A Band-Reconfigurable Antenna Based on Directed Dipole

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IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 62, NO. 1, JANUARY 2014

A Band-Reconfigurable Antenna Based on Directed Dipole Lei Ge, Student Member, IEEE, and Kwai-Man Luk, Fellow, IEEE

Abstract—A novel reconfigurable directed dipole antenna operated in a wideband or four narrowband modes is presented. The wideband mode (0.83–2.5 GHz) is based on a folded bowtie-shaped dipole while the four narrowband modes are based on a lengthreconfigurable thin dipole. A rectangular cavity is introduced to provide the antenna with well-controlled unidirectional radiation patterns. PIN diodes are used as switches at specific locations for choosing different modes. The design procedure is presented in details which maybe a useful guideline. A fully functional prototype is developed and tested, which exhibits good performance. The unidirectional patterns and frequency selective feature make the antenna potentially suitable for fixed facilities in cognitive radio. Index Terms—Cognitive radio, directed dipole, reconfigurable antenna, wideband antenna.

I. INTRODUCTION

T

HE existence of many wireless communication applications, like GSM, 3G, LTE, WLAN, Bluetooth, etc., has made the finite radio-frequency spectrum significantly congested. This attracts increasing interest on the design of band-reconfigurable antennas which can help to use spectrum resources more efficiently whilst guarding against interference. This approach is essential for multimode applications such as cognitive radio (CR) [1], [2]. In general, a possible antenna for cognitive radio consists of a broadband receiving antenna for sensing purpose and a reconfigurable or tunable narrowband communication antenna [3]–[6], or one wide-narrowband reconfigurable antenna for both sensing and communication [7]–[10]. A number of reconfigurable antennas have been demonstrated to meet the first configuration (combing wideband and narrowband antennas using separate excitation ports). In [3], a dual-port wide-narrowband antenna is proposed by a combination of a shorted microstrip patch integrated with a coplanar waveguide (CPW) fed ultrawideband (UWB) antenna. This technique is based on utilizing a relatively large antenna (UWB antenna) that is printed on the top side of a substrate, acting as a ground for a smaller antenna (shorted patch). A dual-port, uniplanar wide-narrowband antenna is demonstrated in [4] by utilizing the space between two tapered slots (a part of an UWB antenna) to integrate a narrowband antenna. In [5], Manuscript received May 31, 2013; revised August 16, 2013; accepted October 11, 2013. Date of publication October 28, 2013; date of current version December 31, 2013. The authors are with the State Key Laboratory of Millimeter Waves, City University of Hong Kong, Hong Kong (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TAP.2013.2287520

a new incorporated planar UWB/reconfigurable-slot antenna is proposed. A slot resonator is precisely embedded in the disc monopole radiator to achieve an individual narrowband antenna. A varactor diode is also inserted across the slot, providing a reconfigurable frequency function. In [6], a wideband and switched subband could be achieved by feeding a different structure through 180 rotational motion. Due to the difficulty in providing sufficient isolation between sensing and communication antennas in limited space, the second approach which uses one antenna for both functions is investigated a lot recently. An L-shaped slot antenna is demonstrated in [7] by using microelectromechanical switches. In [8], a Vivaldi antenna with added switched band functionality by using PIN diodes switches is presented to operate in a wideband or three narrowband modes. The antenna reconfiguration is realized by inserting four pairs of switchable ring slots into the ground plane of the structure. An improved structure is then demonstrated in [9] which uses a single pair of ring slot resonators in the Vivaldi to realize frequency reconfiguration. PIN diode switches are employed at specific locations in the resonator to change its effective electrical length hence forming different filter configurations. Both antennas [8], [9] can obtain directional radiation patterns, but the back lobes are too large which cannot meet the stringent requirements for base station antennas. Although a number of attempts have been proposed to obtain directional radiations for CR antennas [10]–[13], few can achieve good radiation patterns at both wideband and narrowband modes. In this paper, a novel band-reconfigurable antenna based on directed dipole technology is proposed. The antenna is composed of a wideband folded bowtie dipole and a length-reconfigurable narrowband thin dipole. One wideband mode and four narrowband modes can be achieved by controlling the states of the PIN diode switches. The polarizations of the wideband and narrowband modes are maintained as the same linear polarization. To demonstrate its functionality, a prototype is fabricated and measured which shows a wide range of frequency reconfiguration (0.83–2.5 GHz). Moreover, good unidirectional radiation patterns with very low back radiation levels are achieved which indicates the antenna’s capability for outdoor base station in future communication. Details of the proposed design are described as follows. Narrowband and wideband antenna designs are presented in Sections II and III, respectively. In Section IV, the antenna reconfiguration with DC lines is presented and discussed. An example is built with results in Section V followed by conclusions in Section VI.

0018-926X © 2013 IEEE

GE AND LUK: A BAND-RECONFIGURABLE ANTENNA BASED ON DIRECTED DIPOLE

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Fig. 2. Equivalent circuits of the PIN diode: (a) ON state; (b) OFF state.

Fig. 3. Simulated S11 of narrowband antenna with PIN diodes.

TABLE I STATES OF THE SWITCHES FOR DIFFERENT BANDS

Fig. 1. Geometry of the thin directed dipole: (a) 3D view; (b) top view of the dipole; (c) balun. (Unite: mm).

TABLE II VALUES OF THE ELEMENTS OF THE EQUIVALENT CIRCUIT

II. NARROWBAND ANTENNA DESIGN A length-reconfigurable dipole antenna is designed to work as the narrowband antenna. As Fig. 1 shows, the antenna consists of a length-reconfigurable thin dipole, a vertically oriented balun, and a rectangular cavity. The rectangular cavity is used for the dipole to possess directional radiation patterns. The thin dipole is printed on the top layer of the horizontal substrate with and . To feed the dipole, a printed balun ( , ) is placed vertically on the center of the ground. The balun is a microstrip-tostripline transition from a 50 microstrip line to a 70 balanced stripline as Fig. 1(c) describes. The balanced stripline connects to two small printed patches on the bottom layer of the horizontal substrate and then connect to the thin dipole by vias. A number of gaps are created along the thin dipole to divide the dipole into several parts. Some switches are placed to bridge the gaps. Then, the dipole’s length can be controlled by setting the switches ON or OFF. Therefore, the narrowband dipole can operate at different frequencies by setting the states of the switches.

In this design, each switch is realized by two PIN diodes in series, mode Infineon BAR50-02V. Each diode can be forward biased to ON state with a DC voltage that provides 100 mA biasing current, whereas it will be in OFF state if left unbiased. The equivalent circuit of the PIN diode is described in Fig. 2, while the values of the elements at the chosen biasing condition are given in Table II [14]. All the DC lines and the DC decoupling components are not included here and will be discussed later at Section IV. Fig. 3 shows the simulated results of the narrowband dipole with PIN diodes as described byTable I. Four narrowband modes can be achieved at 1.1 GHz, 1.4 GHz, 1.75 GHz and 2.35 GHz, respectively. On the other hand, the operating frequency is out of the band from 0.8 GHz to 2.5 GHz when all the four switches are OFF. To see the ON and OFF effects of the PIN diodes, simulated current distributions of half thin dipole at NB3 mode are

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Fig. 4. Simulated current distribution of half thin dipole with different numbers of diodes as a switch: (a) No diode; (b) One diode as a switch; (c) Two diodes as a switch.

Fig. 6. 3D view of the folded bowtie dipole.

Fig. 5. Simulated radiation efficiency of NB3 with different numbers of diodes as a switch.

shown in Fig. 4. When no diode is used, ON state is represented by short circuit and OFF state is represented by open circuit, which means the switch is ideal. As Fig. 4(a) shows, slight power is coupled to the opened parts of the dipole. When only one diode is used as a switch, some power goes through SW3 as shown in Fig. 4(b). When two diodes are used as Fig. 4(c) shows, the passed power is reduced. Therefore, two diodes can achieve better OFF state than one diode, which is helpful for band selectivity. Fig. 5 shows the simulated radiation efficiency of NB3 when different numbers of diodes are used as a switch. As we can see, very high efficiency can be obtained when no diode is used. When diodes are introduced, power consumption is increased greatly. Moreover, about 3 percent loss is increased when two diodes instead of one are used as a switch at the operating frequency 1.75 GHz. Therefore, two diodes lead to worse ON state than one diode, which brings more power loss. In the proposed design, two diodes are used for switch’s better OFF state at the cost of more power loss. Furthermore, in terms of impedance matching, the S11 of NB3 shifts to lower frequency when one diode is used instead of two. This is because more power goes through SW3 which corresponds to lengthening the dipole. Further work might reveal alternative ways to realize better switches with more ideal ON and OFF states simultaneously.

Fig. 7. Simulated current distribution of the bowtie dipole at 1.5 GHz.

III. WIDEBAND ANTENNA DESIGN A folded bowtie dipole has been chosen as a basic structure of the wideband antenna due to its broadband properties. The antenna is based on [15] where the design guideline can be found, and has been scaled to operate over a band of 0.8–2.5 GHz. As Fig. 6 describes, the wideband antenna is composed of a folded bowtie dipole, a vertically oriented balun, and a rectangular cavity. The balun and the cavity have the same dimensions as that of the narrowband antenna. The bowtie dipole is also printed on the top layer of the horizontal substrate ( , ) and fed by balun. Fig. 7 shows the simulated current distribution of the bowtie dipole at 1.5 GHz which indicates that the current mainly concentrates on the edges of the bowtie. Therefore, some portions in the center of the bowtie as Fig. 6 shows can be removed in order to make space for the thin dipole while maintaining the performance of the wideband antenna. Fig. 8 describes the performances of the wideband antenna with and without the removed portions, which show similar characteristics. Meanwhile, the bowtie dipole without the removed portions can achieve a wideband response from 0.9 to 2.4 GHz. In addition, the bowtie dipole maintains the same linear polarization as that of the narrowband thin dipole antenna, which is needed for the purpose of the application.


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Fig. 8. Simulated S11 and realized gain of the wideband antenna with and without the removed portions.

IV. ANTENNA RECONFIGURATION AND DISCUSSIONS The wide-narrowband antenna reconfiguration is demonstrated here by combining the wideband bowtie antenna and the narrowband length-reconfigurable antenna. Beside the antenna structure, the DC lines are added to bias the switches. A. Layout of Antenna Fig. 9 shows the layout of the proposed antenna which is mainly composed of a folded bowtie dipole, a length-reconfigurable thin dipole, a vertically oriented balun, a rectangular cavity and vertically oriented DC lines. The antenna has the same dimensions as that of the narrowband and wideband antennas. As the antenna geometry is symmetric along both x- and y-axes, only some of the top view layout in Fig. 9(b) is enlarged to describe the structure clearly. There are 5 types of switches, where SW0 and SW0 control the state of the wideband antenna and the other four kinds of switches control the length of the thin dipole. The wideband mode and four narrowband modes can be achieved by choosing different states of the switches as indicated inTable III. Some switches (SW0 ) are placed in the center of the bowtie dipole to cut the bowtie into several parts. When the antenna works at wideband mode, SW0 and SW0 are both ON and the bowtie dipole can work as a whole folded dipole structure. On the other hand, when the antenna works at narrowband mode, SW0 and SW0 are both OFF and the bowtie dipole is cut into several parts. So the radiated signal at the bowtie cannot resonate within the request operating bands. If SW0 is not introduced which means the bowtie dipole is always a whole structure, the wideband performance will not be affected while the narrowband performance will become bad. Fig. 10 shows the comparison of the simulated radiation efficiency at NB3 and NB4 with and without SW0 . As the figure shows, the radiation efficiency is very low when SW0 is not introduced, which results in poor performance of NB3 and NB4. Thus, SW0 is important for the thin dipole getting rid of the bowtie’s interruption. B. Layout of DC Lines Several vertical DC lines are added to connect the antenna to biasing sources through the slots in the ground planes. As Fig. 11 shows, the DC lines with 0.1 mm width are printed on

Fig. 9. Geometry of the proposed antenna: (a) 3D view; (b) top view.

TABLE III STATES OF THE SWITCHES FOR DIFFERENT BANDS

the vertically oriented substrate with and . On the dipoles, i.e. the antenna radiation parts, 22 pF SMD capacitors are used for DC blockage and RF connectivity. The DC lines are isolated from the RF signal by Murata ferrite beads, model BLM18G [16]. This kind of ferrite bead characters very low resistance for DC signal (about 1 ) but very high resistance for RF signal (over 1 for frequency between 0.8

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Fig. 12. Current distribution when no ferrite bead is added along the vertical DC lines.

Fig. 10. Simulated radiation efficiency of NB3 (blue lines) and NB4 (red lines) when SW0 is introduced (dashed lines) and not introduced (solid lines).

Fig. 13. Simulated S11 and realized gain of NB2 with and without DC lines.

point when no DC lines are added while reducing to 5.2 dBi when the DC lines are added. There are two main factors leading to the gain difference. One is the loss introduced by the DC lines. The other one is the beamwidth difference. From a directivity-beamwidth relationship, wider beamwidth will result in lower maximum directivity for directional patterns hence lower broadside gain. For example, it is observed from Fig. 14 that the simulated beamwidth of NB2 with DC lines is wider than that without DC lines. Table IV shows the beamwidth of the two situations. For directional patterns, the maximum directivity is given by

Fig. 11. Geometry of the DC lines.

GHz and 3 GHz). Therefore, some ferrite beads are used as RF chokes while affecting the DC biasing little. Besides, a lot of ferrite beads are used along the vertical DC lines to cut off the radiated RF signal while allowing the DC signal to pass. Fig. 12 describes the current distribution when no ferrite bead is added along the vertical DC lines. As we can observe, the current on the DC lines is as strong as that on the dipole, which will affect the antenna performance a lot. Owning to the addition of many ferrite beads, the vertical DC lines works like absorbers which will affect the resonant frequencies and gains of the narrowband modes. Fig. 13 shows the comparison of the performances of NB2 with and without DC lines. The resonant frequency shifts from 1.5 GHz to 1.2 GHz. Moreover, the broadside gain is about 10.3 dBi at matching

where

beamwidth in one plane, beamwidth in a plane at a right angle to the other [17]. Therefore, the gain degradation can be written as

Thus, about 3.2 dB gain difference is due to the loss of DC lines, which means the efficiency of the antenna is degraded a lot by the DC lines. V. RESULTS A prototype of the antenna as pictured in Fig. 15 was built to verify the proposed design. In the practical fabrication, a number of plastic right-angled fixtures were used to fix and support the PCB substrates. Simulation was accomplished using Ansoft HFSS [18]. Measured results of S11, antenna gain and


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Fig. 14. Simulated radiation patterns of NB2 with and without DC lines (where E-plane is XOZ plane and H-plane is YOZ plane).

TABLE IV BEAMWIDTH OF NB2 WITH AND WITHOUT DC LINES

Fig. 16. S11 and realized gain of the wideband and narrowband modes: (a) Simulated and measured S11; (b) Simulated (dashed lines) and measured (solid lines) realized gain.

Fig. 15. Photo of the fabricated antenna.

radiation patterns were obtained by an Agilent N5230A network analyzer and a Satimo Starlab near-field measurement system. A. Impedance Matching The wideband mode and four narrowband modes can be achieved by choosing different states of the diodes as indicated inTable III. Fig. 16(a) shows the simulated and measured S11 of the antenna with good agreement. The measured impedance bandwidth of wideband mode is 100.3% for from 0.83 to 2.5 GHz. Four narrowband modes can be achieved at 1.05 GHz, 1.2 GHz, 1.5 GHz and 2.1 GHz, respectively. Good agreement between the simulated and measured results is obtained. At each narrowband mode, the measured S11 is less than the simulated one which is due to the introduced loss in practical design. From the comparison between Fig. 2 and Fig. 16(a), the operating bands of the final design shift to lower frequencies because

of the DC lines introduced. Therefore, the band shift should be considered at practical application. The bands can be shifted back by changing the length of the thin dipole. B. Gain and Radiation Pattern Fig. 16(b) shows a broadside gain of 7.6 2.4 dBi can be measured at the wideband mode. The simulated and measured radiation patterns at frequencies of 1.05 GHz, 1.2 GHz, 1.5 GHz and 2.1 GHz are shown in Fig. 17. Over the operating frequencies, the main beam of the radiation pattern is fixed in the broadside direction. The measured front-to-back ratios are above 15 dB and the cross polarization levels are always below 18 dB over the whole frequency band. Fig. 16(b) also describes the simulated and measured broadside gains of the narrowband modes at the operating bands where the measured gains are about 1 dB smaller than the simulated ones. Fig. 18 gives the far-field radiation patterns at the four narrowband modes. The main beam of the radiation pattern is fixed in the broadside direction at all the bands. Furthermore, the measured front-to-back ratios are above 20 dB and the cross polarization levels are all below 18 dB.

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Fig. 17. Radiation patterns of wideband mode at 1.05 GHz, 1.2 GHz, 1.5 GHz and 2.1 GHz.

From the comparison between Fig. 8 and Fig. 16(b), the wideband gain degradation caused by switches and DC lines is about 2 dB, which is much smaller than the degradation at narrowband modes. This is because the radiation part of wideband dipole mainly concentrates on the edges of the bowtie as shown in Fig. 7. Therefore, the ferrite beads on the DC lines do not absorb as much power as that at narrowband modes whose radiation parts are just above the DC lines. On the other hand, much power is lost at narrowband modes because of the switches and DC lines. Although the gains of narrowband modes are low, good radiation patterns are obtained which is important for the

Fig. 18. Radiation patterns of narrowband modes at 1.05 GHz, 1.2 GHz, 1.5 GHz and 2.1 GHz.

purpose of application. Due to the low efficiency of the narrowband antenna, amplifier with higher gain may be needed in the system for practical application. VI. CONCLUSION A novel band-reconfigurable antenna based on directed dipole is proposed. One wideband mode is achieved by a folded bowtie dipole, while four narrowband modes are realized by a length-reconfigurable thin dipole. Different modes are achieved by setting the states of PIN diode switches. Some DC lines are added to bias the switches, which, however, introduce a lot of


losses. A prototype is demonstrated which shows good return loss and unidirectional radiation patterns for each mode. With the frequency selective feature and unidirectional patterns, the proposed antenna may find potential applications in fixed facilities for multimode applications, such as cognitive radio. However, due to the switches and DC lines introduced, the disadvantages are high loss or low efficiency. Moreover, the design is complicate for achieving these functionalities. Further work may focus on improving the antenna efficiency and simplifying the design complexity. REFERENCES [1] J. Mitola and G. Q. Maguire, “Cognitive radios: Making software radios more personal,” IEEE Pers. Commun., vol. 6, no. 4, pp. 13–18, 1999. [2] P. S. Hall, P. Gardner, and A. Faraone, “Antenna requirements for software defined and cognitive radios,” Proc. IEEE, vol. 100, no. 7, pp. 2262–2270, 2012. [3] E. Ebrahimi, J. Kelly, and P. S. Hall, “Integrated wide-narrowband antenna for multi-standard radio,” IEEE Trans. Antennas Propag., vol. 59, no. 7, pp. 2826–2835, Jul. 2011. [4] G. Augustin and T. A. Denidni, “An integrated ultra wideband/narrow band antenna in uniplanar configuration for cognitive radio systems,” IEEE Trans. Antennas and Propagat., vol. 60, no. 11, pp. 5479–5484, Nov., 2012. [5] E. Erfani, J. Nourinia, C. Ghobadi, M. Niroo-Jazi, and T. A. Denidni, “Design and implementation of an integrated UWB/reconfigurable-slot antenna for cognitive radio applications,” IEEE Antennas Wireless Propag. Lett., vol. 11, pp. 77–80, 2012. [6] Y. Tawk and C. G. Christodoulou, “A new reconfigurable antenna design for cognitive radio,” IEEE Antennas and Wireless Propagat. Lett., vol. 8, pp. 1378–1381, Dec. 2009. [7] L. Zidong, K. Boyle, J. Krogerus, M. de Jongh, K. Reimann, R. Kaunisto, and J. Ollikainen, “MEMS-switched, frequency-tunablehybrid slot/PIFA antenna,” IEEE Antennas Wireless Propag. Lett., vol. 8, pp. 311–314, Feb. 2009. [8] M. R. Hamid, P. Gardner, P. S. Hall, and F. Ghanem, “Switched-band Vivaldi antenna,” IEEE Trans. Antennas Propag., vol. 59, no. 5, pp. 1472–1480, May 2011. [9] M. R. Hamid, P. Gardner, P. S. Hall, and F. Ghanem, “Vivaldi antenna with integrated switchable band pass resonator,” IEEE Trans. Antennas Propag., vol. 59, no. 11, pp. 4008–4015, Nov. 2011. [10] M. R. Hamid, P. Gardner, and P. S. Hall, “Reconfigurable log periodic aperture fed microstrip antenna,” in Proc. LAPC Conf. Antennas Propag., Loughborough, U.K., 2009, pp. 237–239. [11] T. Wu, R. L. Li, S. Y. Eom, S. S. Myoung, K. Lim, J. Laskar, S. I. Jeon, and M. M. Tentzeris, “Switchable quad-band antennas for cognitive radio base station applications,” IEEE Trans. Antennas Propag., vol. 58, no. 5, pp. 1468–1476, May 2010. [12] H. H. Moghadam, A. Mirkamali, and P. S. Hall, “Using printed dipole antenna and PIN diodes for wideband frequency reconfiguration,” in Proc. LAPC Conf. Antennas Propag., Loughborough, U.K., 2010, pp. 381–384. [13] Y. Cai, Y. J. Guo, and T. S. Bird, “A frequency reconfigurable printed Yagi-Uda dipole antenna for cognitive radio applications,” IEEE Trans. Antennas Propag., vol. 60, no. 6, pp. 2905–2912, Jun. 2012. [14] BAR50 Series Infineon PIN Diode Datasheet.

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[15] S. W. Qu, J. L. Li, Q. Xue, C. H. Chan, and S. Li, “Wideband and unidirectional cavity-backed folded triangular bowtie antenna,” IEEE Trans. Antennas Propag., vol. 57, pp. 1259–1263, Apr. 2009. [16] BLM18G Series Murata Ferrite Bead Datasheet. [17] C. A. Balanis, Antenna Theory: Analysis and Design, 3rd ed. New York, NY, USA: Wiley, 2005. [18] Ansoft Corp., “HFSS: High frequency structure simulator based on the finite element method,” [Online]. Available: http://www.ansoft.com Lei Ge (S’11) was born in Jiangsu, China, in 1987. He received the B.S. degree in electronic engineering from Nanjing University of Science and Technology, Nanjing, China, in 2009. Currently, he is working toward the Ph.D. degree in electronic engineering at City University of Hong Kong. From September 2010 to July 2011, he was a Research Assistant with the City University of Hong Kong. He received the Honorable Mention at the student contest of 2012 IEEE APS-URSI Conference and Exhibition held in Chicago, US. His recent research interest focuses on wideband antennas, patch antennas, base station antennas, reconfigurable antennas and the antennas for cognitive radio.

Kwai-Man Luk (M’79–SM’94–F’03) was born and educated in Hong Kong. He received the B.Sc. (Eng.) and Ph.D. degrees in electrical engineering from The University of Hong Kong, Hong Kong, in 1981 and 1985, respectively. He joined the Department of Electronic Engineering, City University of Hong Kong, in 1985 as a Lecturer. Two years later, he moved to the Department of Electronic Engineering, Chinese University of Hong Kong, where he spent four years. He returned to the City University of Hong Kong in 1992, and is currently Chair Professor of Electronic Engineering and Director of State Key Laboratory in Millimeter waves (Hong Kong). He is the author of three books, nine research book chapters, over 290 journal papers and 220 conference papers. He has received five US and more than 10 PRC patents. His recent research interests include design of patch, planar and dielectric resonator antennas, and microwave measurements. Prof. Luk is a Fellow of the Chinese Institute of Electronics, PRC, a Fellow of the Institution of Engineering and Technology, U.K., a Fellow of the Institute of Electrical and Electronics Engineers, USA and a Fellow of the Electromagnetics Academy, USA. He is Deputy Editor-in-Chief of PIERS journals. He was a Chief Guest Editor for a special issue on “Antennas in Wireless Communications” published in the Proceedings of the IEEE in July 2012. He was Technical Program Chairperson of the 1997 Progress in Electromagnetics Research Symposium (PIERS), General Vice-Chairperson of the 1997 and 2008 Asia-Pacific Microwave Conference (APMC), General Chairman of the 2006 IEEE Region Ten Conference (TENCON), Technical Program Co-chairperson of 2008 International Symposium on Antennas and Propagation (ISAP), and General Co-chairperson of 2011 IEEE International Workshop on Antenna Technology (IWAT). He received the Japan Microwave Prize at the 1994 Asia Pacific Microwave Conference held in Chiba in December 1994, and the Best Paper Award at the 2008 International Symposium on Antennas and Propagation held in Taipei in October 2008. He was awarded the very competitive 2000 Croucher Foundation Senior Research Fellow in Hong Kong and the 2011 State Technological Invention Award (2nd Honor) of China.