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Jul 30, 2014 - Abstract—A bidirectional buck–boost dc–dc converter and its control is presented for vehicle charging as well as for vehicle-to- grid energy ...
IEEE JOURNAL OF EMERGING AND SELECTED TOPICS IN POWER ELECTRONICS, VOL. 2, NO. 3, SEPTEMBER 2014

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A Bidirectional DC–DC Converter With Overlapping Input and Output Voltage Ranges and Vehicle to Grid Energy Transfer Capability Mehnaz Akhter Khan, Student Member, IEEE, Iqbal Husain, Fellow, IEEE, and Yilmaz Sozer, Member, IEEE Abstract— A bidirectional buck–boost dc–dc converter and its control is presented for vehicle charging as well as for vehicle-togrid energy transfer. The cascaded buck–boost topology allows overlapping input and output voltage ranges and a higher intermediate dc-bus voltage where the electric drivetrain traction inverter can be connected. The intermediate dc-link capacitor voltage is varied to improve the transient performances of the converter. The reference voltage of this capacitor is set based on the input and output voltage levels, and the power demand. The modularity, control flexibility, and excellent transient performance of the converter have been verified with simulation and experiments. Index Terms— Bidirectional converter, cascaded buck-boost converter, dc–dc converter, vehicle to grid, vehicles.

I. I NTRODUCTION

T

HE INCREASED penetration of renewable energy sources into the power grid will help ease our dependence on fossil fuel-based energy sources. The energy throughput variability problem associated with the solar, wind, and ocean renewable energy sources can be alleviated through an interconnected energy storage system. The battery packs in a fleet of electric vehicles can serve as an energy storage system that can deliver energy when required. Therefore, electric vehicles can help stabilize the grid when sufficiently large numbers of them are on service. Electric vehicles can provide the grid support by storing energy in their battery packs during periods of low demand and delivering the energy during periods of high demand. A dc–dc converter with bidirectional power flow capability is needed for power transfer in both directions between the grid and an electric vehicle. The bidirectional dc–dc converter in the plug-in electric vehicles should be capable of power compensation, voltage regulation, and peak shaving [1]. The cascaded buck–boost topology used in [2] and [3] has a low maximum overall efficiency and the efficiency drops significantly for large voltage transfer ratios [4]. Thrimawithana and Madawala [5] presented a contactless

Manuscript received August 22, 2013; revised November 11, 2013 and January 9, 2014; accepted January 10, 2014. Date of publication February 7, 2014; date of current version July 30, 2014. Recommended for publication by Associate Editor Akshay K. Rathore. M. A. Khan and I. Husain are with the Electrical and Computer Engineering Department, North Carolina State University, Raleigh, NC 27606 USA (e-mail: [email protected]; [email protected]). Y. Sozer is with the Electrical and Computer Engineering Department, University of Akron, Akron, OH 44325 USA (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/JESTPE.2014.2305157

bidirectional converter topology with reversible rectifiers on each side of an inductive power transfer system to control the amount and direction of power flow. Garcia et al. [6] used interleaved buck converters to reduce the filtering requirement, and to improve the dynamic response and power management [6]. A multilevel converter topology was used to decrease the voltage stress on the transistors and to reduce the need for large inductors. In the research presented in this paper, a cascaded buck–boost dc–dc converter is presented to transfer power between the vehicle batteries and the grid. The proposed nonisolated dc–dc converter has simpler structure, high reliability, high efficiency, and low parts count compared with other converters in this category [7]–[12]. The discussion in this paper is limited to nonisolated converters, and the readers are referred to the references for discussions on the class of converters with electrical isolation [13]–[15]. The developed converter allows the input and output voltage ranges to overlap with a capacitor bank in the middle providing an intermediate stage. An alternative nonisolated dc–dc converter topology uses an inductor in the intermediate stage [4]. However, the developed topology provides modular design and allows individual modules to be interleaved on either the input or the output side; this allows component size reduction and integration of multiple vehicles into the charging system. II. V EHICLE - TO -G RID P OWER E XCHANGE The bidirectional dc–dc converter is part of the vehicle high-power electrical system that connects the battery with the dc bus of the grid interface; the inverter is assumed to be part of the grid interface infrastructure which will be in the charging station of a parking lot. A number of vehicles with a dc output are envisioned to be connected to the dc bus of the charging station, as shown in Fig. 1. If the battery of the vehicle is undercharged and needs energy, it will then draw power from the grid; conversely, if the battery has reserve capacity more than it needs for the commute and the power demand on the grid is high, then it will deliver power to the grid. The power exchange between the grid and the vehicle is required for a large number of different types of vehicles with different input and output voltage ranges. A bidirectional dc–dc converter that can operate in both buck and boost modes with a wide range of voltage levels in either direction is necessary to achieve the objective [4]. The converter developed and presented has the capability to allow a wide and overlapping input and output voltage ranges.

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Fig. 1.

IEEE JOURNAL OF EMERGING AND SELECTED TOPICS IN POWER ELECTRONICS, VOL. 2, NO. 3, SEPTEMBER 2014

Structure of the V2G interface.

Fig. 3.

Fig. 2.

Proposed dc–dc converter.

The input and output side controls of the proposed converter are independent of each other, which make the topology suitable for multiinput and multioutput applications and EV charging station. The converter topology is suitable for a vehicle architecture where dc-link bus voltage can be flexibly regulated for adding or removing auxiliary components in the dc-link, such as in a heavy hybrid vehicle. This converter can perform boost and buck operation in both directions, which would be useful in certain vehicle systems, such as fuel cell electric vehicles. The comparison of the proposed converter with the converter topology where the inductor is in the middle appears in [16]. III. C ASCADED DC–DC C ONVERTER A. Converter Topology The dc–dc converter plays a vital role for power exchange between vehicle and grid allowing power transfer in either direction according to the need. A boost converter is often used in the electric drivetrain of an electric/hybrid vehicle to boost the battery voltage so that the dc-link voltage supplying the traction inverter operates at a higher voltage. However, the dc-bus voltage range of the charging station may need to be at a lower voltage. The battery voltage also varies around a nominal voltage which could overlap with the dc-bus voltage range of the charging station. The developed converter topology that is capable of operating in both buck and boost modes in either direction is shown in Fig. 2. The topology is suitable for the interfacing the high-voltage battery with the high-voltage dc-bus where an electrical isolation is not required. The cascaded buck–boost converter has an intermediate stage to store energy at a higher voltage and allows the overlap between battery voltage and dc-bus voltage in the whole operating range. The high-efficiency operation is

Flowchart of the operating principles.

achieved in the topology through the switching of only one switch in a bridge, and reducing the voltage stress on the switches through the bridge. The power is transferred between the input and output dc stages utilizing the dc-link capacitor as an intermediate storage unit. The converter has two stages each one of which can act either as a buck converter or a boost converter. For example, during the power flow from the battery to the dc source, stage-1 will act as a boost converter and stage-2 will act as a buck converter. The dc-link capacitor voltage is maintained at a level set by the converter controller. Conversely, when power is to flow from the dc bus to the battery, stage-2 operates in the boost mode and stage-1 operates in the buck mode. Half of the switches are utilized for power flow control in any given direction. The controller turns off the pulse-width modulation signals when power flow direction is required to be changed and waits for the inductor current to reach zero. Then, the controller starts operation with new control commands based on the new operating condition. The flowchart of the controller operation is shown in Fig. 3. The intermediate stage and dc-bus voltage is regulated using two separate PI controllers. The set point of the intermediate capacitor voltage is adjusted according to the battery state-ofcharge and load voltage. In a single-input and single-output application, the set point for the intermediate voltage will be close to the output voltage. In multiple output applications, such as in heavy duty hybrid electric vehicles, this voltage can be set to the larger output voltage level. When there are undesired variations in one of the output voltages due to load disturbances, the intermediate stage is set to a higher level to protect power flow from one output stage to the other lower level output stages. The loads are not affected by one other with this approach. The converter has the inherent capability to enhance the stability of the multioutput system. B. Component Sizing The half bridge switch configuration helps to reduce the switching stresses compared with the single switch topologies;

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Fig. 4.

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CCM inductor current. Fig. 6.

DCM inductor current.

Fig. 7. DCM power transfer as a function of on-time t1 at different input voltages. Fig. 5.

Inductor size depending on the power transfer and current ripple.

switching at higher frequencies helps to minimize the reactive component sizes. The size of the inductors depends on the maximum allowable ripple in the inductor current and the size of the capacitors depends on the maximum allowable ripple on the capacitor voltage. The size of these components can be obtained from the following relations: Vbatt_ max D1_ max T I L1 (Vc1_ max − Vc2_ min )D2_ max T L2 = I L2

L1 =

I L1_ max D2_ max T Vc1 I L2_ max D2_ max T C2 = . Vc2

(1)

C1 =

(2)

Here, T is switching period; D1_ max T and D2_ max T depend on the maximum on-time of S1 , and S4 . I L1 , I L2 , Vc1 , and Vc2 are the maximum allowable ripple currents and ripple voltages of the circuit reactive components. Power transfer characteristics are analyzed separately in the continuous conduction mode (CCM) and discontinuous conduction mode (DCM). Io1 is the inductor steady-state current. Fig. 4 shows the typical CCM inductor current waveform. In CCM, the power transfer equation can be derived as (Vc1 − Vin )2 T Vin 2Vc1 L 1   in Vin T (Vin − Vc1 ) 1 − 2V Vc1

P1 = Vin Io1 + +

2L 1

.

(3)

At different power levels, selection of the inductance value depends on the current ripple allowed in the inductor. The inductance can be chosen from the characteristics of Fig. 5 based on the design specifications for different power levels

Fig. 8.

Pulse positioning. (a) Optimum. (b) Arbitrary.

For example, if an input voltage is 300 V at 10 kW and with a specification of maximum 40% current ripples; the inductor value should be chosen as 600 μH. Fig. 6 shows the typical DCM current waveform. In DCM, the power transfer equation is  2 (Vc1 − V in )2 t22 − t 1 Vin2 t12 − (4) P1 = 2T L 1 2L 1   2 Vin . (5) t2 = t1 1 − Vc1 − V in The power transfer as a function of on-time t1 in the DCM power transfer for three input voltage levels is shown in Fig. 7. C. Optimum Pulse Positioning An optimum positioning of the gate pulses for the input and output stage switches is required to minimize the current ripple in the intermediate stage capacitor current. The optimized pulse positioning also helps to reduce the size of the capacitor. Fig. 8 shows the optimum and arbitrary pulse positioning. The amount of energy stored and transferred via the intermediate stage capacitor depends on the capacitor voltage level and the overlapping percentage of the input and output stage switch pulse positions. If input and output are the same and the pulses are arbitrarily positioned, then more energy may end up being stored in the capacitor before being transferred

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Fig. 9. Energy transfer through the intermediate stage capacitor with % of overlap of two gate signals for (a) Vin = Vo , (b) Vin < Vo , and (c) Vin > Vo .

to the output; this will result in higher ripple in the capacitor current, which will require a larger capacitor. However, with the pulse positioning technique, energy transfer to the output will be guaranteed to be more direct, which will ensure the minimum possible capacitor current ripple. If input and output voltages are the same, the intermediate capacitor do not need to store energy that needs to be transferred to the load with zero overlapping percentage of two gate signals, as shown in Fig. 9(a); the input energy flows to the load directly irrespective of the intermediate stage voltage level. When overlapping percentage is 100%, the voltage change at the intermediate stage is  T /2 1 ILoad T (6) ILoad dt = vc1 = C1 0 C1 2

Fig. 10.

Optimum pulse positioning for Vin > Vo .

Fig. 11. ripple.

Effect of overlapping of two gate signals on the capacitor current

Fig. 12.

Relation between capacitance and current ripple.

and energy transfer from the intermediate stage capacitor is T . (7) 2 For Vin < Vo , an overlap of the input and output stage pulses is essential, but it should be maintained at the minimum level possible. The energy transfer from C1 for the minimum overlap case is Vc1 ILoad TOLmin where TOLmin is the minimum overlapping time. The capacitor energy is a linear function of the overlapping period. When the overlapping percentage is 100%, the energy transfer amount from the intermediate stage capacitor is E c = Vc1 ILoad

E c = Vc1 ILoad TOLmax

(8)

where TOLmax is the maximum overlapping time. The optimum switching condition is when the overlap is minimum, as shown in Fig. 9(b). For Vin > Vo , there is a region where the input and output pulses can be positioned with no overlap, as shown in Fig. 9(c). The energy transfer amount as a function of overlapping percentage is given by %OL =

E c − Iin (Vin − Vo ) T 100. Iin Vo T

(9)

Fig. 10 shows the simulation with optimum pulse positioning where the input voltage is 350 V, intermediate stage reference voltage is 500 V and output reference voltage is 300 V. The optimum pulse positioning minimizes the intermediate capacitor current ripple, which helps to reduce the capacitor size. Fig. 11 shows the effect of overlapping of two gate signals on the capacitor ripple current. The ripple increases with the increase of overlap of two gate signals. For Vin = Vo , minimum overlap is zero. For Vin > Vo , the minimum ripple

is obtained at zero percentage of overlapping. However, when Vin < Vo , there is a minimum overlap, which cannot be avoidable. The analysis has been done for the particular values of Vin = 200 V, intermediate stage voltage, Vc1 = 500 V and Vo = 300 V. For the particular values of Vin and Vo , the ripple is minimum as 20%, as given in Fig. 11. The relationship between capacitance and current ripple is given in Fig. 12 as a reference [17]. For a particular value of capacitor voltage, the current ripple increases with increased capacitance. The minimum overlapping cannot be avoided and current ripple is also unavoidable when the pulses overlap, as shown in Fig. 11. The relationship between the capacitor value and current ripple at different voltage levels is given in Fig. 12.

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Fig. 13.

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Effect of input voltage on the system poles.

IV. C ONTROLLER A NALYSIS AND D ESIGN The controller is designed with parameters that ensure stable operation. A small signal model for the converter system has been developed to aid the selection of the appropriate controller parameters.

Fig. 14.

Effect of input side inductance value on the system poles.

Fig. 15.

Effect of load changing on the system pole pairs.

A. Small Signal Model The stability analysis of the converter for the vehicle-to-grid (V2G) application is presented in this section. Considering the internal resistance of the inductors and capacitors, the system matrix of the converter can be derived in (10), as shown at the bottom of this page. Here, D1 stands for duty cycle of  S1 gate in Fig. 2 and D1 = 1 − D1 , r L1 and r L2 are internal resistances of inductors L 1 and L 2 , respectively. rc1 and rc2 are ESRs of capacitor C1 and C2 , respectively; R L represents the load. The analysis of the system for a 9-kW converter with parameters given in Appendix A shows that the real part of the system poles are negative for the input voltage variations from 100 to 400 V (Fig. 13); this means that the system is stable for the designed input voltage range. The inductors in the converter are sized based on the ripple current limit specified. The ripple current requirement can be relaxed to reduce the values of the inductor. A fault in the inductor may result in its value going all the way down to zero, but the analysis shows that the system is still stable with negative values for the real part of the poles (Fig. 14). The range of load for which the system is stable can be determined from a similar analysis for the given system (Fig. 15). ⎛   D1 r c1 ⎜− L1 + ⎜  ⎜ D1 r c1 ⎜ A=⎜ L2 ⎜  D1 ⎜ ⎝ C1 0 ⎛ 1 ⎞

 −



v˜ c1 =   d˜1 2 s+ s 2 + RD L C1

D12 L 1 C1



D1 r c1 L1 rc1 L2

+

r L2 L2

+

rc2 R L rc2 L 2 +R L L 2



 − DL11

− C11

RL C 2 (rc2 +R L )

 C=







r L1 L1

L

⎜ 1⎟ ⎜ 0 ⎟ ⎟ B =⎜ ⎜ 0 ⎟, ⎝ ⎠ 0

The dynamic behavior of the circuit may shift away from the normal behavior due to disturbance in source, load, circuit parameters, and perturbation in switching time [18]. An average modeling approach can be used to design the controller that manages the dynamics. The perturbation of output voltage depends on the perturbed duty cycle d˜1 and perturbed intermediate stage voltage v˜ c1 . Again, intermediate stage voltage changes with the small change of duty cycle d˜1 . From the average model of the converter, the transfer function from the duty cycle to intermediate voltage is given by

 RL rc2 R L 0 , 0 rc2 + R L rc2 + R L

1 L2

0 0



I L1 C1

 s−

D1 Vc1 I L1 L 1



.

(11)

⎞ 0

⎟ ⎟ ⎟ RL − rc2 L +R L L 2⎟ ⎟ 2 ⎟ ⎟ 0 ⎠ 1 − C2 (rc2 +R L )

D=0

(10)

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Fig. 16.

Fig. 17.

Fig. 18. voltage.

Average model outputs. (a) Intermediate stage voltage. (b) Output

Fig. 19. voltage.

Actual model outputs. (a) Intermediate stage voltage. (b) Output

Average model of the converter.

Converter system with PI controller. TABLE I I NPUT, O UTPUT, AND DC-L INK C APACITOR V OLTAGE R ANGES

The transfer function from output to duty cycle is Vc1

v˜ o  L 2 C 2 =  ˜ d2 s 2 + R L1C2 s +

1 L 2 C2

.

(12)

B. Controller Parameter Selection The analysis with the average model as Fig. 16 helps to evaluate the system behavior due to the perturbation of the system parameters. Two PI controllers have been used as Fig. 17 to control the intermediate stage voltage and output voltage.r k p and ki for the controller have been chosen depending upon the closed loop poles’ position on the left half s-plane. The loop transfer function is l(s) the sensitivity function is μ(s) = 1/(1 + l(s)) and system transfer function is T (s) = l(s)/(1 + l(s)). The fractional change in the system transfer function for a fractional change in the loop transfer function is d T (s)/T (s) = μ(s)dl(s)/l(s) [18]. The sensitivity function of the system should be stable for system stability. For the boost stage the sensitivity function is (13), as shown at the bottom of this page. The boundary conditions for k p1 and ki1 to keep all the closed loop poles in the left half plane are k p1 < D2 /R L I L1 2 D´1 + D´1 Vc1 k p1 ki1 < . I L1 L 1 The discrete time transfer function for the boost stage is (14), as shown at the bottom of this page.

μ (s) = s3 +



D2 R L C1

The sensitivity function for the buck stage is  s 3 + R L1C2 s 2 + L 21C2 s   μ (s) = k V s 3 + R L1C2 s 2 + L 21C2 + Lp22 Cc1 s+ 2

ki2 Vc1 L 2 C2

. (15)

The boundary conditions for k p2 and ki2 for system stability is ki2 > 0 RC2 ki2 Vc1 − 1 . k p2 > Vc1 The average model wave shapes for the intermediate stage voltage and load voltage follow the same pattern as that obtained from the actual model for the same range of PI controller parameters, as shown in Figs. 18 and 19. V. S IMULATION R ESULTS The proposed dc–dc converter is designed and simulated with overlapping input and output voltage ranges. The input and output voltages specified for the design are given in Table I.

 ´ 2 2 s 2 + LD11C1 s s 3 + RD L C1  2 ´ D´ V k I k − L1C1p1 s 2 + LD11C1 + 1L 1c1C1 p1 −

 I L1 ki1 C1

s+

D´1 Vc1 ki1 L 1 C1

(13)



 Vc1 T 2 R L D´1 − 2T R L I L1 L 1 z 2 + 2Vc1 T 2 R L D´1 z + (2T R L I L1 L 1 + Vc1 T 2 R L D´1 ) G (z) =     2 2 4R L L 1 C1 + 2D2 L 1 T + T 2 R L D´1 z 2 + 2T 2 R L D´1 − 8R L L 1 C1 z + 4R L L 1 C1 − 2D2 L 1 T + T 2 R L D´1

(14)

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Fig. 20. Converter voltages and currents for Vc1ref = 500 V and Voref = 300 V.

Fig. 22. Various losses with load variation for a fixed input voltage of 350 V.

Fig. 21. Transient change of the dc-bus voltage (Vo ) for an instantaneous load change.

Fig. 23.

The simulation has been carried out for a nominal battery voltage of 350 V, intermediate stage reference voltage of 500 V and a grid dc-bus voltage of 300 V. A 9-kW load is connected on the dc bus and the current limit on the dc-bus side is maintained at 30 A. The power transfer from the battery to the dc bus is simulated assuming a state-of-charge for the battery higher than its nominal value and dc-bus voltage lower than its nominal value. Fig. 20 shows that the intermediate stage voltage (Vc1 ) reaches its reference voltage of 500-V output dc-bus voltage (Vo ) attains its reference voltage of 300 V and the output current (Io ) is regulated at 30 A to deliver 9 kW of output power. The transient test is carried out with a step change in the load current. Fig. 21 shows the overshoot in the load voltage (Vo ) and load current (Io ) waveforms for step changes in load, although the effect is moderate. The overshoot varies with the different set points for the intermediate capacitor voltage. The overshoot increases with increasing values of the intermediate capacitor reference voltage set point. The inductor DCR loss, conduction loss, switching loss, and efficiency have been calculated using PLECs circuit simulator for different loads. In case of switching loss, the actual IGBT rise time and fall time have been considered. The efficiency results are higher than that of the conventional buck–boost topology [3]. The losses with load variation for an input

voltage of 350 V are shown in Fig. 22. The switching loss can be observed to be the dominant loss for the entire load range. The inductor DCR loss is the loss in the winding dc resistance loss of the inductor. Fig. 23 shows the efficiency variation with the load and input voltage variations. The efficiency level saturates with the increase of load for a particular input voltage; however, the efficiency increases with the increase of input voltage. In lower input voltage, the efficiency decreases with increasing load since the losses at lower input voltage levels is comparatively higher than that at higher input voltage levels.

Efficiency with load and input voltage variations.

VI. E XPERIMENTAL R ESULTS A prototype converter of the proposed topology has been built using a three-phase, six-switch inverter IGBT module, as shown in Fig. 24. The voltage and current ratings of the switches are 1200 V and 100 A, respectively. Each of the three bridge legs has separate current sensors that can provide three branch currents. The rise time and fall time of the IGBT switches are 40 and 65 ns, respectively. The dc-bus capacitor in the intermediate stage of the converter is 3300 μF. In this experiment, the input and output inductors are connected to the midpoints of the two bridge legs for power transfer.

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TABLE IV VARYING O UTPUT R EFERENCE V OLTAGE

Fig. 24.

Three phase IGBT module.

TABLE V VARYING I NPUT V OLTAGE

Fig. 25.

Controller board. TABLE II M AXIMUM L EVELS FOR E XPERIMENT

TABLE III VARYING I NTERMEDIATE S TAGE R EFERENCE V OLTAGE

Fig. 26. Experimental result showing the initial transient response of voltages and currents.

The control algorithm has been developed using microchip dSPIC33 digital signal processor. The controller board shown in Fig. 25 has the processor, voltage sensor, analog input components, and several fault protection circuit components. All of the fault and diagnostic information is processed in the DSP. The power circuitry for the IGBT modules and the controller electronics have been kept separate from the control circuitry for control signal integrity and EMI noise immunity. The inverter and grid side shown in Fig. 2 has been emulated with resistive loads depending on the power transfer direction. The maximum voltage and current levels for the experiments are given in Table II. The parameter values used are given in Appendix A. The experimental results are given in Tables III–V. The results show that the system follows the reference voltage of the intermediate stage and the load voltage remains unchanged (Table III).

The results in Table IV show that load voltage follows the reference value while others remain the same. Table V shows that the intermediate voltage and load voltage remain the same with changes in the input voltage. The results in Tables IV and V demonstrate that this topology allows the overlapping input and output voltage ranges, while accommodating a wide input voltage range. The system is versatile enough to manage any type of reference change or input supply variation or output load variation. Fig. 26 shows the experimental initial transient response where the intermediate stage voltage (Ch2) follows its reference voltage of 170 V and the load voltage (Ch3) follows its reference voltage of 120 V. The result shows the converter response to a step command input. Fig. 27 shows the steady-state condition for 1.5-kW system output with 150-V input voltage, 172-V intermediate reference voltage, and 146-V load reference voltage. The high-frequency spikes in the waveforms are due to measurement noise. The topology can adequately respond to reference changes in both the intermediate stage and load side simultaneously. Fig. 28 shows the response of the intermediate stage voltage

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Fig. 30. Fig. 27.

Experimental result for 3.8-kW load.

Experimental result for 1.5-kW load. TABLE VI E XPERIMENTAL R ESULTS AT H IGHER V OLTAGE L EVELS

Fig. 28.

Transient change in intermediate stage reference voltage.

TABLE VII P OWER AND E FFICIENCY A NALYSIS

Fig. 29.

Transient change in load reference voltage.

(Ch2) for a step change in reference from 140 to 170 V. In this case the load voltage (Ch3) and load current (Ch4) maintain at the same original values before and after the step command. Similarly, the load voltage also follows the reference command change. Fig. 29 shows the load voltage (Ch3) response for a step command change from 90 to 110 V while maintaining the intermediate stage voltage at its previous value. The load current (Ch4) is following the output voltage change since a resistive load has been used in the experiment. The experimental results have been carried out up to 3.8 kW; the intermediate reference voltage is 300 V and the load reference voltage is 180 V in this case. The 3.8 kW experimental results are shown in Fig. 30. Table VI gives the higher voltage test results with the converter. Experimental efficiency has been evaluated at two representative data points, as shown in Table VII. From simulations, the efficiency at these two data points of 4 and 2 kW are found to be 94.41% and 94.2%, respectively. The experimental efficiency results in Table VII are lower than the simulation results since some of the nonlinear switching behavior and

parasitic effects are not modeled in the simulation. The efficiency numbers are also subject to devices selected in the converter. The efficiency numbers can be improved with the selection of better devices. VII. C ONCLUSION A bidirectional dc–dc converter topology that allows overlap of input/output voltage ranges is presented for electric vehicles with V2G capability. The topology has an intermediate dc-link capacitor that enables the input–output voltage overlap. The use of half-bridge switches in the two stages minimizes the switching stresses that lead to higher efficiencies in the converter. The intermediate dc-bus voltage set point control allows improving the transient operation of the power converter. Component sizing, parameter selection, and control with stability analysis of the converter have been presented. Experimental results conform to the transient behavior and the steady-state efficiency obtained in simulation. The proposed topology provides modularity, control flexibility, and excellent transient performance, while allowing individual modules to be interleaved for electric vehicle applications.

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A PPENDIX A PARAMETER S PECIFICATIONS

Mehnaz Akhter Khan (S’11) received the B.Sc. and M.Sc. degrees in electrical and electronic engineering from the Bangladesh University of Engineering and Technology, Dhaka, Bangladesh, in 2006 and 2009, respectively. She is currently pursuing the Ph.D. degree in electrical engineering from North Carolina State University, Raleigh, NC, USA. Her current research interests include in the area of power electronics, renewable energy, and electric and hybrid vehicles.

R EFERENCES [1] M. C. Kisacikoglu, B. Ozpineci, and L. M. Tolbert, “Examination of a PHEV bidirectional charger system for V2G reactive power compensation,” in Proc. APEC Exposit., Feb. 2010, pp. 458–465. [2] R. M. Schupbach and J. C. Balda, “Comparing DC-DC converters for power management in hybrid electric vehicles,” in Proc. IEEE Int. Electr. Mach. Drives Conf., vol. 3. Jun. 2003, pp. 1369–1374. [3] F. Caricchi, F. Crescimbini, and A. Di Napoli, “20 kW water-cooled prototype of a buck-boost bidirectional DC-DC converter topology for electrical vehicle motor drives,” in Proc. Appl. Power Electron. Conf. Exposit., vol. 2. Mar. 1995, pp. 887–892. [4] S. Waffler and J. W. Kolar, “A novel low-loss modulation strategy for high-power bidirectional buck + boost converters,” IEEE Trans. Power Electron., vol. 24, no. 6, pp. 1589–1599, Jun. 2009. [5] D. J. Thrimawithana and U. K. Madawala, “A contactless bi-directional power interface for plug-in hybrid vehicles,” in Proc. Veh. Power Propuls. Conf., Sep. 2009, pp. 396–401. [6] O. Garcia, P. Zumel, A. De Castro, and J. A. Cobos, “Automotive DC-DC bidirectional converter made with many interleaved buck stages,” IEEE Trans. Power Electron., vol. 21, no. 3, pp. 578–586, May 2006. [7] Y.-J. Lee, A. Khaligh, and A. Emadi, “Advanced integrated bidirectional AC/DC and DC/DC converter for plug-in hybrid electric vehicles,” IEEE Trans. Veh. Technol., vol. 58, no. 8, pp. 3970–3980, Oct. 2009. [8] Y. Park, B. Jung, and S. Choi, “Nonisolated ZVZCS resonant PWM DC-DC converter for high step-up and high-power applications,” IEEE Trans. Power Electron., vol. 27, no. 8, pp. 3568–3575, Aug. 2012. [9] S. Park and S. Choi, “Soft-switched CCM boost converters with high voltage gain for high-power applications,” IEEE Trans. Power Electron., vol. 25, no. 5, pp. 1211–1217, May 2010. [10] M. Ortuzar, J. Moreno, and J. Dixon, “Ultracapacitor-based auxiliary energy system for an electric vehicle: Implementation and evaluation,” IEEE Trans. Ind. Electron., vol. 54, no. 4, pp. 2147–2156, Aug. 2007. [11] M. Gerber, J. A. Ferreira, N. Seliger. and I. W. Hofsajer, “Design and evaluation of an automotive integrated system module,” in Proc. Ind. Appl. Conf., vol. 2. Oct. 2005, pp. 1144–1151. [12] O. Hegazy, J. Van Mierlo, and P. Lataire, “Control and analysis of an integrated bidirectional DC/AC and DC/DC converters for plug-in hybrid electric vehicle applications,” J. Power Electron., vol. 11, no. 4, pp. 408–417, Jul. 2011. [13] F. Z. Peng, L. Hui, G.-J. Su, and J. S. Lawler, “A new ZVS bidirectional DC-DC converter for fuel cell and battery application,” IEEE Trans. Power Electron., vol. 19, no. 1, pp. 54–65, Jan. 2004. [14] T. Mishima, K. Akamatsu, and M. Nakaoka, “A high frequency-link secondary-side phase-shifted full-range soft-switching PWM DC-DC converter with ZCS active rectifier for EV battery chargers,” IEEE Trans. Power Electron., vol. 28, no. 12, pp. 5758–5773, Dec. 2013. [15] B. Gu, J.-S. Lai, N. Kees, and C. Zheng, “Hybrid-switching full-bridge DC-DC converter with minimal voltage stress of bridge rectifier, reduced circulating losses, and filter requirement for electric vehicle battery chargers,” IEEE Trans. Power Electron., vol. 28, no. 3, pp. 1132–1144, Mar. 2013. [16] A. Ahmed, M. A. Khan, M. Badawy, Y. Sozer, and I. Husain, “Performance analysis of Bi-directional DC-DC Converters for electric vehicles and charging infrastructure,” in Proc. IEEE ECCE, Sep. 2013, pp. 1401–1408. [17] KEMET Electronics Corporation, Greenville, SC, USA. (2013, Apr. 15). Capacitor Company [Online]. Available: http://www.kemet.com/ [18] J. G. Kassakian, M. F. Schlecht, and G. C. Verghese, “Dynamics and control: An overview,” in Principles of Power Electronics. Reading, MA, USA: Addison-Wesley, 1991, ch. 11, pp. 253–298.

Iqbal Husain (S’89–M’89–SM’99–F’09) received the B.Sc. degree from the Bangladesh University of Engineering and Technology, Dhaka, Bangladesh, and the M.S. and Ph.D. degrees from Texas A&M University, College Station, TX, USA, in 1987, 1989, and 1993, respectively. He is currently the ABB Distinguished Professor with the Department of Electrical and Computer Engineering, North Carolina State University, Raleigh, NC, USA, engaged in teaching and research. He is the Co-Director of the Advanced Transportation Energy Center, and a faculty member of the NSF FREEDM Engineering Research Center, North Carolina State University. He was with the University of Akron, Akron, OH, USA, prior to joining North Carolina State University, where he built a successful electric and hybrid vehicles program. He was a Visiting Professor with Oregon State University, Corvallis, OR, USA, in 2001. He was a Summer Researcher with Wright Patterson AFB Laboratories, Dayton, OH, USA, in 1996 and 1997. His current research interests include the areas of control and modeling of electrical drives, design of electric machines, development of power conditioning circuits, and design and modeling of electric and hybrid vehicle systems. Dr. Husain received the SAE Vincent Bendix Automotive Electronics Engineering Award in 2006, the College of Engineering Outstanding Researcher Award in 2004, the IEEE Third Millennium Medal in 2000, and the IEEEIAS Outstanding Young Member Award in 1998. He is a recipient of the IAS Magazine Paper Award in 2006 and four IEEE-IAS Committee Prize Paper Awards. He was the Distinguished Lecturer of IAS from 2012 to 2013.

Yilmaz Sozer (M’05) received the B.S. degree in electrical engineering from Middle East Technical University, Ankara, Turkey, and the M.S. and Ph.D. degrees in electric power engineering from the Rensselaer Polytechnic Institute, Troy, NY, USA. His master’s and doctoral research was focused on power electronics and the development of control algorithms for electric machines. He is currently an Assistant Professor with the Electrical and Computer Engineering Department, University of Akron, Akron, OH, USA, engaged in teaching and research. Before joining the University of Akron, he was with Advanced Energy Conversion, Schenectady, NY, USA. His current research interests include the control and modeling of electrical drives, alternative energy systems, design of electric machines, integrated and belt-driven starter/alternator systems, high-power isolated dc/dc converter systems, and static power conversion systems that interface energy storage and distributed generation sources with the electric utility. Dr. Sozer was involved in the IEEE activities, which support power electronics, electric machines, and alternative energy systems. He is serving as an Associate Editor for the IEEE T RANSACTIONS ON I NDUSTRY A PPLI CATIONS and the IEEE T RANSACTIONS ON P OWER E LECTRONICS , and the Vice Chair for the IEEE-IAS Renewable and Sustainable Energy Conversion Systems Committee.