A Comparison Study of High-Frequency Isolated DC/AC Converter ...

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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 8, AUGUST 2014

A Comparison Study of High-Frequency Isolated DC/AC Converter Employing an Unfolding LCI for Grid-Connected Alternative Energy Applications Xiaodong Li, Senior Member, IEEE, and Ashoka K. S. Bhat, Fellow, IEEE

Abstract—A high-frequency (HF) isolated dc/ac converter including an unfolding line connected inverter can be used as the interface between a small-scale alternative energy generation system and the utility line. This paper presents the review of operation of several different topologies of HF isolated dc/ac converters. They are designed for illustration purpose and compared for their performance. It is found that the dual-LCL series resonant dc/ac converter can maintain zero-voltage switching (ZVS) operation for all switches with low line-current total harmonics distortion (THD) and high efficiency. Experimental results on a 500-W prototype converter are included for validation purpose.

Fig. 1.

Grid interface by means of an unfolding LCI.

Index Terms—Alternative energy generation, grid connection, high-frequency (HF) isolation, resonant converters.

I. INTRODUCTION ITH an increasing concern of potential energy crisis and environmental problems, renewable energy generation is gaining more attention now. The output of most renewable generation applications is variable dc, which requires a dc/ac conversion stage for grid connection such as photovoltaic, fuel cell [1]–[7]. For other renewable generation with variable ac output, like wind generation [8]–[10], it is also common to first convert the variable ac-to-dc and then do dc/ac conversion to connect to the utility line. To find a suitable dc/ac converter is vital for renewable generation system. From the literature survey, the earliest use of high-frequency link power conversion concepts was reported in [11]–[13]. These papers used thyristors as the switching devices together with resonant tank circuits to achieve load commutation, and cycloconverter was used on the secondary side to generate the line frequency as (LFAC) voltage waveforms. In [14], a twin resonant tank high-frequency (HF) link was used to generate LF voltage waveform. Two half-bridge parallel resonant converters employing thyristors as the switching devices were used. In all the aforementioned schemes switching frequency was limited to about 3–8 kHz due to the use of thyristors, and variable frequency control was used.

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Manuscript received June 30, 2013; revised September 26, 2013 and November 30, 2013; accepted December 22, 2013. Date of current version March 26, 2014. This paper was supported in part by the Natural Sciences and Research Council of Canada. Recommended for publication by Associate Editor Prof. S. K. Mazumder. X. Li is with the Faculty of Information Technology, Macau University of Science and Technology, Taipa, Macau SAR, China (e-mail: [email protected]). A. K. S. Bhat is with the Department of Electrical and Computer Engineering, University of Victoria, Victoria, BC V8W 3P6, Canada (e-mail: [email protected]). Digital Object Identifier 10.1109/TPEL.2013.2296612

In the literature, different options for grid-connection interface have been discussed [15]–[35]. Among those options, the connection with a line connected inverter (LCI) or unfolding inverter (UFI) using forced commutation shows some interesting advantages, such as simple control, low THD, high efficiency [15], [16], [18]–[32]. To match the operation of an LCI, an HF dc/dc front stage is needed to create sinusoidal current link. The dc/ac grid-connected converter including an unfolding LCI is shown in Fig. 1. The input current of the LCI shall be shaped as rectified sinusoidal fluctuating current of 120 Hz (i.e., a series of half sine waves) and in phase with grid voltage by the first conversion stage. Instead of a large inductive line filter, only a small inductor with an HF capacitor or a small HF capacitive filter is used to remove the HF harmonics that resulted from last HF conversion stage. The function of inverter is just to unfold the dc-link (half sine wave) current waveform in alternate half-cycles to produce sinusoidal line current. The switch turn-on point is the zero-crossing point of the line voltage so that ZVS turn-on is achieved. The turn-off is near zero voltage, zero current only once at the end of each LF half-cycle. The switches used could be thyristor or gateturn-off transistor. If thyristors are used with natural commutation [15], [18], the output current has higher THD because the turn-off point has to be a little earlier than zero-crossing point to give enough time for the line current to reach zero. Forced commutation can be applied for lower output THD. However, if the switch used is thyristor, extra circuit is required for forced commutation [15]. The disadvantage is the peak power processed has to be twice the average power, which limits its application in high power application. With the improvement in semiconductor technology, new generation of high-speed insulated-gate bipolar transistor (IGBT) with high power rating are available now and the price is decreasing continuously, which make it reasonable to discuss this grid interface for medium to high power applications also.

0885-8993 © 2013 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications standards/publications/rights/index.html for more information.

LI AND BHAT: COMPARISON STUDY OF HIGH-FREQUENCY ISOLATED DC/AC CONVERTER EMPLOYING AN UNFOLDING LCI

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In this paper, the dc/ac grid-connected converter is expected to have HF transformer isolation, soft switching, high efficiency, reduced electromagnetic interference, low output line current THD, and unity line power factor. The converter working in HF has many advantages: fast transient response, small filter, and small-size transformer when isolation is needed. However, HF operation also brings high switching losses. To overcome the drawback, soft-switching techniques are needed for HF switching. Five different dc/ac topologies to realize grid connections through an LCI are analyzed and designed in Section II. The pros and cons of them are then summarized and compared in Section III. It is shown that the dual-LCL converter has advantages in ZVS operation of all switches in the whole load range, low line current THD, and unity power factor. A prototype of the dual-LCL converter is built and tested in the lab to verify its performance [28]–[32]. II. TOPOLOGIES OF DC/AC GRID-CONNECTED CONVERTER INCLUDING AN UNFOLDING LCI There are many HF transformer isolated dc-to-LF ac converters including an LCI reported in the literature. r Flyback or forward type sinusoidally modulated converter [19]–[22]: the power transfer capability of flyback or forward type converters are limited by the power transformer so that they are mainly used for small power applications. r Push–pull type sinusoidally modulated converter [23]: a 4-kW push–pull converter working at 20 kHz with duty cycle modulated was reported in [23]. It is simple and highly efficient with only two switches. The reported circuit works in hard-switching condition, but it can be changed to soft switching by means of the leakage inductance [37] or resonant tank [38]. The disadvantages associated with a push–pull converter include high switch rating, transformer saturation due to asymmetry, and difficulty in obtaining small leakage inductance. r Hysteresis controlled sinusoidally modulated LCC-type series-parallel resonant converter (SPRC) [15], [16]: due to the limited availability of fast gate-turn-off devices at that time, the asymmetric silicon-controlled-rectifiers were used. The converter was designed to operate below resonance and work in zero-current switching (ZCS). The sinusoidal fluctuating dc-link current is realized through hysteresis current control. Although the switching frequency of the HF converter was fixed when it was running, the frequency of the on/off behavior of the converter was varying as the nature of hysteresis control [15], [16]. The performance can be improved if gate-turn-off devices are used. r Variable frequency sinusoidally modulated series resonant converter (SRC) [24]: the variable frequency sinusoidally modulated SRC can achieve ZCS too [24]. It has relative high peak switch current due to below resonance variable frequency control. The converter works in just continuous current mode at full load and enters discontinuous current mode (DCM) at partial load by varying switching

Fig. 2. Topology 1: variable frequency sinusoidally modulated PWM converter [18].

frequency. To deliver a certain amount of current, the DCM operation has to go with higher peak current compared with continuous current mode. r Fixed-frequency sinusoidally modulated SPRC [36], [39]: the fixed frequency LCC-type SPRC can be controlled by modulating the phase shift between the gating signals of two legs. It is known that it cannot maintain ZVS at light load or for wide variations in input voltage. Since the unfolding stage requires the output of SPRC to be controlled (for variations in voltage as well as load) in a 120-Hz cycle, it is not suitable for this application. Because of operational mode or use of thyristor/bipolar junction transistor, all the aforementioned topologies have some kind of disadvantages, for example, low switching frequency and non-ZVS operation. Here, the following five important topologies are chosen to discuss in detail for potential application in renewable generation systems: 1) variable frequency sinusoidally modulated pulse-width modulation (PWM) converter [18]; 2) fixed-frequency sinusoidally modulated LCL-type SRC [41], [42]; 3) fixed-frequency sinusoidally modulated parallel dual SRC [25]–[27]; 4) fixed-frequency sinusoidally modulated series dual SRC [28]; 5) fixed-frequency sinusoidally modulated series dual-LCL SRC [30]–[32]. A. Topology 1: Variable Frequency Sinusoidally Modulated PWM Converter The first topology discussed here is a variable frequency PWM converter proposed for residential photovoltaic application [18], [29]. As shown in Fig. 2, the full-bridge HF inverter

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Fig. 3.


Topology 2: fixed frequency sinusoidally modulated LCL-type SRC.

is adopted for sinusoidal modulation to create the rectified sine dc-link current. To prevent the saturation of the HF transformer, the flux level is monitored through a flux detector. If the flux level reaches the preset flux limit, it will be clamped by resetting the transformer voltage to zero. Due to the existence of the dc-link inductance, the dc-link current will freewheel through the HF rectifier bridge diodes and the grid, and start to decay. Once the current falls to the sinusoidal reference current limit, the input voltage is applied to the primary side of the HF transformer to boost the current. As the current increases, the transformer flux changes direction and increase until it touches the negative flux limit. As this procedure repeats, a 120 Hz rectified sine-wave dc-link current can be built. Every time when the input voltage is reapplied with changing polarity to the primary side of the HF transformer, the current will change direction with the help of the leakage inductance. The leakage inductance is quite small compared with the dc-link inductance and will decide the speed of current direction change. Just before the end of each LF half-cycle, all the HF switches are turned-OFF so that the dc-link current through Lf decays to zero resulting in natural turn-OFF for the conducting thyristors. Note that thyristors Sa−Sd unfold the dc-link series of half sine-wave currents (Sa and Sd are on for one half-cycle and Sb and Sc for the other halfcycle) shown to sinusoidal current in phase with line voltage. Disadvantage of the LCI inverter using thyristors by the aforementioned method is the increased line current distortion due to the time required for the current to decay to zero and a small dead gap required for reliable commutation of SCRs. Replacing thyristors by bipolar transistors improved the performance [18].

Fig. 4. [26].

Topology 3: fixed frequency sinusoidally modulated parallel dual SRC

B. Topology 2: Fixed-Frequency Sinusoidally Modulated LCL-Type SRC

D. Topology 4: Fixed-Frequency Sinusoidally Modulated Series Dual SRC

Although it is hardly found in the literature, full-bridge resonant converter with fixed frequency phase-shift control may also be used to generate sinusoidal output [40]. Among all resonant type dc/dc converters, the LCL-type SRC with capacitive output filter shown in Fig. 3 has many preferable features, such as low component stress, wide load range of soft-switching operation, ZCS of rectifier diodes, etc. [41].

A phased-modulated DSRC proposed by Pitel [28] is capable of producing sinusoidal output of LF. The topology can be viewed as two conventional series resonant converters connected in series on the secondary sides of the HF transformers, as shown in Fig. 5. The power control is realized by controlling the phase shift between the two bridges. When the phase shift varies from 0◦ to 180◦ , the output level between maximum and

The principle of fixed frequency or PWM power control of resonant converter is to adjust the pulse width of the HF inverter output voltage by shifting the gating signals between two legs of the bridge. With the pulse width controlled between 0◦ and 180◦ , the output can vary between maximum and zero. This power control feature is suitable for sinusoidal modulation, which requires zero output to peak output along the 60-Hz half-cycle. C. Topology 3: Fixed-Frequency Sinusoidally Modulated Parallel Dual SRC The combination of two resonant conversion units might be a good solution, in which the high power could be shared by the two units so that the components stress could be reduced consequently. The first such topology is a parallel connected dual SRC with sinusoidally modulated phase shift [25], [26]. It can be seen in Fig. 4 that two half-bridge SRCs are connected in parallel on the primary side of the HF transformer. By fixing the operating frequency of the dual series resonant converter (DSRC) above the tank resonance frequency, the output power control can be obtained by varying the phase shift between the gating instants of two half-bridges. By varying the phase-shift between 0◦ and 180◦ , the output power can be controlled from maximum to zero. It is noted that the input current of the HF transformer is the algebraic sum of the two current vectors from the two different bridges.


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TABLE I PARAMETERS OF THE FIVE DC/LFAC GRID-CONNECTED TOPOLOGIES FOR THE DESIGN EXAMPLE

Fig. 5. [28].

Topology 4: fixed frequency sinusoidally modulated series dual SRC

Fig. 6. Topology 5: fixed frequency sinusoidally modulated series dual LCLtype SRC [30]–[32].

0 can be produced. It is shown that the input voltage of the HF transformer is the algebraic sum of the voltage vectors from the two bridges. The two bridges are always working at a fixed switching frequency with 50% duty cycle. The switching features are similar for all switches in each bridge. The switching frequency can be set at or higher than the LC tank resonance frequency. Because of phase modulation, it can be concluded that the equivalent loads for the two HF inverters are different: one is an inductive load and the other is a capacitive load [28]. Hence,

Fig. 7. Simulation waveforms of Topology 1 at P o = 500 W: (a) (top) output line current, (middle) tank current is , and (bottom) the primary-side transformer voltage V T on LF scale; (b) switch currents of two legs on HF scale near 90 ◦ ; and (c) switch currents of two legs on HF scale near 30◦ .

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Fig. 8. Simulation waveforms of Topology 1 at P o = 250 W: (a) switch currents of two legs on HF scale near 90◦ ; and (b) switch currents of two legs on HF scale near 30◦ .

Fig. 10. Simulation waveforms on HF scale of Topology 2 at P o = 500 W: (a) switch currents of two legs, parallel inductor current, tank capacitor voltage near 90◦ ; and (b) switch currents of two legs, parallel inductor current, tank capacitor voltage near 30◦ .

the two bridges will draw unequal power from the input. The heating due to power loss would be unequal, which makes it hard to design heat sink. Efforts were made to compensate for the unbalancing by adding additional circuit to the inverter bridge with lagging gating signals for a similar topology with LCC tank in [36], which is not suitable for grid connection with wide variation of phase shift. E. Topology 5: Fixed-Frequency Sinusoidally Modulated Series Dual-LCL SRC

Fig. 9. Simulation waveforms of Topology 2 on LF scale at P o = 500 W: output line current, tank current is , parallel inductor current, tank capacitor voltage v c , and diode current of the HF rectifier.

A phased-modulated dual-LCL-type series resonant converter was proposed in [30]–[32], which combined the features of Topologies 2 and 4, as shown in Fig. 6. It can be predicted that the switch currents in the two bridges are not the same as Topology 4 due to the existence of parallel inductors. However, the ZVS operation of all switches is improved with the help of the parallel inductor.


Fig. 11. Simulation waveforms of Topology 3 on LF scale at P o = 500 W: (from top to bottom) line current, tank current, and tank capacitor voltage in the lagging bridge; tank current and tank capacitor voltage in the leading bridge; diode current of the HF rectifier.

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Fig. 12. Simulation waveforms of Topology 3 on HF scale at P o = 500 W: (a) from top to bottom: switch current and capacitor voltage in the lagging bridge, switch current and capacitor voltage in the leading bridge, near 90 ◦ ; and (b) from top to bottom: switch current and capacitor voltage in the lagging bridge, switch current and capacitor voltage in the leading bridge, near 30◦ .

III. PROS AND CONS OF THE FIVE TOPOLOGIES The topologies explained in the last section are then designed for their performance comparison. For fair comparison, the following specifications are used for designing all the topologies: input dc voltage = 200 V, output single-phase line rms voltage = 208 V, and output average power = 500 W. Based on the design procedure summarized in Appendix A, the calculated design parameters for the five converters discussed in Section II are summarized in Table I. With the converters designed from the specifications, digital simulations of those five converters connected to the grid with close-loop control in one 60-Hz period are done. The simulation plots of the five topologies for both line-frequency scale and HF scale are shown in Figs. 7–15. The simulation results are summarized in Tables II and III. The operation parameters at the zero phase shift and peak output are given in Table II. Table III illustrates the variation of components with respect to the phase shift. Based on the comparisons, the pros and cons of the five dc/ac converters are shown in Table IV. The details of design

procedure (summary given in Appendix A), more simulation waveforms and results can be found in [30]. Due to the inductive dc-link filter, the tank current in Topology 1 is nearly square wave as shown in Fig. 7(b) and (c), which gives moderately low peak current in switches and rectifier diodes. The peak tank current decreases with the instantaneous value of line voltage as shown in Fig. 7(a), which indicates high efficiency at partial output. Thus, in terms of component stresses, Topology 1 is quite good among all candidates. With the help of external tank inductance including the leakage inductance of the HF transformer, switches in both legs can work in ZVS for any load level as shown in Fig. 7(b) and (c) for full load and Fig. 8 for half-load. However, the tank inductance also results in duty-cycle loss of the HF inverter. Additionally, the nearly square-wave current at diode rectifier could induce voltage ringing of diodes at commutation because of the resonance of diode capacitance with the leakage inductance of transformer, which needs extra snubbers to clamp the overshoot

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Fig. 13. Simulation waveforms of Topology 4 on LF scale at P o = 500 W: (from top to bottom) line current, tank current is 1 and tank capacitor voltage v c 1 in the lagging bridge; tank current is 2 and tank capacitor voltage v c 2 in the leading bridge; diode current iD a of the HF rectifier.

voltage across the diodes [42]. The nature of variable switch frequency control brings difficulties to the design of HF transformer, reactive components, and control circuit. In Fig. 9,the envelopes of important parameters of Topology 2 at full load are shown in a 60-Hz period. With a capacitive filter, the diode bridge ringing does not exist in Topology 2. The commutation behaviors of switches in the two bridge legs are different when phase shift is not zero as shown in Fig. 10. Switches in the leg with leading gating signal can always maintain ZVS regardless of the phase shift. However, the switches in the lagging leg may lose ZVS at large phase shift at low load. A small parallel inductor is helpful to extend the ZVS at the cost of large parallel inductor current. Nearly sinusoidal voltage and currents means low dv/dt, di/dt and simplifies filter design. However, the switch peak and rms currents are larger than those of Topology 1 at peak output point due to resonance. In Fig. 11, the envelopes of important parameters of Topology 3 at full load are shown in a 60-Hz period. The peak tank current at peak output is the highest in all topologies. As shown in Fig. 11, the tank inductor current and resonant capacitor voltage can reach about 9.24 A and 199 V, respectively, at the nearly zero phase shift or peak output. Since it is implemented in half-bridge type, those values are expected. The envelopes shown in Fig. 11 also indicate that the peak tank currents and peak capacitor voltages increase as the output voltage declines, which results in high power loss and low efficiency. At zero output, the switch current and capacitor voltage stress reach maximum. The advantage of Topology 3 is full range

Fig. 14. Simulation waveforms of Topology 4 on HF scale at P o = 500 W: (a) switch current is w 1 of the lagging bridge, switch current is w 2 of the leading bridge, capacitor voltages v c 1 and v c 2 near 90◦ ; and (b) switch current is w 1 of the lagging bridge, switch current is w 2 of the leading bridge, capacitor voltages v c 1 and v c 2 near 30◦ .

ZVS operation for all switches in both half-bridges due to fixed frequency control as shown in Fig. 12. Topologies 4 and 5 use more components than any other topologies, which appears to introduce more power loss factors. However, redundancy of the structure could be validated by some benefits which are the key to high power capability. According to simulation results in Figs. 13–15, Topologies 4 and 5 have smaller component stresses over the whole output range than those of single-bridge topologies at the price of two sets of HF inverters. It can be concluded that with same components the power handling capacity of those topologies will double while distributing power losses. Another preferable feature is the high voltage gain. The conventional bridge converters are all buck-type with the primary-side reflected voltage gain no more than one. By means of the series connection of two SRCs, the primary-side reflected voltage gain in Topologies 4 and 5 is improved as high as two in theory, which will not require a high transformer turns ratio when connecting with a high-voltage


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TABLE III CHANGES OF WORKING CONDITIONS FROM PEAK OUTPUT TO ZERO OUTPUT POINT OF THE FIVE TOPOLOGIES FOR THE DESIGN EXAMPLE

TABLE IV GENERAL PERFORMANCE COMPARISONS OF THE FIVE TOPOLOGIES FOR THE DESIGN EXAMPLE

Fig. 15. Simulation waveforms of Topology 5 on LF scale at P o = 500 W. Waveforms from top to bottom are: output voltage and current v o , io ; dc-link current id c ; rectifier output current i3 ; tank current in the leading bridge is 1 ; tank current in the lagging bridge is 2 ; and applied phase-shift angle 2θ.

TABLE II WORKING CONDITIONS NEAR PEAK OUTPUT POINT (90◦ ) FOR THE DESIGN EXAMPLES, LINE VOLTAGE (RMS) = 208 V

grid. The efficiency at partial output is still good as the switch current decreases with the output current. For Topology 4, there are two drawbacks needing improvement. The first one is high resonant capacitor peak voltage in SRC. The second is that only switches in the leading bridge can work in ZVS as shown in Fig. 14. The switches in the bridge with lagging gating signals are always working in ZCS at nonzero phase shift. Topology 5 inherits all advantages of Topology 4, i.e., low component stress and high-voltage gain. With LCL-type tank, the resonant capacitor voltage is found to be smaller than that in regular SRC as shown in Fig. 15. As expected, the ZVS operations of all switches are maintained at all phase shift, even at the maximum phase shift. At the same time, the parallel inductor current is kept no more than 1.2 A. In Table V, the estimated breakdowns of the five converters at rated output power are presented. For fair comparison, the component parameters (such as on-state voltage drop of switches and diodes, etc.) are assumed to be the same. In terms of

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TABLE V ESTIMATED LOSS BREAKDOWNS OF THE FIVE DC/LFAC GRID-CONNECTED TOPOLOGIES FOR THE DESIGN EXAMPLE

efficiency, Topology 2 is the best and Topology 3 is the worst, while the other three have similar performance. In conclusion, Topology 3 is eliminated first due to poor partial output efficiency and high switch current rating. Topology 1 is then removed from the list due to variable frequency control, duty cycle loss, and voltage ringing of diode rectifier. Except for switch ratings, Topology 2 shows satisfying performance compared with Topologies 4 and 5. Thus, Topology 2 is recommended for low power application of a few kilowatts. However, it may not be suitable for high current power applications (e.g., up to about 10 kW) due to the possible loss of ZVS in one switch leg resulting in high power loss. Topologies 4 and 5 achieve lowest switch rating at the price of two bridges. However, the dual-bridge structure enables the potential use of Topologies 4 or 5 in higher power applications. The total voltage-ampere (VA) rating of two transformers in Topologies 4 or 5 is only slightly higher than that of Topology 2. Since Topology 5 can work in ZVS for both bridges, it is worth doing further work to validate its performance. In the lab, a 500-W prototype of Topology 5 was built to verify its performance [30]–[32]. The switch used in the HF inverter stage is warp speed IGBT—G4PC40UD and the switching frequency is chosen at 100 kHz. HF rectifier diodes used were MUR1560, and IGBTs used for the LCI stage were IXGH10N100 A. The HF transformers were built using ferrite HF magnetic cores (core material PC40) ETD49-Z manufactured by TDK Co. The type of inductor cores is A-2541682 using molypermalloy powder (MPP) material. All magnetic cores were wound using Litz wire to reduce the copper losses. Fig. 16 shows the grid voltage and currents for full-load and half-load. It is seen that they are in phase which gives nearly unity power factor from the spectrum of line currents; the THD of line current is found to be 9.2% and 9.49% which are higher than 5% standard. The reason is the distortion in the line voltage in our lab which is used as control reference. The envelope of tank currents and capacitor voltages in two bridges, diode rectifier input voltage, and current in one 60-Hz period are given in Fig. 17. The efficiency at full-load and half-load is 90% and 83%, respectively. IV. CONCLUSION In this paper, a review of the HF link dc/dc converter followed by a UFI for connecting alternate energy sources is

Fig. 16. Experimental results of dc/ac converter with close-loop control interfaced with 208 V single-phase grid. (a) P o = 500 W, line voltage (100 V/div) and current (3.5 A/div); FFT spectrum of line current in amplitude (400 mA/div); and (b) P o = 250 W, line voltage (100 V/div) and current (3.5 A/div); FFT spectrum of line current in amplitude (200 mA/div).

Fig. 17. Experimental results of dc/ac converter with close-loop control interfaced with 208 V single-phase grid delivering P o = 500 W (a) tank current is 1 (top, 4 A/div) and capacitor voltage v c 1 (bottom, 100 V/div) in leading bridge; (b) tank current is 2 (top, 4 A/div) and capacitor voltage v c 2 (bottom, 100 V/div) in lagging bridge; and (c) input voltage v r e c −i n (top, 300 V/div) and current i2 (bottom, 8 A/div) of diode rectifier.

presented. This scheme is found to be simple, highly efficient, and easy to connect with utility line and has low line current THD. The key point of this scheme is to produce a rectified sinusoidal fluctuating current of 120 Hz (i.e., a series of half sine waves) and in phase with grid voltage by the HF dc/dc converter first conversion stage. Five HF dc/LFAC converters are discussed and compared through design and simulation. In conclusion, the fixed-frequency sinusoidally modulated LCL-type SRC shows the best overall performance in terms of efficiency for specified test condition and is recommended for applications of a few kilowatts. When the higher power level is expected, a fixed-frequency sinusoidal-modulated series dual-LCL SRC with wide ZVS range shows a good potential for various


alternate energy applications: larger power processing capacity, full range ZVS, low components stress, nearly unity PF, and low THD in line current. Experimental results are included for validation purpose too. The experiment of a 500-W prototype intended to validate the performance of the converter only. However, due to the features of this topology it should not be difficult to scale-up the design for applications requiring few kilowatts. The THD than required issue might be improved by generating an ideal sine reference signal synchronized with the grid. APPENDIX A A. Converter Designs All the converters designed have the following common specifications: Peak output power Po = 1 kW, Vin = 200 V, and Vo = 295 V. Topology 1: The key point to the design of this converter is the choice of the switching frequency range and maximum duty cycle. The minimum switching frequency is set as 10 kHz, which decides the size of the HF transformer. The maximum duty cycle is chosen as Dm ax = 0.9. The maximum on-time with lowest switching frequency happens at the peak output voltage to deliver the peak output power and is ton =

Dm ax = 45 μs. 2fm in

(A1.1)

The resonant tank values are calculated as follows [40]: M VB2 JF = 28.86 μH 2πfs Po

(A2.3)

Po F = 106.22 nF 2πfs M VB2 J

(A2.4)

Ls =

Cs =

Topology 3: The chosen parameters are: Po = 1 kW, fs = 100 kHz, F =fs /fr = 1.1, Vin = 100 V (half-bridge), Vo = 295 V, and Q = 2πfr Ls /R o = 1, where all the parameters have been defined earlier and R o is the equivalent load resistance at peak output. The converter voltage gain can be evaluated as follows [43]:

M= π2

8

The primary-side reflected output voltage is as follows: Vo = Vin M = 97.34 V.

Ro = Vo 2 / (0.5 Po ) = 18.95 Ω.

(Vin nt − Vo )ton = 5.66 mH. 2Δio

(A1.2)

(A1.3)

(A1.4)

The tank inductance, which includes the leakage inductance of the transformer, is calculated as follows [18]: Ls =

(1 − Dm ax )Vin Vo = 90 μH. 4Po fs

(A1.5)

Topology 2: The chosen parameters are [40]: fs = 100 kHz, F = fs /fr = 1.1, converter gain M = 0.965, J = 0.427, where F is the normalized switching frequency, fs is the switch frequency, fr is the resonance frequency of the tank, J is the normalized primary-side reflected output current with base current IB = 2πfr Ls /Vin , and M is the voltage gain. The primary-side reflected output voltage is as follows: Vo = Vin × M = 193 V.

(A2.1)

The HF transformer ratio is as follows: 1 : nt = Vo : Vo = 1 : 1.53.

(A3.3)

The resonant tank values are given as follows:

The maximum link current ripple is defined to be less than 5%, i.e., Δio = (Po /Vo )(5%) = 0.147 A, then the filter inductance can be obtained as follows: Lf =

(A3.2)

The design of the dual SRC in parallel is same as the design of a single SRC with same output voltage, half output current and half output power. The reflected equivalent resistance of half output power is as follows:

(A2.2)

QF Ro = 33.175 μH 2πfs

(A3.4)

F = 92.385 nF 2πfs QRo

(A3.5)

Ls =

The HF transformer turns ratio is obtained by 1 : nt = V0 : V0 = 1 : 1.63.

= 0.9734. (A3.1)

[(Q/F )(F 2 − 1)]2 + (8/π 2 )2

The primary-side reflected peak output voltage is as follows: V0 = (Vin ) (Dm ax ) = 180 V.

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Cs =

The HF transformer ratio is obtained as follows: 1 : nt = Vo : Vo = 1 : 3.03.

(A3.6)

Topology 4: The chosen parameters are: fs = 100 kHz, F = fs /fr = 1, and Q = 2πfr Ls /Ro = 1. If the circuit is exited at resonance frequency, the converter gain of a single SRC at zero phase-shift would be unity: M = 1. The primary-side reflected output voltage is same as input voltage, Vo = Vin · M = 200 V. The design of the dual SRC in series is the same as the design of a single SRC (given for Topology 3) with same output current, half peak output voltage and half peak output power. Using (A3.3), the reflected equivalent resistance of half peak output power is: Ro = 80 Ω. Using (A3.4) and (A3.5), the resonant tank values are: Ls = 127.3 μH and Cs = 19.89 nF. The HF transformer ratio is: 1: nt = (2 V o = 1:0.74. Topology 5: The approximate optimum operating point [30]–[32] is chosen as: M = 1.94, J = 0.3185, F = 1.1, and K=Lp /Ls = 10. Using (A2.3) and (A2.4) the values of tank components are calculated as: Ls = 43.28 μH and Cs = 70.81 nF. The primary-side reflected output voltage is V o = Vs ×M = 200 × 1.94 = 388 V. The HF transformer turns ratio is

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1: nt = Vo : Vo = 1: 0.76. With K = 10, parallel inductance is Lp = K × Ls = 432.3 μH.

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Xiaodong Li (S’02–M’09–SM’12) received the B.Eng. degree in electrical engineering from Shanghai Jiao Tong University, China, in 1994, and the M.A.Sc. and Ph.D. degrees in electrical engineering from the University of Victoria, BC, Canada, in 2004 and 2009, respectively. From 1994 to 2002, he was with the HongWan Diesel Power Co., Zhuhai, China, as an Electrical Engineer where he conducted maintenance of diesel power generation system. He joined the Faculty of Information Technology, Macau University of Science and Technology, Macau SAR, China, in 2009, where he is an Assistant Professor. His research interests include high-frequency power converter, and fault diagnosis of motor drive. Dr. Li is the recipient of the IEEE PES Best Paper Prize in 2007.

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Ashoka K. S. Bhat (S’82–M’85–SM’87–F’98) received the B.Sc. degree in physics and math from Mysore University, Karnataka, India, in 1972, the B.E. degree in electrical technology and electronics and the M.E. degree in electrical engineering, both with distinction from the Indian Institute of Science, Bangalore, India, in 1975 and 1977, respectively. He also received the M.A.Sc. and Ph.D. degrees in electrical engineering from the University of Toronto, ON, Canada, in 1982 and 1985, respectively. From 1977 to 1981, he was with the Power Electronics Group, National Aeronautical Laboratory, Bangalore, India, as a Scientist, and was responsible for the completion of a number of research and development projects. After working as a Postdoctoral Fellow for a short time, he joined the Department of Electrical and Computer Engineering, University of Victoria, BC, Canada, in 1985, where he is currently a Professor of electrical engineering and is engaged in teaching and conducting research in the ar+ea of power electronics. He was responsible for the development of the Electromechanical Energy Conversion and Power Electronics courses and laboratories in the Department of Electrical Engineering at the University of Victoria. Dr. Bhat received the “Excellence in Teaching Award” from the Faculty of Engineering during the year 2008 and the “Wighton Fellowship” for the year 2010. He is a Fellow of the Institution of Electronics and Telecommunication Engineers, India, and a registered Professional Engineer in the province of British Columbia, Canada.