A Double ZVS-PWM Active-Clamping Forward Converter ... - Ivo Barbi

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Forward Converter, which operates without switching losses from no-load ..... 10 [AI. Fig. 11. Output characteristic of the converter. p 1.: 1.8. 1.7. 1.6. I .5. 1.4. 1.3.

A DOUBLE ZVS-PWM ACTIVE-CLAMPING FORWARD CONVERTER

RenC Tonico BascopC

Ivo Barbi

and

FEDERAL UNIVERSITY OF SANTA CATARINA Department of Electrical Engineering Power Electronic Institute P. 0. Box. 51 19 88.040-970 - Floriandpolis - SC - Brazil Tel.: (55)48-331.9204 - Fax: (55) 48-234.5422 E-mail: [email protected]; [email protected] Abstract - This paper presents a new isolated dc-dc converter, named Double ZVS-PWM Active-Clamping Forward Converter, which operates without switching losses from no-load up to hll-load. The proposed converter is based on two active-clamping forward converters, coupled with a single high frequency transformer. This converter is suitable for high input voltage and high power applications. Operation principle, theoretical analysis, design example and experimental results, taken from a 3kW laboratory prototype, are presented.

hand, they are capable of automatically balance the voltage of the input capacitors associated in series and connected to a dc bus voltage. If one of the voltages is higher, the respective capacitor will transfer energy to the other through the transformer during the on time. In this paper we propose the converter shown in Fig. 1. The converter is composed of two active-clamping ZVSPWM forward converters coupled by single transformer. The proposed converter was derived from the topologies proposed in references [5, 61.This converter is described and analyzed in the following sections.

I. INTRODUCTION

Nowadays, the most used power converter topology in high power dc-dc power supplies applications is the fullbridge zero-voltage-switching pulse-width-modulation (FBZVS-PWM) converter. It is considered one of the best alternative for high power applications and its control strategy is well known. This converter possesses the most desirable characteristic of both, the hard switching PWM and the soft switching resonant converters, avoiding their major drawbacks, such as commutation losses in the first mode and variable switching frequency and high conduction losses in the second mode[ 11. However, the conventional FB-ZVS-PWM converter is not suitable for high input voltage applications because the total input voltage is applied across its blocking switches. Among the alternatives to overcome this drawbacks are the series connection of switches and multilevel topologies. In the series connection of the switches, the static and dynamic sharing of the voltage across the switches is difficult to obtain and requires specific techniques. Multilevel topologies seems to be a more effective solution because they can solve the problem of static and dynamic sharing of the voltage and minimize the electromagnetic interference, since the dv/dt is reduced [2]. Another alternative to solve the problems of the series connection of the switches is the association of two or more converters in series reducing the voltage stress on each switching device. This method is appropriate if a perfect division of the voltage between the converters can be guaranteed at any time. This condition is attainable by coupling the converters with a single high-frequency power transformer as explained in [3, 41. The associated converters coupled by a single transformer are controlled by the same control circuit, in this way, the pulse-width-modulator and drive circuit are identical for all the converters, allowing a good sharing of the voltage across the switches. On the other 0-7803-5160-6/99/$10.00 0 1999 IEEE.

I

I

Fig. 1. The proposed converter.

11. CIRCUIT DESCRIPTION AND

PRINCIPLE OF OPERATION A. Circuit Description

The proposed converter is shown in Fig. 1. It is composed of the following components: main switches SI and S2, auxiliary switches S3 and S4, resonant inductances L 1 and Lr2,high frequency transformer, rectifier diodes D,, and Dn, output filter Lo and CO,input voltage sources 2Vi, clamping capacitors Ccl and Cc2,commutation capacitors C,, and CI?, anti-parallel diodes Dl, DZ,D3 and Dd, and load R,,. B. Principle of Operation

The following assumptions are made to simplify the analysis: 0 the circuit operates in steady-state; a all components are considered ideal; 596

the output inductor filter Lo is large enough to be considered as a current source with a value equal to the load current I,; the voltage over input capacitors are equal (VI=V2=Vi); the resonant inductors are equal (LrI=LrZ=Lr); the clamping capacitors are equal (Ccl=Cc2=Cc); the resonant capacitors are equal (Crl=CrZ=Cr); 0 the transformer leakage inductance is absorbed into Ll and Lrz; 0 there is no difference among the gate signals applied to each pair the of switches (SI,S2and S3,S4),they have identical duty cycles. Fig. 2 shows the topological stages of the converter for a half-period and Fig. 3 shows the main theoretical waveforms for one switching period. The six sequential circuit states are described below: First Stage (to, tl): During this stage, the power is transferred to the load in two ways. Power is transferred from the input source VI through switch S1 and also, there is a power transfer to the load when capacitor Cc2 discharges through switch S4. The switches S2 and S3 are in “off’ state and the voltage over them is equal to the voltage over the clamping capacitors C,I and C,Z. Second Stage (tl, t2): At instant tl, switch SI is tuned-off with zero voltage. Capacitor Crlbegins to charge linearly with a constant current. This stage finishes when the voltage over SI is VI, and over S3is Vc-VI. Third Stage (tz, t3): At instant t2, the transformer voltage is zero, in such a way that, the capacitor C,,begins the resonance with LrI. Also, the output current I, starts the freewheeling process. When the voltage over Sl reaches V, this stage finishes. Fourth Stage (4, t4): At instant t3, diode D3 is direct biased and starts conducting the current through Lrl, which decreases linearly. In the same manner, the current through Lr2 decreases linearly. During this stage the load current keeps freewheeling through diodes DrI and Dn. Switch S3 must be gated on before diode D3 is reverse biased. This stage finishes when the switch S4 is turned OK Fifth Stage (t4, t5): When switch S4 is turned off, the current through L2is deviated to resonant capacitor Cr2.This capacitor is discharged in a resonant way. During this stage the load current keeps freewheeling. This stage finishes when voltage over capacitor Cr2 is null and the diode D2 starts to conduct. Sixth Stage (t5, ts): At the instant t5, diode D2 is direct biased and begins conducting the Lr2 resonant inductor current. During this condition, the switch S2 must be gated to conduct. The current through Lr2 decreases linearly until it reaches zero. Thereafter, this current changes direction and flow through switch S2. When the sum of the currents in LrI and Lr2 is equal to -nI, this stage finishes. During this stage there is no power transferred to the load; therefore, only a reduction in the duty cycle occurs.

597

b

LII

-

-

a ) First Stage

,_I

VI

v2

-

b ) Second Stage L.1

c ) - Third Stage

d

- Fourth Stage

U2

W y2

la

DI2

1”lrLlrr

ER

-

f ) Sixth Stage Fig. 2. States of operation of the converter.

n : transformer turns ratio (NJ"); At, : no power transfer time. The no power transfer time in half period is: At, =

n.1, .L, 42-0)

(3 1

v, The reduction of duty cycle in the period T, is given by (4)

AD=- 2.A1, Ts

Substituting(3) into (4), we obtain AD =

2 . f , .L, . n . l , .(2-0)

(5)

v,

Substituting (1) and (3) into (2), the average output voltage - is obtained as 2.f, .L, . n . l , (6) v, = n . v i D

.[-

1

VI

C. Commutation Analysis

This converter presents soft turn-on and turn-off commutation during its operation. Turn-on : to analyze this commutation consider the fifth stage (the main switch S2tums-on). Equation (7) is obtained for no load (critical) operation condition of the converter. Therefore,

b

,I

a

U

11

IS

)6

Fig. 3. Main waveforms of the converter.

111. THEORETICAL ANALYSIS

Qon

A. Clamping Characteristic The clamping voltage and input voltage ratio (VJVJ can be obtained determining the average voltage over the main switch SI and then, applying loop voltage equations. So,

The clamping characteristic is shown in Fig. 12.

B. Output Characteristics

= ~o eton

where: i:switching ti-equency and resonance ti-equency ratio 9 on : displacement angle of the oscillation fiequency ton:discharge time of the resonant capacitor, Displacement angle e on as function of during the turnon is shown in Fig. 4. The discharge time of the resonant capacitor Cr2is given by 8,".i ton = -

r

2.n.f,

In the sixth stage, a reduction in the duty ratio occurs because S2 is gated on but output freewheeling maintains zero voltage across +e transformer. According to the waveforms shown in Fig. 3, and considering that the commutation time is much smaller than the switching period, the average output voltage is given by

where: Vo : output voltage; D: duty cycle; Vc : clamping voltage; Ts : switching period;

eon

'I

Iradl

>:

0

O B

om

"a

0 0

(I,

0,:

at,

01s

Fig. 4. Displacement angle e on as function of during tum-on.

-

f

598

011

-

0:

Turn off: In this case, it is analyzed the turn-off of the switch S I which happens in the second and third stages. During this switching, the resonant capacitor Crl is charged. The equation (1 1) is also obtained for no load operation condition of the converter. Therefore,

eoff = @ , * t o f f (12) Displacement angle eonas function of T during the turnoff is shown in Fig. 5. @a

Id1

1) Resonant inductor L,: The resonant inductor is defined by the specified maximum reduction of duty cycle and is calculated from (5). L, =

*AD,, 200 .0.2 = 5.6pH 2 . f. .n .I, .(2 - 0-) = 2.1. lo5 .0.6.50.(2 - 0.8)

2) Clamping capacitor CC:To determine the capacitance it is consider that the resonance period of the clamping capacitor and resonance inductor is three times the switching period Ts. Therefore, 2.x . = 3 . T,

24

3) Resonant capacitor C,: To determine the capacitance it is necessary to find the oscillation frequency between L, and C,. For this purpose, Fig 5 and equation (13) - are used. From the curve of Dmi,=0.65, values of [email protected] f are: ,8 = 0.56rad

22 2 ,I 1.6

1.4

1.1 I

f = 0.1

0..

From equation (13), the commutation time is:

06

L . 02 0

o

om

am

am

an

0.12

a14

016

at,

The oscillation frequency is obtained from equation (12).

-0 2 t

Fig. 5. Displacement angle , 8 as

function of

7 during turn-off.

The charge time of the resonant capacitor Crlis given by

Iv. SIMPLIFIED DESIGN EXAMPLE A methodology and design procedure is presented in this section.

5.6

CO =

output power; input voltage; output voltage; output current; switching frequency.

V0=60V 10=50A f,=1 OOkHZ

B. Determination of passive components Assuming ideal switches and diodes .and considering: maximum duty cycle Dm, =0.8 maximum duty cycle reduction AD- =0.2 The transformer turns ratio is calculated from (6)

- (2-D")

.%-

(2-0.8) 60 - o.6 (0.8-0.2)200 The clamping voltage is calculated from (1) 2 .Vi = -. 200=333.34\/ V, = (2- 0.8) (2- Dmx) n-NS

N,

(Dmx-ADw)

V,

1

lo6

= 4.6nF

(2. x a1 IO6)'

4) Output filter: The output filter can be calculated in the same way as for a conventional hll-bridge converter. This filter is designed for a maximum current ripple AIh=5A (1 0% of I,) and a maximum voltage ripple AV,=0.48 (0.8% of Vo).

A. Input Data

P0=3000w 2Vi=400V

1

c, =

A',

=

5

P.X.~,.AV, 2 . x . 1 . 1 0 ~ .0.48

= 17pF

Maximum allowable series resistance of output capacitor

COmust be: AVO

- 0,48

RSE = -- -= 0,096Q 5

AkO

C. Semiconductors voltage and current stresses

1) Main and auxiliary switches: the maximum voltage across the blocking switches is: vs,szs,,

=

2.V = -= 2.200 (2-D-) (2-0.8)

333.34u

The rms current through the main switches (SI, Sz) and auxiliary switches (S3, S4) can be calculated by means of the normalized current values determined in [7]; for AD,,=0.2 we have: Isr.sz,

= 0.04

1s3.s4m,

= 0.015

Hence,

599

2) Output rectifier diodes: in the output rectifier, the diode reverse voltage is:

v,.

2.200 2.200 = 0,6.= 200.0V (2- 0.8)

=n.

(2 - D,,)

The interaction of the transformer leakage inductance with the rectifier capacitance during the reverse-recovery processes causes overshoots of the diode reverse voltage. The overshoots can be controlled using soft-recovery rectifiers and a clamping circuit such as that shown in Fig. 6. The diode's average current is given by lDrlD,Z,vp = '0 =

2

In the same way, to determine the external resonant capacitor the output intrinsic MOSFET's capacitances C,, are subtracted; C,, are of 870pF and 980pF. In this section experimental waveforms obtained for an output power of P0=2780W and P0=90W are presented. The waveforms obtained for P0=2780W are shown in Figs. 7 and 8. On the other hand, the waveforms obtained for P0=90W are shown in Figs. 9 and 10. These results c o n f m the soft commutation features of the converter. The output characteristic, the clamping characteristic and the esciency of the converter are shown in Figs. 11, 12 and 13 respectively. They confirm the theoretically predicted results.

50 = 25.OA 2

V. EXPERIMENTAL RESULTS To verify the practical aspects of the proposed converter, a prototype was built with the following components: APT5012LNR - 500V, 42A, 0.1252 MOSFET Si,S2 JRFP460 - 500V, 20A, 0.27R MOSFET s3, s 4 HFA5OPA60C - 600V, 50A diode Drl, Dr2 6pF/400V - polypropylene capacitor CCI,Cc2 3300pF, 350V - electrolytic capacitor C1,C2 2.7nF/1.6kV - polypropylene capacitor Crl, C,, Ferrite core EE75/50 - IP12; Trl NpI=Nm=lO turns, N s ~ = N s ~6=turns 21pH - Ferrite core EE65/26 IP12; LO Nh=12 turns 3.9pH Ferrite core EE42/15 IP12; L,1, Lr2 NLrl=NLR=7 turns. Cwl, C m 0.1 pF, 630V polypropylene capacitor Rb1, Rm2 33kW 5W - resistor DDrl,DDr2 MUR440 - diode

Ref2

20 O m V

200~5

Fig. 7. Voltage and current in switch SI. (10Ndiv.; 100V/div.; 2uddiv)

-

-

-

Ref2

20 OmV

2 Oops

Fig. 8. Voltage and current in switch S4. ( 1 ONdiv.;100Vldiv.; 2usldiv.)

1VI

Fig. 6. Circuit diagram of the laboratory prototype. Ref2

For inductor design, the leakage transformer inductance is subtracted of the resonant inductance calculated. In the prototype, the leakage inductance is around 1.7pH.

600

i o omv

2 oovs

L,,

,

Fig. 9. Voltage and current in switch SI. 2Ndiv.; lOOV/div.; 2uddiv.)

VI. CONCLUSIONS An new active-clamping ZVS-PWM dc-dc converter based in two single forward-converters was proposed in the paper. Analysis of the clamping characteristic, output characteristic, commutation characteristic and design methodology are presented. Experimental results show that the converter presents the following features: 0 Soft commutation of switches SI, S1, S3, S4 for any load current. The power flow from the input to the output is controlled with only two switches, SI and Sz. In this way, conduction losses of the converter are low. Besides, the current flowing through the auxiliary switches SS and S4 is very low. 0 Overshoots voltages across the switches caused by the transformer leakage inductance are reduced by the use of active clamping circuits. 0 Good sharing of the voltage across the switches is observed.

Fig. 10. Voltage and current in switch S4. (2Ndiv.; lOOV/div.;2us/div.)

a4 72

a 40

36 24

2

IZ0 0

20

I0

Fig. 1 1 . Output characteristicof the converter.

p

J. A. SabatB, V. Vlatkovic, R. B. Ridley, F. C. Lee, and B. H. Cho, “Design Considerations for High-Voltage High-Power Full-Bridge Zero-Voltage-Switching PWM Converter”, in IEEE Applied Power Electronics Conference (APEC) Rec., 1990, pp. 275-284. E. Deschamps and I. Barbi, ”A New DC-to-DC ZVS PWM Converter for High Input Voltage Applications”, in IEEE Power Electronics Specialists’ Conference (PESC) Rec., 1998, pp. 967-972. M. Miller, A. Buffh and U. Carlsson, “High Frequency ZVS for High Power Rectifiers”, in International Telecommunication Energy Conference (MTELEC) Proc., 1993, pp. 424-430. N. Kutkut, G. Luckjitt and D. Divan, “A Dual Bridge High Current DC-to-DC Converter with Soft Switching Capability”, in IEEE Industry Applications Society (IAS) Conf. Rec., 1997, pp. 1398-1405. [SI B. Carsten, “Design Techniques for Transformers Active Reset Circuits at High Frequencies and Power Levels”, in High Frequency Power Conversion (HFPC) conf. proc., 1990, pp. 235-246. [6] C. M. C. Duarte, and I. Barbi, “A Family of ZVS-PWM Active-Clamping DC-to-DC Converters: Synthesis, Analysis, Design, and Experimentation”, in IEEE Transactions on Circuits and Systems, august 1997, ~01.44,NO.8. pp. 698-704. (71 R. Torrico Bascope, “Isolated ZVS-PWM dc-dc Converters with Two Input, Single Transformer and Symmetrical Rectifier”, Doctoral Thesis in Preparation, Power Electronic Institute-Federal University of Santa Catarina, Brazil, 1998.

so

40

30

10 [AI

1.: 1.8

1.7 1.6

I .5 1.4 1.3 1.2

1.1

I 0

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

I

0.9

D Fig. 12. Clamping characteristicof the converter.

’* I%] 95 w 85

75 70

e5 09

55 I

0

I

12W

I

18MI

I

24au

Fig. 13. Measured efficiency of the converter.

REFERENCES

I 3wo

Po WI

601

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