A Dual-Band Parallel Doherty Power Amplifier for ... - IEEE Xplore

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IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 60, NO. 10, OCTOBER 2012

A Dual-Band Parallel Doherty Power Amplifier for Wireless Applications Andrei Grebennikov, Senior Member, IEEE, and James Wong, Member, IEEE

Abstract—In this paper, a novel dual-band transmission-line parallel Doherty amplifier architecture for active antenna arrays and base-station applications in next-generation communication systems is presented. The carrier and peaking amplifiers using GaN HEMT Cree CGH40010P devices are designed based on the reactance compensation technique to provide optimum Class-E impedance seen by the device output at the fundamental frequency across the wide frequency range achieving drain efficiencies over 73% across the frequency range from 1.7 to 2.7 GHz. In a single-carrier WCDMA operation mode with a peak-to-average ratio of 6.5 dB, high drain efficiencies of 40%–45% can be achieved at an average output power of 39 dBm with an of about 30 dBc at center bandwidth frequencies of 2.14 and 2.655 GHz. Index Terms—Broadband Class E, Doherty amplifier, efficiency, GaN HEMT, reactance compensation, RF power amplifier, transmission line.

I. INTRODUCTION

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N NEXT-GENERATION fourth-generation (4G)/fifth-generation (5G) telecommunication systems, it is required that the radio transmitter in general and power amplifiers as its key part in particular operate with high efficiency over a wide frequency range to provide multiband and multistandard operation. Besides, in these systems with increased bandwidth and high data rate using an orthogonal frequency-division multiplexing (OFDM) transmission mode, the transmitting signal is characterized by high peak-to-average power ratios due to wide and rapid variations of the instantaneous transmitting power. Therefore, it is very important to provide high efficiency at maximum output power and at lower power levels typically ranging from 6-dB backoff and less over a wide frequency bandwidth. Different 3GPP long-term evolution (LTE) advanced bands for 4G/5G systems with up to 40-MHz channel bandwidths are expected to be covered: tri-band (SMH, CLR, GSM) 0.7–0.9 GHz, tri-band (DCS, PCS, IMT) 1.8–2.1 GHz, dual-band (IMT and IMT-E) 2.1–2.6 GHz, or even multiband 1.8–2.6 GHz. By using GaN HEMT technology and innovative Doherty architectures, average efficiencies of 50%–60% for output powers ranging from 5 to 50 W can be achieved that significantly reduces cost, size, and power consumption of the transmitters. Moreover, power-amplifier miniaturization Manuscript received June 26, 2012; revised July 07, 2012; accepted July 17, 2012. Date of publication August 16, 2012; date of current version September 27, 2012. A. Grebennikov is with Bell Laboratories, Alcatel–Lucent, Dublin 15, Ireland (e-mail: [email protected]). J. Wong is with Alcatel–Lucent Telecom, Swindon, SN5 7YT, U.K. (e-mail: [email protected]). Digital Object Identifier 10.1109/TMTT.2012.2210906

Fig. 1. Block diagram of conventional and modified Doherty amplifiers.

and integration for small-cell applications are vitally needed keeping the same high-performance capability. For a conventional Doherty amplifier with a quarter-wave impedance transformer and a quarter-wave output combiner, the measured power-added efficiency (PAE) of 31% at backoff power levels of 6–7 dB from the saturated output power of about 43 dBm was achieved across the frequency range of 1.5–2.14 GHz [1]. To improve the broadband performance of a conventional Doherty amplifier, an output network can be composed of two quarter-wave impedance inverters with reduced impedance transformation ratios [2]. For broadband combining, an output quarter-wave transmission line with fixed characteristic impedance can be replaced by a multisection transmission line with different characteristic impedances, which allows the frequency range from 2.2 to 2.96 GHz to be covered [3]. In this case, the broadband matching is realized by applying the simplified real frequency technique with the desired frequency-dependent optimum impedances. However, nonlinear optimization of the entire Doherty amplifier system makes the design complicated enough in terms of circuit simulation and results in a sufficiently large size of the final board implementation. This paper introduces and describes a novel parallel Doherty architecture for different wireless applications, which allows high efficiency across a wide frequency range and backoff output powers to be achieved using a simple transmission-line load network. To further maximize bandwidth efficiency, the

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GREBENNIKOV AND WONG: DUAL-BAND PARALLEL DOHERTY POWER AMPLIFIER FOR WIRELESS APPLICATIONS

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Fig. 3. Reactance compensation Class-E circuit with lumped elements and transmission line and its performance.

Fig. 2. Load-network schematics and broadband properties.

broadband Class-E reactance compensation transmission-line approach for carrier and peaking amplifiers is used. II. PARALLEL ARCHITECTURE A multiband capability of the conventional two-stage Doherty amplifier, whose block schematic is shown in Fig. 1(a), can be achieved when all of its components are designed to provide their corresponding characteristics over the required bandwidth of operation. In this case, the carrier and peaking amplifiers should provide broadband performance when, for example, their input and interstage matching circuits are designed as broadband and the load network generally can represent a low-pass structure with two or three sections tuned to the required frequencies. In a broadband Class-E mode, the

load network can be composed of the consecutive series and parallel resonant circuits using lumped or transmission-line elements according to reactance compensation technique. For a multiband operation with the center frequency ratio at each of the frequency bands of 2 or greater, the input divider can be configured as a multisection Wilkinson power divider or coupled-line directional coupler. In a dual-band operation mode, an input power splitter can represent a -shape or -shape stub tapped branch-line coupler composed of four dual-band quarter-wavelength transmission lines, and an impedance inverter network introducing a 90 phase shift can be based on a - or -type transmission-line impedance-inverting section with proper selected transmission-line characteristic impedances and electrical lengths, where shunt elements are provided by the open-circuit stubs [4], [5]. The delay transmission line at the input of the peaking amplifier can be constructed in a similar way as the multiband impedance transformer at the output of the carrier amplifier by allowing the phase of the signal transmitted through the carrier amplifier path to match the phase of the signal in the peaking amplifier path. However, it should be noted that it is not easy to design a multiband impedance transformer that provides two separate matching options simultaneously: first, to operate in a 50- environment without affecting power-amplifier performance in a high-power region, and secondly, to provide an impedance matching from 25 to 100 in a low-power region. The multiband output combiner required to combine the output powers from the carrier and peaking amplifiers and match the resulting 25- impedance to the standard load impedance of 50 can be realized using the two quarter-wave transmission lines where the characteristic impedance of the first transmission line can be equal to and the characteristic impedance of the second transmission line can be equal to for an intermediate impedance of 35 . Generally, a simple two-stepped transmission-line impedance transformer can provide a two-pole

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Fig. 4. Idealized circuit schematic of broadband Class-E GaN HEMT power amplifier.

Fig. 5. Simulated small-signal

versus frequency.

response with a different characteristic impedance ratio and different electrical lengths of the transmission-line sections [6]. III. BROADBAND LOAD NETWORK The classical two-stage Doherty amplifier has limited bandwidth capability in a low-power region since it is necessary to provide an impedance transformation from 25 to 100 when the peaking amplifier is turned off, as shown in Fig. 2(a), thus resulting in a loaded quality factor at a 3-dB output-power reduction level, which is sufficiently high for broadband operation. The parallel architecture of a two-stage Doherty amplifier with modified modulated load network, whose block schematic is shown in Fig. 1(b), can improve bandwidth properties in a low-power region by reducing the impedance transformation ratio by a factor of 2. In this case, the load network for the carrier amplifier consists of a single quarter-wave transmission line required for impedance transformation, the load network for the peaking amplifier consists of a 50- quarter-wave transmission line followed by another quarter-wave transmission line required for impedance transformation, and the quarter-wave transmission line at the input of the carrier amplifier is necessary for phase compensation. Both impedance-transforming quarter-wave transmission lines, having a characteristic impedance of 70.7

Fig. 6. Simulated results of broadband Class-E GaN HEMT amplifier.

each, provide a parallel connection of the carrier and peaking amplifiers in a high-power region by parallel combining of the two 100- impedances at their output into a 50- load, with 50- impedances at their inputs seen by each amplifier output. In a low-power region below an output-power backoff point of 6 dB, when the peaking amplifier is turned off, the required impedance of 100 seen by the carrier-amplifier output is achieved by using a single quarter-wave transmission line with a characteristic impedance of 70.7 to match with a 50-


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Fig. 7. Circuit schematic of dual-band parallel GaN HEMT Doherty amplifier.

load, as shown in Fig. 2(b). This provides a loaded quality factor , resulting in a 1.73 times wider frequency bandwidth, as shown in Fig. 2(c) by curve 1, compared with a conventional case (curve 2). Since the load network of the peaking amplifier contains two quarter-wave transmission lines connected in series, this provides an overall half-wavelength transmission line, and an open circuit at the peaking-amplifier output directly translates to the load providing a significant isolation of the peaking-amplifier path from the carrier-amplifier path in a wide frequency range. The input in-phase divider and phase-compensating transmission line can be replaced by a broadband coupled-line 90 hybrid coupler. From Fig. 2(c), it follows that the use of a parallel Doherty architecture can provide a broadband operation within 25%–30% around center bandwidth frequency with minimum variation of the load-network transfer characteristic. As a result, a dual-band operation can be easily provided by this architecture, for example, in 1.8-GHz (1805–1880 MHz) and 2.1-GHz (2.11–2.17 GHz) or 2.1-GHz and 2.6-GHz (2.62–2.69 GHz) WCDMA/LTE frequency bands, respectively. IV. BROADBAND CLASS-E POWER AMPLIFIER The conventional design of a high-efficiency switch-mode Class-E power amplifier requires a high factor to satisfy the necessary harmonic impedance conditions at the output device terminal. However, if a sufficiently small value of the loaded

quality factor is chosen, a high-efficiency broadband operation of the Class-E power amplifier can be realized by applying the reactance compensation technique. For example, a simple network consisting of a series resonant circuit tuned to the fundamental frequency and a parallel inductor provides a constant load phase angle of 50 in a frequency range of about 50% [7]. Usually, the bandwidth limitation in power amplifiers comes from the device low transition frequency and large output capacitance; therefore, silicon LDMOSFET technology has been the preferred choice up to 2.2 GHz. As an alternative, GaN HEMT technology enables high efficiency, large breakdown voltage, high power density, and significantly higher broadband performance due to higher transition frequency and smaller periphery, resulting in smaller input and output capacitances and less parasitics. It is very difficult to maintain efficiency at a high level over very wide frequency bandwidth. For a Class-E load network with shunt capacitance, a PAE above 50% was achieved within the frequency range from 1.9 to 2.4 GHz [8]. To increase highefficiency frequency bandwidth, the broadband Class-E technique based on a reactance compensation principle with a combination of the series and shunt resonant circuits can be used [9]. In this case, a PAE over 53% was observed in a frequency bandwidth of 2.1–2.7 GHz with output power variations from 9.3 to 12.7 W at a supply voltage of 40 V [10].

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Fig. 8. Simulated small-signal


and

versus frequency.

Fig. 3(a) shows the example of a reactance-compensation load network for a Class-E power amplifier with shunt capacitance including a series transmission line and a parallel resonant circuit. In this case, the reactance of the Class-E load network with shunt capacitance and series inductance varies similar to that of the series resonant circuit with positive slope, whereas the required negative slope is provided by the parallel resonant circuit. Selection of the proper characteristic impedance and electrical length of the series transmission line enables the magnitude of the two slopes to be made identical so as to achieve a constant total reactance and phase of the load network impedance over a wide frequency range. The simulation results at the fundamental frequency show that the resistance varies from 35 at 30 MHz to 68 at 70 MHz, as shown in Fig. 3(b) by curve 1, whereas the load-network phase varies from 27 to 40 in more than octave bandwidth from 33 to 80 MHz (curve 2). Generally, very broadband power-amplifier design employs an input lossy matching circuit to minimize the input return loss and output power variations over very wide frequency bandwidths with an output network to compensate for the device output reactance [11], [12]. As an alternative, to provide an input broadband matching over an octave bandwidth, it is possible to use a multisection matching transformer consisting of stepped transmission-line sections with different characteristic impedances and electrical lengths [9]. Such an input matching structure is convenient in practical implementation since there is no need to use any tuning capacitors. Fig. 4 shows the idealized simulation setup of a 10-W broadband Class-E power amplifier circuit designed to operate over a frequency bandwidth from 1.7 to 2.7 GHz and based on a GaN HEMT Cree CGH40010 (or CGH27015) device, where both the input matching circuit and load network are composed of ideal transmission lines. The nominal Class-E load resistance can be calculated for W, V, and V as (1) where is the output power at the fundamental frequency, is the drain supply voltage, and is the saturation voltage defined from the device output current–voltage characteristics [9]. In this case, the transmission-line parallel resonant circuit in the broadband Class-E load network having a 25- load is

Fig. 9. Simulated broadband capability of parallel Doherty amplifier.

Fig. 10. Simulated results of dual-band Doherty amplifier.

represented by the open- and short-circuit stubs replacing the lumped capacitor and inductor, respectively, each having a characteristic impedance of 50 and electrical length of 45 at 2.0 GHz. An additional series transmission line with 35- characteristic impedance and quarter wavelength at the high bandwidth frequency of 2.7 GHz is used to match an idealized 25load with a standard 50- load.


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Fig. 11. Test board of dual-band GaN HEMT Doherty amplifier.

Fig. 14. Measured results of dual-band GaN HEMT Doherty amplifier without linearization at 2.655 GHz.

Fig. 12. Measured results for small-signal

- and

-parameters.

Fig. 15. Measurements of ACLR with and without DPD for LTE signal.

Fig. 13. Measured results of dual-band GaN HEMT Doherty amplifier without linearization at 2.14 GHz.

Fig. 5 shows the simulation results for the small-signal -parameters versus frequency demonstrating the in-band return loss for a broadband Class-E power amplifier. As a result, an output power of more than 41 dBm with a power gain of around 10 dB was simulated for an input power of 31 dBm, as shown in Fig. 6(a). In this case, drain efficiency over 73% and PAE over 67% were achieved across the required frequency range from 1.7 to 2.7 GHz, as shown in Fig. 6(b). Previously, drain efficiency greater than 60% was achieved between 1.8 and 2.3 GHz with a 45-W GaN HEMT Cree CGH40045F

device using a distributed second-harmonic termination with short-circuit stubs [13]. V. SIMULATION Fig. 7 shows the simulated circuit schematic of a parallel GaN HEMT Doherty configuration, where the carrier and peaking amplifiers are based on broadband transmission-line Class-E power amplifiers, each having an idealized circuit structure shown in Fig. 4. The input matching circuits and output load network are based on microstrip lines with their parameters corresponding to a 20-mil RO4360 substrate. Special care was taken for modeling of the device input and output package leads to account for finite values of their inductances. The ideal 90 hybrid coupler is used at the Doherty amplifier input to split signals between the carrier and peaking amplifying paths,

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TABLE I PERFORMANCE SUMMARY OF BROADBAND/MULTIBAND DOHERTY AMPLIFIERS

which also provides a 90 phase shift at the input of the carrier amplifier path required for a parallel Doherty power amplifier. Offset lines of equal electrical lengths are used at the output of the carrier and peaking amplifiers. The electrical lengths of both offset and combining microstrip lines were optimized to maximize efficiency at saturated and backoff output power levels by providing high-impedance condition seen by the carrier amplifier when the peaking amplifier is turned off. Fig. 8 shows the simulation results for the small-signal - and -parameters versus frequency demonstrating the wide bandwidth capability of a parallel transmission-line GaN HEMT Doherty power amplifier covering a frequency range from 2.0 to 2.8 GHz with a power gain over 10 dB. In this case, an input return loss defined from the magnitude of is less than 5 dB over the frequency bandwidth of 2.1–2.9 GHz. Fig. 9 demonstrates the broadband capability of a parallel Doherty structure, where the carrier and peaking amplifiers are based on a broadband transmission-line reactance compensation Class-E technique. In an amplifier saturation mode with an input power of 36 dBm, a drain efficiency of around 70% with an average output power of more than 43 dBm and a gain variation of about 1 dB was simulated across the frequency range of 2.0–2.8 GHz, as shown in Fig. 9(a). At the same time, high drain efficiencies over 50% at backoff output powers of 5–6 dB from saturation can potentially be achieved across the frequency range of 2.1–2.7 GHz, as shown in Fig. 9(b). This means that the practical implementation of a parallel Doherty power amplifier, the simulation setup of which is shown in Fig. 7, can provide a highly efficient operation in two cellular bands of 2.11–2.17 and 2.62–2.69 GHz without any tuning of the amplifier load-network parameters, either with separate or simultaneous dual-band transmission of WCDMA or LTE signals.

The large-signal simulations versus input power have been done at two center bandwidth frequencies of 2.14 and 2.655 GHz with optimized circuit parameters to achieve maximum performance. Fig. 10 shows the simulated large-signal power gain and drain efficiencies of a dual-band transmission-line GaN HEMT parallel Doherty amplifier based on a 20-mil RO4360 substrate with the carrier gate bias V, peaking gate bias V, and dc supply voltage V. In this case, a linear power gain of about 11 dB was achieved at an operating frequency of 2.655 GHz, as shown in Fig. 10(b), whereas a slightly higher linear power gain of about 12 dB was achieved at lower operating frequency of 2.14 GHz, as shown in Fig. 10(a). In a large-signal operation mode, high drain efficiencies of 64% and 53% were simulated at backoff output powers of 39 dBm ( 4-dB backoff) and 37 dBm ( 6-dB backoff), respectively, at both center bandwidth frequencies. Here, a peak efficiency point near 4-dB backoff output power at high bandwidth frequency is clearly seen, while high efficiency maintains almost constant at high output powers at low bandwidth frequency. VI. IMPLEMENTATION The dual-band transmission-line GaN HEMT Doherty power amplifier was fabricated on a 20-mil RO4360 substrate. An input splitter represents a broadband coupled-line coupler from Anaren, model 11306-3, which provides a maximum phase balance 5 and amplitude balance 0.55 dB across the frequency range of 2–4 GHz. Fig. 11 shows the test board of a dual-band parallel two-stage Doherty power amplifier based on two 10-W Cree GaN HEMT power transistors CGH40010P in metal–ceramic pill packages. The input matching circuit, output load network, and gate and


drain bias circuits (having bypass capacitors on their ends) are fully based on microstrip lines of different electrical lengths and characteristic impedances according to the simulation setup shown in Fig. 7. Special care should be taken in order to minimize the input and output lead inductances of the packaged GaN HEMT device, which can significantly affect the power-amplifier performance such as power gain, output power, and efficiency. VII. MEASUREMENTS Fig. 12 shows the measured small-signal and parameters across the frequency range of 1.8–3.0 GHz, where the magnitude of varies from 4.8 to 13.2 dB and the magnitude of varies between 13 and 15 dB for equal gate bias voltages for carrier and peaking amplifiers, providing a total quiescent current of 200 mA. Here, the simulated small-signal -parameters with increasing values at lower bandwidth frequencies because of an ideal 90 input coupler used in the simulation setup are also shown. Significant variations of the measured -parameters can be explained by a sufficiently high amplitude imbalance of the broadband coupler and some nonidentity of the peaking and carrier amplifying paths including devices gate lead inductances. For a single-carrier 5-MHz WCDMA signal with a peak-toaverage ratio (PAR) of 6.5 dB, a drain efficiency of 45% with a power gain of about 10 dB and lower than 30 dBc at 2.14 GHz, as shown in Fig. 13, and a drain efficiency of 40% with a power gain of about 11 dB and around 30 dBc at 2.655 GHz, as shown in Fig. 14, were achieved at an average output power of 39 dBm. In both cases, optimization of the gate bias voltages for carrier (Class-AB mode) and peaking (Class-C mode) amplifiers were provided. As an example of linearization capability, Fig. 15 shows the results of applying an Optichron DPD linearizer to a dual-band transmission-line GaN HEMT Doherty power amplifier with a single-carrier 10-MHz LTE signal having a PAR of 7.6 dB at a center bandwidth frequency of 2.14 GHz, resulting in a corrected output power of 38.6 dBm with a drain efficiency of 45%, of 55 dBc, and of 57 dBc for optimized gate bias voltages applied to the carrier and peaking amplifiers. A comparison Table I shows the performance summary of various practical broadband/multiband Doherty power amplifiers implemented in LDMOSFET or GaN HEMT technologies and using symmetrical or asymmetric Doherty configurations with similar or close output powers. VIII. CONCLUSION A novel dual-band transmission-line parallel Doherty amplifier architecture for active antenna arrays and base-station applications in next-generation communication systems is presented. The carrier and peaking amplifiers using GaN HEMT Cree CGH40010P devices are designed based on the reactance compensation technique to provide optimum Class-E impedance seen by the device output at the fundamental frequency across the wide frequency range achieving the drain efficiencies over 73% across the frequency range from 1.7 to 2.7 GHz. In a single-carrier WCDMA operation mode with

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a PAR of 6.5 dB, high drain efficiencies of 40%–45% can be achieved at an average output power of 39 dBm with an of about 30 dBc at center bandwidth frequencies of 2.14 and 2.655 GHz. REFERENCES [1] K. Bathich, A. Z. Markos, and G. Boeck, “A wideband GaN Doherty amplifier with 35% fractional bandwidth,” in Proc. 40th Eur. Microw. Conf., 2010, pp. 1006–1009. [2] K. Bathich, D. Gruner, and G. Boeck, “Analysis and design of dualband GaN HEMT based Doherty amplifier,” in Proc. 6th Eur. Microw. Integr. Circuits Conf., 2011, pp. 248–251. [3] G. Sun and R. H. Jansen, “Broadband Doherty power amplifier via real frequency technique,” IEEE Trans. Microw. Theory Tech., vol. MTT-60, no. 1, pp. 99–111, Jan. 2012. [4] H. Zhang and K. J. Chen, “A stub tapped branch-line coupler for dualband operations,” IEEE Microw. Wireless Compon. Lett., vol. 17, no. 2, pp. 106–108, Feb. 2007. [5] W. Chen, S. A. Bassam, X. Li, Y. Liu, K. Rawat, M. Helaoui, F. M. Ghannouchi, and Z. Feng, “Design and linearization of concurrent dual-band Doherty power amplifier with frequency-dependent power ranges,” IEEE Trans. Microw. Theory Tech., vol. MTT-59, no. 10, pp. 2537–2546, Oct. 2011. [6] C. Monzon, “A small dual-frequency transformer in two sections,” IEEE Trans. Microw. Theory Tech., vol. MTT-51, no. 4, pp. 1157–1161, Apr. 2003. [7] J. K. A. Everard and A. J. King, “Broadband power efficient Class E amplifiers with a non-linear CAD model of the active MOS device,” J. Inst. Electron. Radio Eng., vol. 57, pp. 52–58, Mar. 1987. [8] H. Xu, S. Gao, S. Heikman, S. I. Long, U. K. Mishra, and R. A. York, “A high-efficiency Class-E GaN HEMT power amplifier at 1.9 GHz,” IEEE Microw. Wireless Compon. Lett., vol. 16, no. 1, pp. 22–24, Jan. 2006. [9] A. Grebennikov, RF and Microwave Power Amplifier Design. New York: McGraw-Hill, 2004. [10] M. P. van der Heijden, M. Acar, and J. S. Vromans, “A compact 12-watt high-efficiency 2.1–2.7 GHz Class-E GaN HEMT power amplifier for base stations,” in IEEE MTT-S Int. Microw. Symp. Dig., 2009, pp. 657–660. [11] Y.-F. Wu, R. A. York, S. Keller, B. P. Keller, and U. K. Mishra, “3–9-GHz GaN-based microwave power amplifiers with L–C–R broadband matching,” IEEE Microw. Guided Wave Lett., vol. 9, no. 8, pp. 314–316, Aug. 1999. [12] K. Krishnamurthy, D. Green, R. Vetury, M. Poulton, and J. Martin, “0.5–2.5 GHz, 10 W MMIC power amplifier in GaN HEMT technology,” in IEEE Compound Semiconduct. Integr. Circuits Symp. Dig., 2009, pp. 1–4. [13] J. Kim, F. Mkadem, and S. Boumaiza, “A high efficiency and multiband/multi-mode power amplifier using a distributed second harmonic termination,” in Proc. 40th Eur. Microw. Conf., 2010, pp. 1662–1665. [14] J. H. Qureshi, N. li, W. C. E. Neo, F. van Rijs, I. Blednov, and L. C. N. de Vreede, “A wideband 20 W LDMOS Doherty power amplifier,” in IEEE MTT-S Int. Microw. Symp. Dig., 2010, pp. 1504–1507. [15] K. Bathich, A. Z. Markos, and G. Boeck, “Frequency response analysis and bandwidth extension of the Doherty amplifier,” IEEE Trans. Microw. Theory Tech., vol. MTT-59, no. 4, pp. 934–944, Apr. 2011.

Andrei Grebennikov (M’99–SM’04) received the Dipl. Ing. degree in radio electronics from the Moscow Institute of Physics and Technology, Moscow, Russia, in 1980, and the Ph.D. degree in radio engineering from the Moscow Technical University of Communications and Informatics, Moscow, Russia, in 1991. He amassed his long-term academic and industrial experience working with the Moscow Technical University of Communications and Informatics, Moscow, Russia, the Institute of Microelectronics, Singapore, M/A-COM, Cork, Ireland, Infineon Technologies, Munich, Germany, and Linz, Austria, and Bell Laboratories, Alcatel-Lucent, Dublin, Ireland, as an Engineer, Researcher, Lecturer, and Educator. He has lectured as

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a Guest Professor with the University of Linz, Linz, Austria. He has authored or coauthored over 80 papers. He authored five books dedicated to RF and microwave circuit design. He holds 20 European and U.S. patents. Dr. Grebennikov has presented short courses and tutorials as an invited speaker at the IEEE Microwave Theory and Techniques Society (IEEE MTT-S) International Microwave Symposium (IMS), European and Asia–Pacific Microwave Conferences, the Institute of Microelectronics, Singapore, Motorola Design Centre, Penang, Malaysia, the Tomsk State University of Control Systems and Radioelectronics, Tomsk, Russia, and Aachen Technical University, Aachen, Germany.

James Wong (M’99) received the B.Eng. degree (with honors) in electrical and electronics engineering and Ph.D. degree in engineering from the University of Surrey, Surrey, U.K., in 1999 and 2003, respectively. He has previously worked within the telecommunications industry (Nokia Networks, Filtronic PLC, Nujira Ltd, Astrium EADS), during which time he has been involved on various amplifier designs and architectures. He is currently with Alcatel-Lucent Telecom, Swindon, U.K., as their RF Power Amplifier Specialist, leading their global Advance Technologies Power Amplifier Team. He holds several European patents with several patent applications pending. His current research and industrial activity covers advanced amplifier architectures and linearization techniques.