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IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 63, NO. 1, JANUARY 2015

A Dual-Loop Antenna Design for Hepta-Band WWAN/LTE Metal-Rimmed Smartphone Applications Yong-Ling Ban, Yun-Fei Qiang, Zhi Chen, Kai Kang, and Jin-Hong Guo

Abstract—A simple direct-fed dual-loop antenna capable of providing hepta-band WWAN/LTE operation under surroundings of an unbroken metal rim in smartphone applications is proposed. The greatest highlight of this proposed antenna is that it provides a simple and effective multiband antenna solution for an unbroken metal-rimmed smartphone. The unbroken metal rim with 5 mm in height embraces the system circuit board of 130 70 mm . Two no-ground portions of 10 70 mm and 5 70 mm are set on the top and bottom edge of the system circuit board, respectively. In-between the two separate no-ground portions, there is a system ground of 115 70 mm connected with the unbroken metal rim via a small grounded patch which divides the unbroken metal rim into two strips. Finally, a dual-loop antenna is formed by combining the inner system ground and two strips. This proposed dual-loop antenna is capable of covering GSM850/900/DCS/PCS/UMTS2100/LTE 2300/2500 operating bands. Detailed design considerations of the proposed antenna are described and both experimental and simulation results are also presented and discussed. Index Terms—Loop antenna, unbroken metal rim, WWAN/LTE antenna.

I. INTRODUCTION

I

N recent years, smartphones have entered into a rapid development period and have gradually become the main communication tools [1]. Furthermore, a smartphone with an unbroken metal rim has become an obvious trend. The metal rim can not only provide sufficient mechanical strength to extend the service life of the smartphone, but also can possess a wonderful appearance, which is very desirable for consumers. Manuscript received April 27, 2014; revised October 14, 2014; accepted November 03, 2014. Date of publication November 07, 2014; date of current version December 31, 2014. This work was supported in part by the National Higher-education Institution General Research Development Project (2013ZX03001024), in part by the National Science and Technology Specific Projects of China (ZYGX2013J013), in part by the National Science Fund of China (61471098), and in part by the China Scholarship Fund and OATF, UESTC. Y.-L. Ban, Y.-F. Qiang, and K. Kang are with the Institute of Electromagnetics and School of Electronic Engineering, University of Electronic Science and Technology of China (UESTC), Chengdu 611731, China (e-mail: [email protected]). Z. Chen is with the National Key Lab of Science and Technology on Communication, University of Electronic Science and Technology of China (UESTC), Chengdu 611731, China. J.-H. Guo is with Department of Biomedical Engineering, Nanyang Technological University, 70 Nanyang Drive, Singapore 63745. Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TAP.2014.2368573

Not surprisingly, the performance of the previous antennas designed for smartphones will be affected dramatically, if the metal rim, without any modification, is placed around the housing of the smartphone. In [2], it has presented a detailed explanation about how the unbroken metal rim affects the performance of the internal antenna. The unbroken metal rim sets up a bad feedback link to the internal antenna which introduces an undesired coupling between the unbroken metal rim and the internal antenna. The undesired coupling affects the performance of the internal antenna adversely which will increase the design difficulty for antenna designers to achieve multiband of an antenna. Recently, several promising solutions [2], [3] have been demonstrated which can resolve the effects of the metal rim. For example, in [2], it has shown a method to reduce the effects of the metal rim by inserting three gaps and two grounded patches. Besides, by judiciously choosing the locations of the gaps and the grounded patches, this method can alleviate the effects of the metal rim. In [3], it has presented a compact slot antenna of 15.5 56.5 mm by adding several grounded patches between the bottom system ground and the unbroken metal rim. The two slots are fed by the same feeding strip, which can cover five WWAN bands of GSM850/900/DCS/PCS/UMTS2100 operation. Seen from the above discussion, both of them occupy too much space of the PCB and the width of the narrow edges of these two antennas is always more than 15 mm, which are not suitable for narrow-frame antenna designs [4], [5]. In addition, another promising candidate for the metal-rimmed smartphone antenna is exciting and employing the different chassis' characteristic modes [6]. However, its biggest drawback is the narrow-band operations. In order to widen the impedance bandwidth, many effective techniques have been reported in [7]–[13]. The usual effective techniques of widening the impedance bandwidth include the matching network [7]–[10], coupled-fed [11] and reconfigurable technique [12], [13]. In [7], it has introduced a novel antenna structure combining a nonself-resonant CCE and a self-resonant ILA antenna occupying only 750 mm . However, it needs a matching network which needs a correct selection of low-loss components. In [8], it has proposed a small antenna system using nonresonant planar elements for 2G, 3G, and 4G occupying about 700 mm . In [11], it has presented a coupled-fed dual-loop antenna capable of providing eight-band WWAN/LTE operations. This method indeed can widen the impedance matching but it will increase the difficulty

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BAN et al.: DUAL-LOOP ANTENNA DESIGN FOR HEPTA-BAND WWAN/LTE METAL-RIMMED SMARTPHONE APPLICATIONS

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of antenna tuning in the final optimization process, because its performance is very sensitive to the coupled gaps between the feeding strip and shorting strip. In [12], the reconfigurable antenna is controlled by one p-i-n diode to choose the antenna mode between loop antenna mode and an inverted-F antenna mode. Unfortunately, the p-i-n diode will also introduce insertion loss. To alleviate these problems, a simple direct-fed dual-loop antenna capable of providing hepta-band WWAN/LTE operation under surroundings of an unbroken metal rim in smartphone applications is proposed in this work. The unbroken metal rim is directly fed by a mini coaxial feed line and then connected to the system ground by a small grounded patch. Consequently the unbroken metal rim is divided into two stripes and a dual-loop antenna is formed by combining the inner system ground and two strips. The biggest merit of the proposed antenna is that it keeps the integrity of the metal rim and sufficient bandwidth to cover hepta-band WWAN/LTE operation. Hence, it is very promising for metal-rimmed smartphone applications. The organization of this paper is arranged as follows. In Section II, the antenna geometry and design are described. The design process and parameters are discussed more in detail in Section III. The proposed dual-loop antenna was fabricated and the test results of the measured S parameter, gain, and radiation patterns are shown in Section IV, in which the effects of adding components and the user's hand, the specific absorption rate (SAR) values and state-of-the-art comparison are also presented. Some conclusions are given in Section V. II. PROPOSED ANTENNA CONFIGURATION Fig. 1(a) shows the geometry of the proposed dual-loop antenna formed by an unbroken metal rim for WWAN/LTE smartphones, whose detailed structure and optimized dimensions are given in Fig. 1(b). As illustrated in Fig. 1(a), a 0.8-mm thick FR4 substrate of relative permittivity 4.4 and loss tangent 0.024 is used as the system circuit board. The system circuit board of 130 70 mm is embraced by an unbroken metal rim whose height is of 5 mm and thickness is of 0.3 mm. The distance between the system circuit board and the metal rim is 2 mm [3]. Two no-ground portions of 10 70 mm and 5 70 mm are set on the top and bottom edge of the system circuit board, respectively. In-between the two separate no-ground portions, there is a system ground plane with the length of 115 mm and the width of 70 mm. mini coaxial feed line is Seen from Fig. 1(b), a employed to excite the antenna connected to the feeding point (point A) and the grounded point (point B). The distance between the feeding point and the bottom edge of the system circuit board is 25 mm. The unbroken metal rim directly fed mini coaxial feed line is connected to the system by a ground by a small grounded patch. The unbroken metal rim is divided into two stripes by the grounded patch. Finally, the dual-loop antenna is formed by combining the inner system ground and two strips. The distance between the grounded patch and he bottom edge of the system circuit board is 50 mm. is about 260 mm which The length of the Loop 1 loop mode (at 0.67 GHz) as the allows it to generate a

Fig. 1. Proposed antenna configuration: (a) Geometry of the metal-rimmed antenna for hepta-band operations in smartphone applications. (b) Detailed dimensions of the proposed antenna (Unit: mm).

fundamental mode. The high-order resonant mode of Loop 1 such as and modes are also excited. The is about 156 mm (about at length of the Loop 1.13 GHz) which can provide two high-order resonant modes ( and modes). The two fundamental modes of the Loop 1 and Loop 2 generate a wide bandwidth to cover the GSM850/900 operation. The desired upper-band DCS/PCS/UMTS2100/LTE2300/2500 operation is provided by the high-order modes of Loop 1 and Loop 2. III. DESIGN PROCESS AND PARAMETER STUDY A. Analyses of Resonant Modes and Current Distributions To fully comprehend the excited modes of the proposed dual-loop antenna, two reference antennas (Ref-1 and Ref-2) are introduced. Fig. 2(a) shows the simulated input reactance of the proposed antenna and the reference antennas, the case with Loop 1 only (Ref-1) and the case with Loop 2 only (Ref-2). The modes generated by the Ref-1 and the Ref-2 are marked as

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IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 63, NO. 1, JANUARY 2015

Fig. 3. (a) Simplified equivalent circuits of the proposed antenna. (b) Simulated input resistance of the proposed antenna and the case with Ref-1 (Loop 1 only), and the case with Ref-2 (Loop 2 only).

Fig. 2. (a) Simulated input reactance and (b) simulated S parameters, respectively, of the proposed antenna, the case with Loop 1 only (Ref-1) and the case with Loop 2 only (Ref-2).

and (2) The input impedance of the proposed antenna is marked as Hence, the input impedance of can be written as

.

and , respectively, where represents the -th order modes. Seen from the Fig. 2, the proposed dual-loop has five resonant modes ( or ). Fig. 2(b) shows the corresponding simulated S parameters of the Fig. 2(a). The multiresonant character for both Ref-1 and Ref-2 can be easily seen from Fig. 2(b). The bandwidth of each resonant mode of Ref-1 and -2 is not wide enough to cover the whole operating frequency. However, by combining the multiresonant modes provided by the Ref-1 and Ref-2, the proposed antenna can provide two wide operating bands of 824–960 MHz and 1710–2690 MHz, respectively. In addition, seen from Fig. 2(a), two modes of are not appeared in the proposed antenna while the other modes are still remaining. This is explained as follows. Fig. 3(a) shows a simplified equivalent circuit of the proposed antenna which is formed by the Ref-1 and Ref-2 antenna in parallel. The radiation impedance of Ref-1 (Loop1 only) and Ref-2 (Loop2 only) are marked, respectively, as

and simultaneously, then according Lastly, if to (4), the reactance of input impedance is

(1)

(7)

(3) where and are the input resistance and input reactance of the proposed antenna respectively. Combining (1), (2) with (3), the input reactant can be written as

(4) If

, the (4) can be simplified as (5)

If

, the (4) can be simplified as (6)

BAN et al.: DUAL-LOOP ANTENNA DESIGN FOR HEPTA-BAND WWAN/LTE METAL-RIMMED SMARTPHONE APPLICATIONS

Seen from Fig. 3(b), for the mode , both the values of and (about 20 ) are comparable to the absolute value of (about 23 ). For the other mode , the values of and are about 45 and the absolute value of is about 70 . Hence, according to (5), the absolute value of is just smaller than the absolute value of , but can not approaching to be zero. That's why the two modes of and are not appeared in the proposed antenna. However, for the mode , the values of (about 4 ) and (about 30 ) are very small relative to the absolute value of (about 223 ). The values of and are about 28 and 140 , respectively, for the mode . According to (5), the value of will approximate to be zero indicating these two modes of , and will still exist in the proposed antenna. Again referring to Fig. 3(b), for the mode , both the values of and (about 18 ) are small relative to the absolute value of (about 50 ). For the other mode , the value of (about 30 ) is very small relative to the value of (about 215 ) and the absolute value of (about 122 ). According to (6), the value of will approximate to be zero which means these two modes will appear in the proposed antenna modes. For the last two modes of and which have a same resonant frequency at about 2.93 GHz. The (7) shows the two modes of and can be preserved in the proposed antenna. All of the modes marked in Fig. 2(a) have been analyzed reasonably. In order to distinguish the resonant modes of the proposed antenna, simulated surface vector current distributions at frequencies of 0.67 GHz, 1.12 GHz, 1.98 GHz, 2.33 GHz and 2.93 GHz, which corresponds to the modes of (or ), are plotted in Fig. 4(a)–(e), respectively. Seen from the Fig. 4(a), the simulated surface current has a null point along the Loop 1 at 0.67 GHz indicating that the proposed antenna operates in which is the fundamental mode of the Loop 1. The other modes of and can be analyzed easily from Fig. 4 and the surface current distributions of the corresponding resonant frequencies in Fig. 4(b)–(d). However, an interesting phenomenon appears on the surface current distribution of the proposed antenna at 2.93 GHz. Seen from Fig. 4(e), although the two modes and have a same resonant frequency at 2.93 GHz, the surface current is mainly distributed along the Loop 2, which can be explained by using a simplified equivalent circuit and the input impedance of Ref-1 and Ref-2 shown in Fig. 3. Seen from Fig. 3(a), the radiation impedance of Ref-1 is larger than the radiation impedance of Ref-2 at 2.93 GHz. Hence, the current flowing into is smaller than the current flowing into . Finally, the current flows into will be very large and more obvious which agrees the simulated surface current in Fig. 4(e). B. Parameter Study of the Proposed Antenna The two no-ground portions play an important role in the impedance matching of the proposed antenna. The effects of the

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Fig. 4. Simulated surface current distributions in the proposed antenna at the corresponding resonant frequencies of (a) 0.67 GHz; (b) 1.12 GHz; , (b) , (c) (c) 1.98 GHz; (d) 2.33 GHz; (e) 2.93 GHz. (a) , (d) , (e) .

size of the no-ground portions are also analyzed, where the simulated results are included in Fig. 5(a) and (b). Fig. 5(a) shows the influence on the antenna performance when varying the width of the top no-ground portion. Large effects on the impedance matching of the frequencies over the lowerband are seen when the length varied from 5 to 15 mm For mm, the impedance matching over the lower-band is not good. As increasing the width , the improved impedance matching of the lower-band is obtained. In this study, the width of the top no-ground portion is chosen as 10 mm for good impedance matching and minimizing the size of the top no-ground portion. Seen from Fig. 5(b), the width of the bottom no-ground portion affects the impedance matching over the lower-band and upper-band. The impedance matching is very poor if without the bottom no-ground portion. Taking into account both bandwidth and miniaturization of the proposed antenna, The width of 5 mm is a suitable choice for the bottom no-ground portion. Effects of the positions of the feeding point and grounded point are also analyzed in Figs. 6 and 7, respectively. With the increase of the length between the feeding point and the bottom edge of the system circuit board, the length of Loop 1 will reduce while the length of Loop 2 will increase. Therefore, seen from the Fig. 6, the corresponding resonant frequencies of and are shifted to higher frequencies and the corresponding resonant frequencies of , and are shifted to lower frequencies.

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IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 63, NO. 1, JANUARY 2015

Fig. 7. Simulated input impedance for the proposed antenna as a function of the length Lg between the ground point and the bottom edge of the system circuit board (other dimensions are the same as given in Fig. 1).

Fig. 5. Simulated S parameters for the proposed antenna as a function of of the top no-ground portion and (b) the width of the (a) the width bottom no-ground portion (other dimensions are the same as given in Fig. 1).

Fig. 6. Simulated input impedance for the proposed antenna as a function of the length Lf between the feeding point and the bottom edge of the system circuit board (other dimensions are the same as given in Fig. 1).

Furthermore, for the lower-band, the variations of the resonant frequencies are very small while the variations of the

resonant frequencies are quite large for the upper-band. This can be mainly explained as follows. The variations of the lengths of Loop 1 and Loop 2 relative to its resonant lengths of the fundamental modes ( and ) are very small which will result in small effect on the corresponding resonant frequencies. However, for the upper-band, the variation of the lengths of Loop 1 and Loop 2 relative to their resonant lengths of the high order modes ( and ) are relatively large which will result in significant effect on the corresponding resonant frequencies. Similar variations of the resonant frequencies of the proposed antenna can be seen in Fig. 7 when altering the length between the grounded point and the bottom edge of the system circuit board. All of these variations can be explained by the above theories in the foregoing paragraphs. In addition, according to Figs. 6 and 7, the input resistance of the proposed antenna can be tuned closer to 50 Ohms by adjusting the feeding and grounding point locations. The obtained bandwidths can cover the two operating bands of 824–960 MHz and 1710–2690 MHz. Seen from the above parameter studies about the proposed antenna, the method to cover LTE700 band maybe achieved by reducing the length of Lg and Lf which can shift the fundamental resonant frequencies to lower-frequencies. In order to achieve good impedance matching for the desired bandwidth, some matching circuit maybe needed. At present, the typical thicknesses of smartphones are in the range of 6.2–10.7 mm [15]. The proposed unbroken metal rim whose height is of 5 mm can suit for the slimmest smartphones at present. However, the thickness of the metal rim is not fixed for different smartphones. Therefore, the discussion about the effect on the performance of the proposed unbroken metal rim with different heights is necessary. Results of the simulated S parameters of the proposed antenna with different heights are shown in Fig. 8. Tiny effects on the impedance matching of the frequencies over the lower- and upper-band are seen from Fig. 8 when the height of the metal rim varied from 5 to 9 mm. Therefore, this proposed unbroken metal-rimmed antenna can suit for many styles of smartphones with different profiles.

BAN et al.: DUAL-LOOP ANTENNA DESIGN FOR HEPTA-BAND WWAN/LTE METAL-RIMMED SMARTPHONE APPLICATIONS

Fig. 8. Simulated S parameters of the proposed antenna as a function of the height of the unbroken metal rim (other dimensions are the same as given in Fig. 1).

Fig. 9. Photos of the manufactured printed antenna for hepta-band WWAN/LTE operation in a smartphone: (a) front side; (b) back side; (c) testing environment.

IV. MEASUREMENT AND ANTENNA PERFORMANCE A. Free Space The proposed unbroken metal-rimed antenna for hepta-band operations was fabricated and tested, as shown in Fig. 9. From Fig. 9, it can be seen that the unbroken metal rim is fixed by several foams with thickness of 2 mm. The simulated results are obtained by Ansoft HFSS version 13, and the measured results are tested by an Agilent N5247A vector network analyzer which the testing environment has shown in Fig. 9(c). The part of the cable at the outer system ground may affect the antenna performances. Hence, the ferrite bead used in the measurements is mainly to reduce this effect. The measured and simulated S parameters for the prototype are presented in Fig. 10. The measured impedance bandwidth based on [14]–[19], which is widely used as the design specification of the WWAN/LTE smartphone antennas, is seen to cover 3:1 VSWR the desired 824–960 MHz and 1710–2690 MHz bands for the hepta-band WWAN/LTE operations. The radiation characteristics, the gain and the total efficiency of the proposed antenna are also measured in the SATIMO microwave anechoic chamber. Fig. 11 plots the measured radiation patterns at 86 MHz, 925 MHz, 1830 MHz, 2100 MHz, and

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Fig. 10. Simulated and measured S parameters against frequency for the proposed unbroken metal rim antenna.

2520 MHz, respectively, which are similar to those of many reported LTE/WWAN antennas [21]–[23]. Dipole-like radiation patterns at 860 MHz and 925 MHz are seen in Fig. 11(a) and (b), and omnidirectional radiation in the azimuthal plane (xy-plane) is observed. The high-order resonant frequencies at 1830 MHz, 2100 MHz, and 2520 MHz are also plotted in Fig. 11(c), (d), and (e) from which the complementary and radiation in the xy-plane are seen. Hence, the patterns of the proposed antenna at upper-band are still suitable for the practical communication environments. This is mainly because the transmitting wave transmitted from a base station will have comparable levels for the two components of and after reflection, scattering and diffraction. For example, in the case of the xy-plane pattern at 2100 MHz in Fig. 11(c), at , 180 , the components are both nulls but the components are not nulls, which can ensure a smartphone can receive at least one component of or to guarantee the quality of the communication. Therefore, these complementary patterns of the proposed antenna in smartphone can provide the robustness for the practical communication environments. In addition, dipole-like radiation patterns in Fig. 11(a) and (b) can be explained as follows. Fig. 12 shows the evolution from the traditional folded dipole to our proposed antenna. Fig. 12(a) shows a traditional folded dipole ( mode). In Fig. 12(b), the traditional folded dipole is divided into two portions, where the top portion (solid line) is provided by the antenna element and the bottom portion (dotted line) is simplified from the system ground. It means that the antenna element plus the system ground behave as a dipole (the antenna element is one part of folded monopole and the PCB is the other part of folded monopole). Hence, the antenna element in Fig. 12(b) is called folded monopole antenna. Fig. 12(c) is just a simple deformation from Fig. 12(b). Fig. 12(d) is achieved from Fig. 12(c) by turning the bottom (dotted line) to the top-side. In Fig. 12(d), the outer portion (solid line) is provided by the metal rim and the inner portion is still simplified from the system ground. Finally, comparing Fig. 12(d) with Fig. 4(a), the similar current distributions along

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IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 63, NO. 1, JANUARY 2015

Fig. 11. Measured 2-D radiation patterns at (a) 860 MHz, (b) 925 MHz, (c) 1830 MHz, (d) 2100 MHz, and (e) 2520 MHz for the proposed antenna (red line is , blue line is , Unit: dBi).

and the corresponding total efficiency is larger than 60%. As a result, the measured radiation characteristics of the proposed antenna within the operating band are suitable for meeting the requirement for smartphone systems [24]. B. Effect of Handset Components

Fig. 12. Evolution of our proposed antenna. (a) Traditional folded dipole. folded monopole (b) Folded monopole (the antenna element is one part of folded monopole). (c) Deformed and the system ground is the other part of folded monopole. (d) Simplification from our proposed antenna (the solid line is provided by the antenna element and the dotted line is simplified from the system ground. Both the solid dots and hollow dots denote the current nulls where the current direction will change to the opposite direction).

the outer metal rim and inner system ground are obtained. Hence, the entire evolution from the traditional folded dipole to our proposed antenna is shown in Fig. 12 which has explained the dipole-like radiation patterns at lower frequency bands. In addition, seen from Fig. 12(d), the current distribution along the outer portion (solid line) is opposite to the current distribution along the inner portion. The proposed antenna can still radiate effectively, which is due to that our proposed antenna behaves as an asymmetric dipole causing the current distribution . The measured total efficiency and antenna gain of the fabricated antenna are presented in Fig. 13. For the lower-band of GSM850/900 (824–960 MHz), the antenna gain varies from about 1.2 to 2.0 dBi and the total efficiency is about 62–79%, which are acceptable for practical application. For the desired upper bands of DCS/PCS/UMTS2100/LTE2300/2500 (1710–2690 MHz), the obtained antenna gain is 1.0–3.9 dBi

The antenna performance will be affected if some components closed to the proposed antenna. Hence, it is necessary to study the impacts of mounting the display upon the system ground and having some components (such as speaker and USB) in the no-ground portions. Fig. 14 shows the configuration of the proposed antenna with a display and two other components. The display with size 113 68 2 mm is directly mounted upon the system ground. The volumes of the speaker and USB are 8 18 3 mm and 6 8 3 mm , respectively. All these components are assumed to be perfect electric conductor (PEC) for the simulation. The simulated S parameter and total efficiency are shown in Fig. 15(a) and (b). Seen from Fig. 15(a), the primary effect of these components is deterioration in the impedance bandwidth of the original antenna, especially at lower frequencies. However, it still can provide sufficiently bandwidth to cover GSM850/900 operations. To further investigate the effect on antenna performance when adding some components, the simulated total efficiency is shown in Fig. 15(b). The total efficiency is defined as the ratio of radiated-to-stimulated power of the antenna. For the lower-band [see Fig. 15(b)], the simulated efficiency dropped by 2.5% on the average. For the upper-band, the difference between these two simulated efficiency is about 5.3% on the average. C. Effect of Hand The effects of the user's hand on the performance of proposed antenna have also been studied. Fig. 16 shows the human hand model directly gripping the handset at different position: at the top of the handset (Top position), in the middle of the handset

BAN et al.: DUAL-LOOP ANTENNA DESIGN FOR HEPTA-BAND WWAN/LTE METAL-RIMMED SMARTPHONE APPLICATIONS

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Fig. 13. Measured antenna gain and total efficiency (mismatching loss included) across the operating bands for the proposed antenna.

Fig. 14. Configuration of the proposed antenna with a display and two other components (a speaker and a USB).

(Middle position), and at the bottom of the handset (Bottom position). The relative permittivity and conductivity of the human hand model [28] is depended on the frequency. Hence, the relative permittivity and conductivity in the simulations are set as dispersive parameters whose values are obtained from [28]. The simulated S parameters and total efficiencies for these configurations are shown in Fig. 17(a) and (b), respectively. Seen from Fig. 17(a), it is evident that the matching bandwidths are widened for different gripping positions. This is mainly because the hand as a very lossy medium must absorbs a lot of power and then improves the bandwidth when touching the metal rim. The absorption losses with different hand-gripping positions are shown in Fig. 17(b). Compared to the case of the antenna without the hand, the simulated efficiencies drop from 75% to 23%–32% on the average in the lower-band. For the upper-band, the simulated efficiencies for different gripping positions are stabilized around 30% on the average. D. SAR Performance The SAR simulation model and the simulated SAR values for 1-g head tissues are shown in Fig. 18 and Table I, which are provided by CST version 2012. In the simulation, the smartphone handset is placed close to the head phantom ear with a distance of 1 mm and is inclined to the vertical line shown in the figure by 60 . The input power for the SAR testing is 24 dBm for 859 and 925 MHz and 21 dBm for 1795, 1920, 2045,

Fig. 15. Simulated results of the antenna with some components. (a) S parameter; (b) total efficiency (mismatching loss included).

Fig. 16. Configuration of hand grip smartphone at different positions (the metal rim is directly touched by the hand in each simulated mode).

2350 and 2595 MHz and the corresponding simulated 1-g SAR values and the S parameter all listed in Table I. It can be seen that the obtained SAR values for the proposed antenna are below the SAR limit of 1.6 W/kg for 1-g tissue which demonstrates that

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TABLE I SIMULATED SAR VALUES FOR 1-G HEAD ISSUE

TABLE II OF ANTENNA DESIGN IN THIS AND IN RECENT PUBLICATIONS

COMPARISON

PAPER

Fig. 17. Simulated results of hand grip at different positions. (a) S parameters; (b) total efficiency (mismatching loss included).

Fig. 18. Simulation model of SAR value for the proposed antenna.

the presented antenna is acceptable for practical smartphone application [25]–[27]. E. State-of-the-Art Comparison In order to identify the performance of the proposed antenna with respect to the state-of-the-art, the proposed antenna is com-

pared to designs presented in recent publications. Only antennas whose chassis were surrounded by the metal rim are chosen for a fair comparison. The key performances we care are the integrity of the metal rim, bandwidth and the total efficiency. Compared to [2], the proposed antenna can keep the integrity of the metal rim which is desirable for consumers. Notice that the proposed antenna's area is larger than the antenna in [3], [29] and [30]. However, the performances of the proposed antenna are still acceptable with respect to the wide bandwidth and total efficiency. Furthermore, the large no-ground portion in the proposed design can be reused which has been demonstrated in Part IV (B). Therefore, the presented comparison shows that the proposed antenna has good performance with respect to the integrity of the metal rim, bandwidth and total efficiency. V. CONCLUSION A simple direct-fed dual-loop antenna capable of providing hepta-band WWAN/LTE operation under surroundings of an

BAN et al.: DUAL-LOOP ANTENNA DESIGN FOR HEPTA-BAND WWAN/LTE METAL-RIMMED SMARTPHONE APPLICATIONS

unbroken metal rim in smartphone applications is proposed and studied in this paper. By combining the multiresonant character of the dual-loop antenna, the proposed antenna can provide two wide operating bands of 824–960 MHz and 1710–2690 MHz, respectively. A prototype of the proposed unbroken metal-rimmed antenna has been successfully designed, fabricated, and measured. The obtained measured results including S parameter, antenna peak gain, and total efficiency are presented, which can meet the requirements for smartphone systems. Furthermore, the simulated SAR values of the proposed antenna are less than the 1.6 W/kg for 1-g head tissue In addition, the greatest highlight is that the proposed antenna keeps the integrity of the metal rim very well, which is very promising for metal-rimmed smartphone applications. REFERENCES [1] J. Anguera et al., “Advances in antenna technology for wireless handheld devices,” Int. J. Antennas Propag., vol. 2013, 2013. [2] Q. X. Guo et al., “Interaction between internal antenna and external antenna of mobile phone and hand effect,” IEEE Trans. Antennas Propag., vol. 61, no. 2, pp. 862–870, Feb. 2013. [3] B. Yuan et al., “Slot antenna for metal-rimmed mobile handsets,” IEEE Antennas Wirel. Propag. Lett., vol. 11, pp. 1334–1337, 2012. [4] K. Ishimiya, C. Y. Chiu, and J. I. Takada, “Multiband loop handset antenna with less ground clearance,” IEEE Antennas Wirel. Propag. Lett., vol. 12, pp. 1444–1447, 2013. [5] Y. L. Ban et al., “Low-Profile narrow-frame antenna for seven-band WWAN/LTE smartphone applications,” IEEE Antennas Wirel. Propag. Lett., vol. 13, pp. 463–466, 2014. [6] H. Li, Z. T. Miers, and B. K. Lau, “Design of orthogonal MIMO handset antennas based on characteristic mode manipulation at frequency bands below 1 GHz,” IEEE Trans. Antennas Propag., vol. 62, no. 5, pp. 2756–2766, May 2014. [7] J. Ilvonen et al., “Design strategy for 4G handset antennas and a multiband hybrid antenna,” IEEE Trans. Antennas Propag., vol. 62, no. 4, pp. 1–1, Apr. 2014. [8] J. Anguera, A. Andújar, and C. Garc´ıa, “Multiband and small coplanar antenna system for wireless handheld devices,” IEEE Trans. Antennas Propag., vol. 61, no. 7, pp. 3782–3789, Jul. 2013. [9] R. Valkonen, M. Kaltiokallio, and C. Icheln, “Capacitive coupling element antennas for multi-standard mobile handsets,” IEEE Trans. Antennas Propag., vol. 61, no. 5, pp. 2783–2791, May 2013. [10] M. Zheng, H. Y. Wang, and Y. Hao, “Internal hexa-band folded monopole/dipole/loop antenna with four resonances for mobile device,” IEEE Trans. Antennas Propag., vol. 60, no. 6, pp. 2880–2885, Jun. 2012. [11] K. L. Wong and M. T. Chen, “Small-size LTE/WWAN printed loop antenna with an inductively coupled branch strip for bandwidth enhancement in the tablet computer,” IEEE Trans. Antennas Propag., vol. 61, no. 12, pp. 6144–6151, Dec. 2013. [12] Y. Li et al., “A compact hepta-band loop-inverted F reconfigurable antenna for mobile phone,” IEEE Trans. Antennas Propag., vol. 60, no. 1, pp. 389–392, Jan. 2012. [13] Y. Li et al., “Compact heptaband reconfigurable loop antenna for mobile handset,” IEEE Antennas Wirel. Propag. Lett., vol. 10, pp. 1162–1165, 2011. [14] C. H. Ku, H. W. Liu, and S. Y. Lin, “Folded dual-loop antenna for GSM/DCS/PCS/UMTS mobile handset applications,” IEEE Antennas Wirel. Propag. Lett., vol. 9, pp. 998–1001, 2010. [15] Y. L. Ban et al., “Small-size multiresonant octaband antenna for LTE/WWAN smartphone applications,” IEEE Antennas Wirel. Propag. Lett., vol. 13, pp. 619–622, 2014. [16] S. Wang and Z. W. Du, “A compact octaband printed antenna for mobile handsets,” IEEE Antennas Wirel. Propag. Lett., vol. 12, pp. 1347–1350, 2013. [17] S. C. Chen and K. L. Wong, “Small-size 11-band LTE/WWAN/WLAN internal mobile phone antenna,” Microw. Opt. Technol. Lett., vol. 52, no. 11, pp. 2603–2608, 2010. [18] C. K. Hsu and S. J. Chung, “Compact antenna with U-shaped open-end slot structure for multi-band handset applications,” IEEE Trans. Antennas Propag., vol. 62, no. 2, pp. 929–932, Feb. 2014.

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[19] Y. L. Ban et al., “Low-profile printed octaband LTE/WWAN mobile phone antenna using embedded parallel resonant structure,” IEEE Trans. Antennas Propag., vol. 61, no. 7, pp. 3889–3894, Jul. 2013. [20] Y. Li, Z. J. Zhang, and J. F. Zheng, “Compact heptaband reconfigurable loop antenna for mobile handset,” IEEE Antennas Wirel. Propag. Lett., vol. 10, pp. 1162–1165, 2011. [21] Y. L. Ban et al., “Small-size wideband monopole with distributed inductive strip for seven-band WWAN/LTE mobile pone,” IEEE Antennas Wirel. Propag. Lett., vol. 12, pp. 7–10, 2013. [22] J. H. Lu and Z. W. Lin, “Planar compact LTE/WWAN monopole antenna for table computer application,” IEEE Antennas Wirel. Propag. Lett., vol. 12, pp. 147–150, 2013. [23] C. M. Peng et al., “Bandwidth enhancement of internal antenna by using reactive loading for penta-band mobile handset application,” IEEE Trans. Antennas Propag., vol. 59, no. 5, pp. 1728–1733, May 2011. [24] Y. Cao, B. Yuan, and G. F. Wang, “A compact multiband open-ended slot antenna for mobile handsets,” IEEE Antennas Wirel. Propag. Lett., vol. 10, pp. 911–914, 2011. [25] Y. L. Ban, C. L. Liu, and L. W. Li, “Small-size coupled-fed antenna with two printed distributed inductors for seven-band WWAN/LTE mobile handset,” IEEE Trans. Antennas Propag., vol. 61, no. 11, pp. 5780–5784, Nov. 2013. [26] K. L. Wong et al., “Small-size internal eight-band LTE/WWAN mobile phone antenna with internal distributed LC matching circuit,” Microw. Opt. Technol. Lett., vol. 52, no. 10, pp. 2244–2250, 2010. [27] J. H. Lu and J. L. Guo, “Small-size octaband monopole antenna in an LTE/WWAN mobile phone,” IEEE Antennas Wirel. Propag. Lett., vol. 13, pp. 548–551, 2014. [28] C. Gabriel, “Tissue equivalent material for hand phantoms,” Phys. Med. Biol., vol. 52, no. 14, pp. 4205–4210, July 2007. [29] S. H. Kim et al., “A compact GPS and WLAN PIFA for full metalrimmed mobile handset using the ground bridges,” in Proc. Asia-Pacific Microwave Conf., 2012, pp. 648–650. [30] J. W. Zhong, K. K. Chen, and X. W. Sun, “A novel multi-band antenna for mobile phone with metal frame,” in Proc. 8th Int. Conf. Wireless Communications, Networking and Mobile Computing, Sep. 21–23, 2012, p. 1, 4.

Yong-Ling Ban was born in Henan, China. He received the B.S. degree in mathematics from Shandong University, China, the M.S. degree in electromagnetics from Peking University, China, and the Ph.D. degree in microwave engineering from the University of Electronic Science and Technology of China (UESTC), Chengdou, Sichuan, China, in 2000, 2003, and 2006, respectively. In July of 2006, he joined the Xi'an Mechanical and Electric Information Institute (from China North Industries Group Corporation) as a microwave engineer. He then joined Huawei Technologies Co., Ltd., Shenzhen, China, first as a RF antenna design engineer and then as senior design engineer. At Huawei, he designed and implemented various terminal antennas for 15 data card and mobile phone products customized from leading telecommunication industries like Vodafone. Since September 2010, he has been an Associate Professor of microwave engineering with UESTC. His research interests include wideband small antennas for 3G/LTE handheld devices, MIMO antenna decoupling techniques, and smart antennas for wireless AP. He is the author of over 40 referred journal and conference papers on these topics. Prof. Ban holds 15 granted and pending Chinese and overseas patents.

Yun-Fei Qiang was born in Anhui, China, in 1990. He received the B.S. degree in applied physics from the Hefei University of Technology, Heifei, China, in 2012. He is currently pursuing the M.S. degree at the University of Electronic Science and Technology of China (UESTC), Chengdou, Sichuan, China. His main research interests are multiband smartphone antennas for wireless communications, especially for narrow-frame antennas and metal-rimmed antennas designs for smartphone applications, as well as base station antennas and antenna synthesis.

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IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 63, NO. 1, JANUARY 2015

Zhi Chen received the B.Eng., M.Eng., and Ph.D. degrees in electrical engineering from the University of Electronic Science and Technology of China (UESTC), Chengdou, Sichuan, China, in 1997, 2000, 2006, respectively. In April 2006, he joined the National Key Lab of Science and Technology on Communications (NCL), UESTC, where he was a Professor. He was a visiting scholar at the University of California, Riverside, CA, USA, during 2010 and 2011. His current research interests include wireless communication and signal processing, microwave communication, and THz communication. Dr. Chen has served as a reviewer for various international journals and conferences, including the IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, the IEEE TRANSACTIONS ON SIGNAL PROCESSING, etc.

Kai Kang was born in 1979. He received the B.Eng. degree in electrical engineering from the Northwestern Polytechnical University, China, in 2002, and the joint Ph.D. degree from the National University of Singapore, Singapore, and Ecole Supérieure D'électricité, France, in 2008. From 2006 to 2010, he was a Senior Research Engineer at the Institute of Microelectronics, A*STAR, Singapore. From 2009 to 2010, he was an Adjunct Assistant Professor at the National University of Singapore. From 2010 to 2011, he was a Principle Engineer at Globalfoundries. Since June 2011, he has been with the University of Electronic Science and Technology of China (UESTC), Chengdou, Sichuan, China, where he is now Professor and Vice Dean of the School of Electronic Engineering. His research interests are RF and mm-wave integrated circuits design and modeling of on-chip devices.

Jin-Hong Guo received the B.E. degree in electronic engineering in 2010 from the University of Electronic Science and Technology of China (UESTC) and the Ph.D. degree in 2014 in biomedical engineering from Nanyang Technological University (NTU), China. Dr. Jinhong was a Research Engineer for RFIC design at VIRTUS Integrated Circuit Design Lab, NTU, from March 2011 to August 2011. From August 2011 to March 2014, he was a Research Assistant with the Applied Microfluidicological Lab jointly with the Institute of Microelectronics, A*STAR, Singapore. In April 2014, he co-founded the JESON Medical Co., Ltd., where he is now the Chief Scientist. Dr. Jinhong has published many patents which have been commercialized by JESON. He has authored more than 40 journals and international conference papers. His research interests include micro/nano solid-state biosensors, electrofluidics, acoustofluidics, optofluidics sensors and actuators, nano-electromagnetics, and microwave cytometry. He also serves as a reviewer for many reputable journals, including various IEEE TRANSACTIONS, as well as Lab On a Chip, Biomicrofluidics, Biomedical Microdevice, Microfluidics and Nanofluidics, Electrophoresis, Progress In Electromagnetic Research, etc.