A New Interleaved Active-Clamp Forward Converter with Parallel Input

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conduction loss and switch loss. [8] proposes a two active-clamp forward converter with a single transformer and a full-wave rectifier structure, only two rectifier ...

A New Interleaved Active-Clamp Forward Converter with Parallel Input and Series-Parallel Output Guoxing Zhang, Xinke Wu, Wei Yuan, Junming Zhang, Zhaoming Qian (IEEE Senior Member) College of Electrical Engineering, Zhejiang University Zheda Road 38#, Hangzhou China [email protected] Abstract-A new interleaved active-clamp forward converter with parallel input and series-parallel output is proposed in this paper. With the proposed topology the secondary windings of the transformers work in series when the input voltage is low but in parallel when the input voltage is high. Therefore, the duty cycle is not limited within 0.5 and the steady state conversion gain of the converter can be improved. Accordingly, the current stress and the conduction loss of the primary side are reduced. Interleaved operation can reduce the current ripple in the output capacitor. A prototype with 36-72V input and 12V/20A output is built up to verify the theoretical analysis.

I.

in the filter is identical. But the one-choke approach is less efficient than the two-choke approach because of the higher conduction loss and switch loss. [8] proposes a two active-clamp forward converter with a single transformer and a full-wave rectifier structure, only two rectifier diodes are needed. This paper proposes a new interleaved active-clamp forward topology with a series-parallel rectifier structure. The proposed topology employs an extra diode to connect the secondary sides of the transformers in series based on the one-choke approach. Thus, the two interleaved channels can operate with duty cycle more than 0.5. Compared with the conventional interleaved active-clamp forward converter with a common LC output filter, the proposed converter has the following advantages. a. With the enlarging of maximum duty cycle, the conversion gain at low input voltage is improved, which is suitable for the wide input voltage applications. b. Due to the increase of the turn ratio, conduction loss and current stress in the primary sides are reduced. c. The current ripple through the output capacitor can be reduced. d. Lower voltage stress across the rectifier diodes.

INTRODUCTION

With the rapid development of the industry electronics, forward converter with its simplicity and high efficiency is widely used [1]. In order to keep the transformer balance, many magnetic reset methods have been proposed. In a conventional forward converter, an auxiliary winding in series with a diode is required to reset the transformer core. A snubber capacitor is used to reset the core in the resonant forward converter, and the voltage stress across the switches is too high. The RCD clamp method has been proposed and analyzed to reduce the voltage stress of the switch devices [2-3]. However, the energy stored in the magnetizing inductor is dissipated on the resistor and the conversion efficiency is limited. The active clamp technique [4-6] can reduce the voltage stress and realize ZVS soft switching, thus it improves the conversion efficiency. For the request of lower voltage, higher current and smaller size, various interleaved forward converters have been proposed [7-12]. The interleaving of topologies can be implemented in many ways. One method is paralleling the outputs of two converter modules directly with a common output filter capacitor, which can reduce the output current ripple. However, current sharing could be a problem, and the maximum duty cycle is limited to 0.5. The topology with two paralleled modules at the input of a common LC output filter operates quite different from the converter with two separate output-filter inductors [7]. With the same output current ripple, the output filters in the two converters could be designed at the same size due to the energy stored

978-1-422-2812-0/09/$25.00 ©2009 IEEE

II. OPERATION PRINCIPLE The conventional interleaved active-clamp forward converter is shown in Fig. 1. The two channels of the converter are connected in parallel on the primary side and share the freewheeling diode and the output LC filter on the secondary side. With the duty cycle less than 0.5, the two interleaved channels operate in parallel. Figure 2 shows the proposed two interleaved active-clamp forward topology with a new rectifier structure. Compared with the conventional rectifier structure, the new topology has an extra diode which connects the secondary side of the transformers in series. Thus the two interleaved channels can operate with duty cycle more than 0.5. According to the operation of the rectifier structure, the operating process of the converter can be divided into two cases. In case 1, the duty cycle is less than 0.5, and the converter operates just as the conventional

40

converter, which will not be elaborated here. In Case 2 with the duty cycle more than 0.5, the two interleaved channels work in series or parallel, the key waveforms at steady state are given in Fig. 3. The detailed operation of each mode is presented as follows. Lo

Lf1 Cc1 Vin

Lm1

Q2

Mode 1 (t0-t1): In this mode, Q1 and Q3 turn on, so the voltage across transformer primary windings is in the same direction. It results in that the secondary sides of the two channels work in series. The output current iLo flows through D4 and Lo and increases linearly. The voltage across D1, D2 is Vin/n, and the voltage across D3 is 2Vin/n. Mode 2 (t1-t2): When Q1 turns off at t1, the primary current of T1 charges the resonant capacitor Cr1 and the resonance among Lf1, Lm1, Cr1 occurs. The rectifier diode D4 keeps on and the others stay off. Thus the voltage across D1 declines until it falls to zero at t2. Meanwhile the voltage across D3 declines from 2Vin/n to Vin/n. Mode 3 (t2-t3): When the voltage across Cr1 reaches Vin at t2, the rectifier diode D4 and D1 both are conducting. So in this mode the primary voltage of T1 remains zero and the magnetizing current ILm1 remains constant. The resonance occurs among Lf1 and Cr1. Mode 4 (t3-t4): At time t=t3, the voltage across Cr1 increases to Vin+VCc1, and the primary current of T1 circulates through the intrinsic diode of auxiliary switch Q2. Mode 5 (t4-t5): The auxiliary switch Q2 is turned on at time t=t4. iD4 decreases to zero and iD1 increases through ILo at time t=t5. The state that T1 is clamped at zero is over at the end of this mode. Mode 6 (t5-t6): In this mode, the primary voltage of T1 is negative and the power transferred to the load through transformer T2. The magnetizing current iLm1 starts to decrease from positive to negative. The voltage across D4 is VCc1/n and the voltage across D2 is (Vin+ VCc1)/n. Mode 7 (t6-t7): When the auxiliary switch is turned off at t6, the primary voltage of T1 is still negative and the resonance capacitor Cr1 begins to discharge to Lf1 and Lm1. the resonance among Lf1, Lm1, Cr1 occurs. In this mode, the voltage across Cr1 decreases from Vin+VCc1 to Vin. Meanwhile VD2 falls from Vin/n to zero and VD4 falls from (Vin+VCc1)/n to Vin/n. Mode 8 (t7-t8): The capacitor voltage VCc1 falls to Vin at time t=t7, and the voltage across the primary winding of T1 is clamped to zero. Thus the magnetizing current ilm1 keeps constant. The resonance between Lf1 and Cr1 occurs. The second side current commutation begins at t7 between D1 and D4. Mode 9 (t8-t9): In this mode, the switch Q1 is turned on. The rectifier diode D1 and D4 both are conducting. Mode 10 (t9-t10): When the current flowed through D2 decreases to zero at the beginning of this mode, the commutation of second side is ended. The primary voltage of T1 is Vin, thus the voltage across D1 is Vin/n and across D3 is 2Vin/n. This mode is the same as mode 1. Since the two interleaved channels operate symmetrically, the operation of next half cycle is similar to the modes described above. But during the next half cycle, the roles of the two channels are exchanged.

T1

·

+

ILo

·

Ro V o

Co

D3

-

Q1

Cr1 D1

IQ1

D2

Lf2 Cc2 Lm2

Q4

·

T2

·

Cr2

Q3 IQ3

Fig.1 Conventional interleaved forward converter Lo

Lf1 Cc1 Vin

Lm1

Q2

·

T1

+

ILo

·

D3

Co

Ro Vo -

Q1

Cr1 D4

IQ1

D1

D2

Lf2 Cc2 Lm2

Q4

Q3

·

T2

·

Cr2

IQ3

Fig.2 Proposed interleaved forward converter

Vg1 Vg2 Vg3 Vg4 V

in

1− D

VQ1 V

Vin in

1− D

VQ3 VV

in

in

V

n(1 − D ) Vin D1 Vn in n(1 − D ) Vin n

VD2 2V

in

VD3

n Vin n

Vin D n(1 − D)

VD4 IQ1 IQ3 ILm1 ILm2 t0

t1 t 2 t3 t4 t5

t6 t7 t8 t9 t10

Fig.3 Key waveforms of the proposed converter in case 2

978-1-422-2812-0/09/$25.00 ©2009 IEEE

41

Lo

Lf1 Cc1 Vin

Lm1

Q2

·

Q1

T1

D3

Co

Ro

Q2

Vin

D1

Lm2

·

T2

Cc2

·

Lm1

·

T1

·

D3

Co

Ro

Lm2

·

·

Lm1

Q1

D1

T1

·

Ro

Co

D3

D4

D1

T2

Cc2

·

Cr2

D2

n=

Lo ·

Q1

T1

D3

Co

Ro

Vin

·

Lm1

Q2

T1

Q1

Cr1 D4

D1

·

Ro

Co

D3

Cr1 D4

D2

Lf2

D1

Lm2

·

T2

Cc2

·

Cr2

Q3

·

Q1

T1

D3

Co

Ro

·

Lm1

Q2

Vin

T1

Q1

Cr1 D4

D1

·

o

Co

D3

Ro

D4

D1

D2

Lf2

Cc2 Lm2

·

T2

Cc2

·

Cr2

Q3

·

Lm2

Q4

T2

·

Cr2

Q3

Mode 7

Mode 8

ΔI L = o

Q2

·

Q1

T1

Cc1

·

D3

Co

Ro

Vin

Q2

D1

T1

·

D3

Co

D4

D1

D2

Lf2

Cc2 Lm2

·

T2

·

Cr2

Mode 9

Cc2 Q4

Lm2

Q3

·

T2

Vo + V f

Vo (0.5 − D)

(3)

Lo f s

(Vo −

Vin

)(1 − D) n Lo f s

(4)

Figure 5 shows the current ripple with the same output-filter inductance (Lo=10μH), where it is in function of the input voltage. Obviously the current ripple of the proposed converter is much lower than that of the conventional interleaved forward.

Ro

Cr1

D2

Lf2

Q3

·

Q1

Cr1 D4

Q4

Lm1

·

Cr2

Mode 10

ΔILo(A)

Vin

Lm1

Lo

Lf1

Lo

Lf1 Cc1

(2)

Vin _ min Dmax

Since the proposed converter has two operating cases according to the different duty cycle, it should be analysed separately. In case 1, the expression of current ripple in the output-filter inductor is the same as the conventional converter as given in (3). But for case 2(when D>0.5), the current ripple can be calculated by

Cr1

D2

Lf2

ΔI L =

Lo

Lf1 Cc1

·

Ns

=

B. Current ripple comparison For the conventional converter, the frequency of the output filter is twice the switch frequency, so the peak to peak current ripple in the output-filter inductor is given in

·

Mode 6 Lo

Lf1 Lm1

T2

Cr2

Q3

Mode 5 Cc1

·

Lm2

Q4

NP

D2

Lf2

Cc2

(1)

n

where Vf is the forward voltage drop of the rectifier diode. With a bigger Dmax, the turn ratio of the proposed converter can be larger than the conventional converter, which helps to reduce the primary side currents of the proposed converters. Thus the primary conduction loss and current stress of the proposed converter is lower than the conventional converter.

Lo

Lf1 Cc1

·

2D

·

Mode 4

Lf1 Lm1

T2

Cr2

Q3

Mode 3 Cc1

·

Lm2

=

where D is the duty cycle of the main switch and n is the turn ratio of the transformer. And the turn ratio here is given as:

Cr1

D2

Q4

Q3

Q4

Vo Vin

Lf2

Cc2

Q2

·

Lo

Q2

Vin

Cr1 D4

Vin

T2

Lf1 Cc1

Lf2

Q4

D2

Mode 2 Lo

Q1

Vin

D1

Cr2

Q3

Lf1 Cc1

·

Lm2

Q4

Mode 1

Q2

Cr1 D4

Cr2

Q3

Q4

Ro

Co

D3

Lf2

Cc2

Q2

·

D2

Lf2

Vin

T1

Q1 D4

Q4

·

Lm1

Cr1

proposed rectifier structure here the maximum duty cycle can be larger than 0.5. Thus the utilization of each channel can be improved. From the analysis above, the conversion gain according to different duty cycle in overall input voltage range can be expressed as:

Lo

Lf1 Cc1

·

III. OPTIMAL DESIGN CONSIDERATIONS A. The duty cycle and conversion gain It is known that the conventional interleaved active-clamp forward works with the duty cycle less than 0.5 because of the current sharing problem. But with the

978-1-422-2812-0/09/$25.00 ©2009 IEEE

Conventional Proposed

Vin(V) Fig.5 Current ripple with same filter inductance

42

The filter inductor can be designed according to (3) and (4). With a same maximum current ripple about 15% of the full load current, the filter inductor of the proposed converter can be much smaller than the conventional converter.

C. The voltage stress across the rectifier diodes The voltage stress of the rectifier diodes in the proposed converter are given in (5) to (7) respectively. VD = VD = 1

2

Vin n(1 − D)

⎧ Vin Vo ⎪⎪ n = 2 D VD = ⎨ ⎪ 2Vin = Vo ⎪⎩ n D

=

Vo 2 D(1 − D)

(5)

TableⅠ 0 < D ≤ 0.5

(6)

COMPARISON OF KEY PARAMETERS BETWEEN CONVENTIONAL AND PROPOSED CONVERTER

3

0.5 < D < 1

Conventional forward Q1,Q2

Vo ⎧ 2Vin D ⎪⎪ n(1 − D ) = 1 − D VD = ⎨ ⎪ Vin D = Vo ⎪⎩ n(1 − D ) 2(1 − D ) 4

0 < D ≤ 0.5

T1,T2

0.5 < D < 1

For the conventional converter, the voltage variation is just as case 1 (0

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