A NEW TYPE OF MONOLITHIC BROADBAND FREQUENCY DOUBLER BY USING AN ARRAY OF PSEUDOMORPHIC HEMTs Stefan Simion Military Technical Academy Regina Maria 81-83, Bucharest, Romania Phone : +40.1.3354660/192, Fax : +40.1.3355763, e-rnail :
[email protected]
SUMMARY In this paper, it is proposed a new type of monolithic frequency doubler, consisting of an array of pseudomorphic HEMTs, separated by inductances. As equivalent circuit, the doubler contains two nonlinear lines, one for the gate and one for the drain, which are coupled by the gate to drain capacitance. It is shown that the frequency bandwidth of this type of doubler is much larger compared to those obtain by other means. For the example presented in this paper, the +I-3 dB frequency bandwidth exceeds three octave with a conversion eficiency of -30%.
1. INTRODUCTION
equivalent circuit, the following expression have been used for a 200pm x 0.35p.m HEMT [4] :
The microwave broadband frequency multipliers are extensively used in many applications such as satellite communications, radar systems, microwave instrumentation and so on. As the frequency increases, these circuits have to be designed as distributed components. So, new types of frequency doublers have been designed and monolithically integrated, based on soliton generation on nonlinear transmission lines [I]. These types of frequency doublers consist of an array of reverse biased Schottky varactor diodes, separated by inductances, which is essentially a passive circuit. In order to improve the conversion efficiency and also the frequency bandwidth, the circuit should contain active devices. Distributed amplifiers, having very large frequency bandwidth, operating into the small signal regime, consisting of an array of FETs and HEMTs, have been realised [2],[3]. As equivalent circuit, these types of amplifiers consist of two lines, one for the gate and one for the drain, coupled by the equivalent capacitance between the gate and the drain. In this paper, a nonlinear analysis of a distributed HEMT amplifiers is performed by a computer simulation method. For the HEMT, it is used a nonlinear model, developed and experimentally verified in [4]. The aim of this paper is to analyse the possibility to use this type of circuit as a frequency doubler.
2. CIRCUIT DESCRIPTION The equivalent circuit of the pseudomorphic HEMT is given in Fig.1, where the parasitic elements have not been included. For the elements of this
Cgs = Cgso[l+ tanh(Vgs - 0.04SVis)].
(1)
[1+ tanh(0.4Vds)]
where
V,, is the voltage across the nonlinear capacitance C,,, Vdsis the voltage between the drain and the source, I p k is the drain-source current for Vd,=O and Vg,=0.55V and C,, and Cgd, are the gate-source and the gatedrain capacitances for V,=Vd,=0. Using (1)-(4), in Fig.2, the variations of the Cgs/Cgso,Cgd/Cgdo,Ids/Ipk and Rds (for IPk=26.3mA)are graphically drawn versus the V,, and Vds. In this paper, the equivalent circuit shown in Fig. 1 will be used to analyse a distributed frequency doubler, realised by using a number n of cells, each cell consisting of a small width pseudomorphic HEMT device, connected be inductances, as shown in Fig.3.
0-7803-5 139-8/99/$10.00 0 1999 IEEE
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Gate
Source
Drain
where w[pm] is the gate width of the lumped HEMT for one cell of the circuit shown in Fig.3.
Source
From Fig.3, it is observed that the equivalent circuit of the distributed frequency doubler consists of two nonlinear lines : the nonlinear gate line (much similar with the wellknown nonlinear transmission line [l]) and the nonlinear drain line. These lines are not completely separated, the capacitance Cgd connect each other, periodically, at every cell.
Fig.1 : The nonlinnear equivalent circuit of the HEMT
In order to compute the elements of the each cell, knowing Cgso, Cgdo, R,, Ipk, Cds for gate width of 200ym, the following formulas may be used :
The inductances which connect these small HEMTs are L, and Ld for the gate and the drain line respectively. The losses of these inductances by the skin effect are RLgand RM,respectively.
0
3000
-
E 2000 '
S
0
A
rn
2 I000
I
2
0
E3~2.2: The 3D representation of the cgs!cgso, Cgd/Cgdo,Ids/Ipk and Rds (for IPk=26.3mA) , versus the V,, and Vds. '2
288
I
dmcLd
Fig.3 : The equivalent circuit of the distributed frequency doubler, realised by using a number of pseudomorphic HEMTs
-
3. DESIGN AND NUMERICAL RESULTS
Cgs,echiv =
-
The circuit shown in Fig.3 has been analysed by computer simulation using a noncommercial software. The analysis has been performed in the time domain, applying a finite difference method on the differential equations which describe the circuit. The design of the circuit has been performed, taking into account that for a good conversion efficiency of the frequency doubler, the loads of the gate and drain lines, Rg,loadand Rd,load,respectively, and also Rg,biasand &bias should be matched with respect to the characteristic impedance of the gate and drain lines. For this aim, the following relations have been used
-
c g s ( v g s = Vg,bias I 2 , v d s = Vd,bias 12) -
Cgd,i = Cgd,echiv . (1 -tg,
-
-
Cgd,o = Cgd,echiv
-
'
(1 f g,
.R ) '
R) / ( g m R )
Cgd,echiv = CgS (vgs = Vgbias 1 2, v d s = Vd,bias
2,
In the relations written above, g,,, is the transconductance and R=5OQ. The voltage Vg.biasand Vd,bias have been choosen to maximize the nonlinearity of the gate line, the drainsource resistance and the drain-source current (see Fig.2). The elements of the circuit have been computed for n=15, a gate width w=20pm, a characteristic impedances of the gate and drain lines equals to 5OQ, by using the relations (1)-(12). Therefore, the circuit has been analysed by computer
-
simulation for the following input data : L g = Rd,load = Rd,bias =
(12)
-
_
L d =O. lnH,
C L g = CLd = 7 F ,
-
-
-
-
-
-
R d =3Q,
Cgso=
Cgdo =15fF, c d s =20fFF,Ipk =2.72mA, Rg,bias=&bias where : -
-
Cg,echiv = Cgs,echiv
-
f
Cd,echiv = Cds -k F L d
CLg
.f
f
Cgd,o
Cgd,i
--Rg,load=Rd,load=5OQ ,
v g ,bias=().4 v vd.bi as= 1 5 V Vg= VamPlsin(2nft),Va,,~=0.6V. In Fig.4 is shown the representation of the normalized output power on the second harmonic, Pz,
289
7
3
versus the input frequency (the values were normalized to P2 for an input frequency of 20 GHz). From this figure it is observed that the bandwidth of this type of frequency doubler is much larger compared to those obtained by using other types of circuits or nonlinear transmission lines. For this example, the attainable bandwidth is about SOGHz, exceeding three octave, for output power levels varying with +I-3dB around the value for P2 at 20 GHz. Also, the conversion efficiency, computed for an input frequency of 2SGHz, as the ratio between the output power P2 and the input power from the microwave generator (in this example this power is equal to 0.9mW) is -30%.
0.60
-0.10
0.50 0:
:
-5
-10 :
Vamp,= 0.6 V
V X
0 F
-15 r L
. =0.4V
g.bias
'd.bias
= l5
"
1
I 0 10 20 30 40 50 60 70
-20
- 0.20
Input frequency [GHz] Fig.4 :
550
The normalized output power on the second harmonic, versus the frequency
5.OO/di v. T i m e Cpsl
600
(b) 7.10
In Fig.5 are shown the waveforms to the input of the gate line, to the output of the gate line and to the output of the circuit (on the load resistance of the drain line), for an input frequency f=45GHz. It is observed the formation of the solitons on the gate and drain line.
4. CONCLUSIONS In this paper, the performances of a new possible frequency doubler have been evaluated by computer simulation. The results of these simulations shown that the bandwidth of the frequency doubler, consisting of a number of HEMTs operating in the nonlinear regim, is larger compared to the results obtained by other types of frequency doubler. These preliminary results will be continued on the optimization of the performances, in order to increase the conversion efficiency.
550
5.0OAdiv.
600
T i m e Cpsl (c) Fig.5 : The waveforms to the input of the gate line (a), to the output of the gate line (b) and to the output of the circuit, on the drain line (c)
REFERENCES [ I ] E. Carman et al, IEEE Microwave Guided Wave Letters, vol.1, no.2, February 1991, pp.28-31. [2] Y. Imai et al, IEEE Microwave Guided Wave Letters, vol.6, no. 10, October 1996, pp.357-359. [3] J. Pus1 et al, IEEE MTT-S Digest, 1995, pp.16611664.
L41 I. et al, IEEE Transactions cm Microwave Theory and Techniques, vo1.40, no.12, l9g21 PP.2258-2266.
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