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A Novel Dual-Inductor Boost Converter With Ripple Cancellation for High-Voltage-Gain Applications Ching-Shan Leu, Member, IEEE, Pin-Yu Huang, Student Member, IEEE, and Ming-Hui Li

Abstract—A new full-wave rectifier is proposed in this paper. By integrating the proposed rectifier with a classic current-fed converter as an example, a dual-inductor boost converter with ripple cancellation is presented. It has several features, such as high voltage gain with smaller transformer turns ratio, recovery of the transformer secondary leakage energy, low voltage stress on the rectifier diodes, and nonpulsating input and output currents. These properties make it desirable for high-frequency highefficiency high-voltage-gain applications, such as the renewable energy source power system. In addition to the circuit analysis and design, a 150-kHz 24–36-V-input 200-V/600-W-output converter prototype is implemented and tested to demonstrate its feasibility. Index Terms—Current-fed boost converter, current ripple cancellation, high voltage gain, lossless snubber, renewable energy.

I. I NTRODUCTION

I

N ADDITION TO experiencing a supply shortage in recent years, fossil fuel causes serious environmental pollution problems. Therefore, developing a high-efficiency renewable energy power system, as shown in Fig. 1, has become an urgent issue of concern. Due to wide-range low dc output voltage, the renewable energy sources (solar cell as an example) cannot directly support the ac or dc electrical appliances. Consequently, a step-up converter is required to provide high voltage gain from low dc output voltage to generate 200-V high dc voltage as the input voltage of the cascaded dc–ac inverter or dc-dc converter. Several step-up converter topologies, classified with the voltage-fed configuration and current-fed configuration, have been proposed in the literature [1]–[17]. By employing the voltage-fed configuration, however, a large number of input electrolyte capacitor is required to meet the severe ripple current characteristic of a renewable energy source. In addition, the required voltage boosting function is only realized by using a transformer with large turns ratio. As a result, large parasitic capacitances and leakage inductance are induced [18], [19]. These parasitic components generate high voltage and high current spikes on the power devices. Therefore, a voltage-fed converter is not suitable for high-voltage-gain applications. Manuscript received July 31, 2009; revised December 11, 2009 and March 10, 2010; accepted March 22, 2010. Date of publication April 29, 2010; date of current version March 11, 2011. C.-S. Leu and P.-Y. Huang are with the Department of Electrical Engineering, National Taiwan University of Science and Technology, Taipei 10673, Taiwan (e-mail: [email protected]; [email protected]). M.-H. Li was with the Department of Electrical Engineering, National Taiwan University of Science and Technology, Taipei 10673, Taiwan. He is now with the Skynet Electronics Company, Ltd., Taipei 11570, Taiwan (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TIE.2010.2048835

Fig. 1.

Block diagram of a renewable energy power conversion system.

On the contrary, the current-fed configuration has a continuous input current. Thus, the number of input electrolytic capacitor can be minimized. Moreover, part of the voltage gain can be provided by the current-fed circuit itself; a smaller turns ratio transformer can be used, and the problems caused by the transformer parasitic components are alleviated. However, a full-wave rectifier and filter stages are still needed to obtain the required dc output voltage. Employing a centertapped rectifier as an example, the rectifier diodes and the output filter capacitor suffer from high voltage spike and high current ripple, respectively. In addition to utilizing a highvoltage-rating rectifier diode or a turn-off RC snubber circuit, a large number of the output filter capacitor has to be used. Thus, the converter efficiency and power density performance are degraded. To enhance the converter performance, therefore, a new fullwave rectifier is proposed [20], and a novel current-fed dualinductor boost converter with ripple cancellation (DIBCRC) as an application example is presented in this paper. Several features, such as high voltage gain with smaller transformer turns ratio, recovery of the transformer secondary leakage energy, low voltage stress on the rectifier diode, and continuous input and output currents, can be obtained. These properties make the converter desirable for high-voltage-gain applications, such as a solar-cell-powered system. In addition to the circuit analysis and design, a 150-kHz 24–36-V-input 200-V/600-W output converter prototype is implemented and tested to demonstrate its feasibility. II. A NALYSIS AND C IRCUIT O PERATION The circuit diagram of the DIBCRC is shown in Fig. 2. It comprises two input inductors L1 and L2 , one transformer Tr with one primary winding P1 and four identical secondary windings S1 −S4 , two switch pairs Q1 −Q3 and Q2 −Q4 , three clamping capacitors C1 , C2 , and C3 , one output capacitor Co ,

0278-0046/$26.00 © 2010 IEEE

LEU et al.: NOVEL DIBCRC FOR HIGH-VOLTAGE-GAIN APPLICATIONS

Fig. 2.

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Circuit diagram of the DIBCRC.

Fig. 4. Equivalent circuits for different operation stages of the DIBCRC.

Fig. 3.

Key waveforms of the DIBCRC.

and two series-connected diode pairs D1 −D2 and D3 −D4 . The key waveforms of the DIBCRC are shown in Fig. 3. To simplify the analysis of the presented converter, several assumptions are made. All semiconductors are ideal. Moreover, all capacitors C1 , C2 , C3 , and Co are sufficiently large that the voltage across each capacitor is constant during one switching period. The four leakage inductances Lk1 −Lk4 are identical. There are six operation stages within one switching cycle during the steady-state operation. However, only three

of them need to be described since the remaining stages are analogous. The equivalent circuit of each stage during a half period is shown in Fig. 4(a)–(c), and the operation principles are analyzed as follows. Stage 1 [t0 −t1 ] VGS2 is provided to turn on Q2 at t0 . Both Q1 and Q2 are turned on during this time interval t0 − t1 , i.e., Tcharge . The voltage across the transformer primary is thus shorted, and two inductor currents iL1 and iL2 increase linearly. Without being forward biased, the four rectifier diodes D1 , D2 , D3 , and D4 are all turned off. Part of the load currents are thus provided by the clamping capacitors C1 and C2 through

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(+)

(−)

C1 −S1 −Lk1 −R−S4 −Lk4 −C1 (−) S2 −Lk2 −C2 , respectively.

(+)

and C2 −S3 −Lk3 −R−

Stage 2 [t1 −t2 ] At t1 , Q1 is turned off, and the body diode of Q3 is turned on. The inductor current iL2 increases continuously as (1), and the inductor current iL1 decreases linearly as (2) Vi (t − t0 ) L2 Vc−Vi iL1 (t) = iL1 (t1 ) − (t − t1 ). L1

iL2 (t) = iL2 (t0 ) +

(1) (2)

The voltage across the transformer primary winding P1 is clamped to VC3 , which is the sum of the input voltage and the inductor voltage vL1 . The input power is transferred to the load via transformer windings S1 and S2 during the time interval t1 −t3 , i.e., Ttransfer . In addition to providing the load current, part of the input power is used to charge the clamping capacitors C1 and C2 through S4 −S2 −Lk2 −D2 −D1 −C1 −Lk4 −S4 and S3 −C2 −D2 −D1 −S1 −Lk1 −Lk3 −S3 , respectively. The average voltage across C1 or C2 is thus clamped to VO because the voltages across S1 −S4 or S2 −S3 cancelled each other. Moreover, the voltage stresses of D3 and D4 are equal to VO due to the turning on of D1 and D2 . At t2 , the current through Q3 reaches zero and starts to reverse. Q3 should be turned on before t2 so that zero-voltage switching operation can be achieved.

Fig. 5.

Key current waveforms of the rectifier circuit.

Fig. 6.

Current ripple distribution during the t1 −t2 time interval.

Stage 3 [t2 −t3 ] During this time interval, the operation is identical to that in [t1 −t2 ] except that the current through Q3 is negative until Q1 is turned on at t3 . The operations of the following stages during another half period are analogous. At t0 , Q2 is turned on again to start another switching cycle. As mentioned, the proposed rectifier circuit has an output current ripple cancellation mechanism so that the rectifier output current Io1 is a continuous waveform. Several key current waveforms are thus shown in Fig. 5 to explore the criteria. Accordingly, (3) has to be met ΔIs1 + ΔIs3 = 0

ΔIs2 + ΔIs4 = 0.

(3)

On the other hand, the current ripples during switch tran+ sients, for instance t− 1 to t1 , are shown in Fig. 6(a) and (b), and the diode current ripple ΔID can be derived as ΔID = |ΔIs1 | + |ΔIs4 | = |ΔIs2 | + |ΔIs3 |.

(4)

To simplify the analysis, the equivalent series resistance of the clamping capacitors and output filter capacitor can be ignored without making significant deviations. Consequently, the current ripple distribution among the secondary windings S1 , S2 , S3 , and S4 can be calculated as follows: Lk4 |ΔIs1 | = ΔID Lk1 + Lk4 Lk3 |ΔIs2 | = ΔID Lk2 + Lk3

(5) (6)

Lk2 Lk2 + Lk3 Lk1 |ΔIs4 | = ΔID . Lk1 + Lk4

|ΔIs3 | = ΔID

(7) (8)

Applying (5)–(8) into (3), the criteria to achieve a continuous rectifier output current can be obtained as Lk1 Lk4 = . Lk3 Lk2

(9)

Practically, one of the conditions, namely, LK1 = LK4 and LK2 = LK3 , can be approximately realized by using the sandwich winding scheme in the transformer construction. Moreover, the currents Ia , Ib , and Ic labeled in Fig. 5 can be calculated by applying the ampere–second balance

LEU et al.: NOVEL DIBCRC FOR HIGH-VOLTAGE-GAIN APPLICATIONS

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on the clamping capacitor under Lk1 = Lk4 and Lk2 = Lk3 assumptions Io 2 Io (2 · D − 1) Ib = 4(1 − D)

Ia =

Ic =

Io (2D − 3) . 4(D − 1)

(10) (11) (12)

+ − − During t+ 0 < t < t1 or t2 < t < t3 time interval, the output current Io1 is calculated as

Io1 = 2 · Ia = Io .

TABLE I M AIN PARAMETERS OF DIBCRC

voltage Vo . Thus, the clamping capacitors C1 and C2 can be expressed as

(13) C1,2 >

+ − − During t+ 1 < t < t2 or t3 < t < t0 time interval, the output current Io1 is calculated as

Io1 = Ic − Ib = Io .

(14)

Therefore, the current Io1 is equal to the load current Io all the time. Ideally, the output capacitor Co can be eliminated. III. D ESIGN C ONSIDERATIONS It can be seen from the previous descriptions that there are two phases, namely, Tcharge and Ttransfer , within each half of the switching cycle. The Tcharge and Ttransfer time intervals are given as follows: 1 Tcharge = D − (15) · Ts 2 Ttransfer = (1 − D) · Ts .

(16)

From the volt–second balance, the converter voltage gain can be derived as Vo 2N = Vin (1 − D)

(17)

where the duty cycle of each switch must be greater than 50% and the turns ratio of the transformer can be calculated as N=

(1 − Dmax ) · Vo . 2 · Vin,min

(18)

Because S1 −S4 and S3 −S2 have the same number of turns with opposite polarities, the average voltage across the clamping capacitors C1 and C2 can be calculated as follows: VC1 = Vo − VS1 + VS4 = Vo

(19)

VC2 = Vo − VS3 + VS2 = Vo .

(20)

Moreover, the voltage ripples of the clamping capacitors C1 and C2 are defined to be smaller than 1% of the steady-state

25 · (2Dmax − 1) · TS Rmin

(21)

where Rmin is the full-load resistance. IV. E XPERIMENTAL R ESULTS The presented dual-inductor boost converter with output current ripple cancellation is implemented with 24–36-V input voltage and 200-V/3-A output power operated at 150-kHz switching frequency. The main parameters of the DIBCRC are listed in Table I, and the experimental results are shown as follows. Fig. 7 shows the key voltage waveforms of the DIBCRC captured at low-line and full-load operation condition. VDS1 and VDS2 are clamped to Vo /N (200/2.4 = 83.3 V), as shown in Ch3 and Ch4. The Ch5–Ch8 waveforms show the voltage across D1 to D4 which are clamped to the output voltage Vo (200 V). Due to the leakage energy being absorbed by the clamping capacitors C1 and C2 , all the diodes D1 −D4 are free of voltage spikes without adding an RC snubber. Fig. 8 shows the key current waveforms of the DIBCRC operating with low-line and full-load condition. Ch3 and Ch4 are the current waveforms of the input inductors L1 and L2 , respectively. The output current Io1 shown in Ch9 is the sum of Is1 (Ch5) and Is3 (Ch6). It is a continuous waveform due to the cancellation of the opposite pulsating current ripples of Is1 and Is3 . Because the leakage inductance criteria are not met well, a small current ripple can be seen. Moreover, the current waveform of the output filter capacitor ICo is shown in Ch10. Due to the output current ripple cancellation mechanism, the output capacitor can be significantly reduced. Finally, the measured efficiency of the proposed converter with different input voltage and load conditions is shown in Fig. 9. The DIBCRC has lower efficiency at low-line conditions because of the higher conduction loss in the primary. As shown, a maximum 95.58% efficiency is achieved at 36-V-input and 200-V/1.6-A-output operating condition. V. C ONCLUSION By integrating the proposed rectifier circuit with a conventional current-fed boost converter, a dual-inductor current-fed

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Fig. 7. Key experimental voltage waveforms under low-line and full-load operation condition.

boost converter with 150 kHz, 24–36 V input, and 200 V/600 W output prototype has been implemented and tested. In addition to having a high voltage gain (8.3 = 200 V/24 V) with a

Fig. 8. Key experimental current waveforms under low-line full-load operation condition.

small-turns-ratio (2.4 = 24/10) transformer, several additional advantages over the conventional current-fed boost converter can be obtained.

LEU et al.: NOVEL DIBCRC FOR HIGH-VOLTAGE-GAIN APPLICATIONS

Fig. 9.

Measured efficiency of the DIBCRC power stage.

First, the secondary leakage energies are absorbed and recovered so that the rectifier diodes are free of voltage spikes without adding an RC snubber circuit. Next, the rectifier output current is a continuous waveform due to having a built-in current ripple cancellation mechanism. Therefore, the number of required output filter capacitors can be reduced, resulting in increasing the power density performance. Moreover, two series-connected 250-V Schottky diodes can be used to replace one 500-V fast recovery diode due to having built-in voltage-clamping and voltage-sharing features. As a result, the conduction loss is reduced, and a maximum 95.58% efficiency is obtained. These features make the proposed rectifier circuit desirable for high-voltage-gain applications. R EFERENCES [1] L. Tang and G.-J. Su, “An interleaved reduced component count multivoltage bus DC/DC converter for fuel cell powered electric vehicle applications,” IEEE Trans. Ind. Appl., vol. 44, no. 5, pp. 1638–1644, Sep./Oct. 2008. [2] L. Zhu, “A novel soft-commutating isolated boost full-bridge ZVS-PWM DC–DC converter for bidirectional high power applications,” IEEE Trans. Power Electron., vol. 21, no. 2, pp. 422–429, Mar. 2006. [3] K. C. Tseng and T. J. Liang, “Novel high-efficiency step-up converter,” Proc. Inst. Elect. Eng.-Elect. Power Appl., vol. 151, no. 2, pp. 182–190, Mar. 2004. [4] S.-J. Jang, C.-Y. Won, B.-K. Lee, and J. Hur, “Fuel cell generation system with a new active clamping current-fed half-bridge converter,” IEEE Trans. Energy Convers., vol. 22, no. 2, pp. 332–340, Jun. 2007. [5] E.-H. Kim and B.-H. Kwon, “High step-up resonant push–pull converter with high efficiency,” IET Power Electron., vol. 2, no. 1, pp. 79–89, Jan. 2009. [6] X. Kong and A. M. Khambadkone, “Analysis and implementation of a high efficiency, interleaved current-fed full bridge converter for fuel cell system,” IEEE Trans. Power Electron., vol. 22, no. 2, pp. 543–550, Mar. 2007. [7] P. Thounthong, P. Sethakul, S. Rael, and B. Davat, “Design and implementation of 2-phase interleaved boost converter for fuel cell power source,” in Proc. 4th IET Conf. PEMD, Apr. 2–4, 2008, pp. 91–95. [8] M. H. Todorovic, L. Palma, and P. N. Enjeti, “Design of a wide input range DC–DC converter with a robust power control scheme suitable for fuel cell power conversion,” IEEE Trans. Ind. Electron., vol. 55, no. 3, pp. 1247–1255, Mar. 2008. [9] R.-J. Wai, C.-Y. Lin, R.-Y. Duan, and Y.-R. Chang, “High-efficiency power conversion system for kilowatt-level stand-alone generation unit with low input voltage,” IEEE Trans. Ind. Electron., vol. 55, no. 10, pp. 3702–3714, Oct. 2008.

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[10] R.-J. Wai, C.-Y. Lin, R.-Y. Duan, and Y.-R. Chang, “High-efficiency DC–DC converter with high voltage gain and reduced switch stress,” IEEE Trans. Ind. Electron., vol. 54, no. 1, pp. 354–364, Feb. 2007. [11] S. Jalbrzykowski and T. Citko, “Current-fed resonant full-bridge boost DC/AC/DC converter,” IEEE Trans. Ind. Electron., vol. 55, no. 3, pp. 1198–1205, Mar. 2008. [12] T.-F. Wu, Y.-S. Lai, J.-C. Hung, and Y.-M. Chen, “Boost converter with coupled inductors and buck–boost type of active clamp,” IEEE Trans. Ind. Electron., vol. 55, no. 1, pp. 154–162, Jan. 2008. [13] R. Sharma and H. Gao, “Low cost high efficiency DC–DC converter for fuel cell powered auxiliary power unit of a heavy vehicle,” IEEE Trans. Power Electron., vol. 21, no. 3, pp. 587–591, May 2006. [14] J. Wang, F. Z. Peng, J. Anderson, A. Joseph, and R. Buffenbarger, “Low cost fuel cell converter system for residential power generation,” IEEE Trans. Power Electron., vol. 19, no. 5, pp. 1315–1322, Sep. 2004. [15] J. L. Duarte, M. Hendrix, and M. G. Simoes, “Three-port bidirectional converter for hybrid fuel cell systems,” IEEE Trans. Power Electron., vol. 22, no. 2, pp. 480–487, Mar. 2007. [16] K. Jin and X. Ruan, “Hybrid full-bridge three-level LLC resonant converter—A novel DC–DC converter suitable for fuel-cell power system,” IEEE Trans. Ind. Electron., vol. 53, no. 5, pp. 1492–1503, Oct. 2006. [17] R.-J. Wai, L.-W. Liu, and R.-Y. Duan, “High-efficiency voltage-clamped DC–DC converter with reduced reverse-recovery current and switchvoltage stress,” IEEE Trans. Ind. Electron., vol. 53, no. 1, pp. 272–280, Feb. 2006. [18] S. C. Kim, S. H. Nam, S. H. Kim, D. T. Kim, and S. H. Jeong, “High power density, high frequency, and high voltage pulse transformer,” in Proc. Dig. Tech. Papers PPPS, 2001, vol. 1, pp. 808–811. [19] M. A. Perez, C. Blanco, M. Rico, and F. F. Linera, “A new topology for high voltage, high frequency transformers,” in Proc. Appl. Power Electron. Conf. Expo., Mar. 5–9, 1995, vol. 2, pp. 554–559. [20] C.-S. Leu, “Low voltage stress power converters,” U.S. Patent 7 515 439, Apr. 7, 2009.

Ching-Shan Leu (M’96) received the B.S. degree from the Department of Electrical Engineering, National Taiwan Ocean University, Keelung City, Taiwan, in 1974, the M.S. degree from the Department of Electrical and Computer Engineering, The Ohio State University, Columbus, in 1985, and the Ph.D. degree from the Department of Electrical and Computer Engineering, Virginia Polytechnic Institute and State University, Blacksburg, in 2006. From 1978 to 2003, he was an Assistant Researcher with the Chung-Shan Institute of Science and Technology, Longtan, Taiwan. Since 2006, he has been with the Department of Electrical Engineering, National Taiwan University of Science and Technology, Taipei, Taiwan, where he is currently an Assistant Professor. His research is focused on high-frequency power conversion technology for renewable energy applications.

Pin-Yu Huang (S’10) was born in Taipei, Taiwan, in 1985. He received the B.S. degree from the Department of Electrical Engineering, Tamkang University, Taipei, in 2007. He is currently working toward the M.S. degree in the Department of Electrical Engineering, National Taiwan University of Science and Technology, Taipei. His research is focused on high-frequency power conversion technology for renewable energy applications.

Ming-Hui Li was born in Taoyuan, Taiwan, on December 21, 1981. He received the B.S. degree from the Department of Electrical Engineering, National United University, Miaoli, Taiwan, in 2003, and the M.S. degree from the Department of Electrical Engineering, National Taiwan University of Science and Technology, Taipei, Taiwan, in 2009. He is currently with the Skynet Electronics Company, Ltd., Taipei. His research interests include switching-mode dc–dc power converters.

IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 58, NO. 4, APRIL 2011

A Novel Dual-Inductor Boost Converter With Ripple Cancellation for High-Voltage-Gain Applications Ching-Shan Leu, Member, IEEE, Pin-Yu Huang, Student Member, IEEE, and Ming-Hui Li

Abstract—A new full-wave rectifier is proposed in this paper. By integrating the proposed rectifier with a classic current-fed converter as an example, a dual-inductor boost converter with ripple cancellation is presented. It has several features, such as high voltage gain with smaller transformer turns ratio, recovery of the transformer secondary leakage energy, low voltage stress on the rectifier diodes, and nonpulsating input and output currents. These properties make it desirable for high-frequency highefficiency high-voltage-gain applications, such as the renewable energy source power system. In addition to the circuit analysis and design, a 150-kHz 24–36-V-input 200-V/600-W-output converter prototype is implemented and tested to demonstrate its feasibility. Index Terms—Current-fed boost converter, current ripple cancellation, high voltage gain, lossless snubber, renewable energy.

I. I NTRODUCTION

I

N ADDITION TO experiencing a supply shortage in recent years, fossil fuel causes serious environmental pollution problems. Therefore, developing a high-efficiency renewable energy power system, as shown in Fig. 1, has become an urgent issue of concern. Due to wide-range low dc output voltage, the renewable energy sources (solar cell as an example) cannot directly support the ac or dc electrical appliances. Consequently, a step-up converter is required to provide high voltage gain from low dc output voltage to generate 200-V high dc voltage as the input voltage of the cascaded dc–ac inverter or dc-dc converter. Several step-up converter topologies, classified with the voltage-fed configuration and current-fed configuration, have been proposed in the literature [1]–[17]. By employing the voltage-fed configuration, however, a large number of input electrolyte capacitor is required to meet the severe ripple current characteristic of a renewable energy source. In addition, the required voltage boosting function is only realized by using a transformer with large turns ratio. As a result, large parasitic capacitances and leakage inductance are induced [18], [19]. These parasitic components generate high voltage and high current spikes on the power devices. Therefore, a voltage-fed converter is not suitable for high-voltage-gain applications. Manuscript received July 31, 2009; revised December 11, 2009 and March 10, 2010; accepted March 22, 2010. Date of publication April 29, 2010; date of current version March 11, 2011. C.-S. Leu and P.-Y. Huang are with the Department of Electrical Engineering, National Taiwan University of Science and Technology, Taipei 10673, Taiwan (e-mail: [email protected]; [email protected]). M.-H. Li was with the Department of Electrical Engineering, National Taiwan University of Science and Technology, Taipei 10673, Taiwan. He is now with the Skynet Electronics Company, Ltd., Taipei 11570, Taiwan (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TIE.2010.2048835

Fig. 1.

Block diagram of a renewable energy power conversion system.

On the contrary, the current-fed configuration has a continuous input current. Thus, the number of input electrolytic capacitor can be minimized. Moreover, part of the voltage gain can be provided by the current-fed circuit itself; a smaller turns ratio transformer can be used, and the problems caused by the transformer parasitic components are alleviated. However, a full-wave rectifier and filter stages are still needed to obtain the required dc output voltage. Employing a centertapped rectifier as an example, the rectifier diodes and the output filter capacitor suffer from high voltage spike and high current ripple, respectively. In addition to utilizing a highvoltage-rating rectifier diode or a turn-off RC snubber circuit, a large number of the output filter capacitor has to be used. Thus, the converter efficiency and power density performance are degraded. To enhance the converter performance, therefore, a new fullwave rectifier is proposed [20], and a novel current-fed dualinductor boost converter with ripple cancellation (DIBCRC) as an application example is presented in this paper. Several features, such as high voltage gain with smaller transformer turns ratio, recovery of the transformer secondary leakage energy, low voltage stress on the rectifier diode, and continuous input and output currents, can be obtained. These properties make the converter desirable for high-voltage-gain applications, such as a solar-cell-powered system. In addition to the circuit analysis and design, a 150-kHz 24–36-V-input 200-V/600-W output converter prototype is implemented and tested to demonstrate its feasibility. II. A NALYSIS AND C IRCUIT O PERATION The circuit diagram of the DIBCRC is shown in Fig. 2. It comprises two input inductors L1 and L2 , one transformer Tr with one primary winding P1 and four identical secondary windings S1 −S4 , two switch pairs Q1 −Q3 and Q2 −Q4 , three clamping capacitors C1 , C2 , and C3 , one output capacitor Co ,

0278-0046/$26.00 © 2010 IEEE

LEU et al.: NOVEL DIBCRC FOR HIGH-VOLTAGE-GAIN APPLICATIONS

Fig. 2.

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Circuit diagram of the DIBCRC.

Fig. 4. Equivalent circuits for different operation stages of the DIBCRC.

Fig. 3.

Key waveforms of the DIBCRC.

and two series-connected diode pairs D1 −D2 and D3 −D4 . The key waveforms of the DIBCRC are shown in Fig. 3. To simplify the analysis of the presented converter, several assumptions are made. All semiconductors are ideal. Moreover, all capacitors C1 , C2 , C3 , and Co are sufficiently large that the voltage across each capacitor is constant during one switching period. The four leakage inductances Lk1 −Lk4 are identical. There are six operation stages within one switching cycle during the steady-state operation. However, only three

of them need to be described since the remaining stages are analogous. The equivalent circuit of each stage during a half period is shown in Fig. 4(a)–(c), and the operation principles are analyzed as follows. Stage 1 [t0 −t1 ] VGS2 is provided to turn on Q2 at t0 . Both Q1 and Q2 are turned on during this time interval t0 − t1 , i.e., Tcharge . The voltage across the transformer primary is thus shorted, and two inductor currents iL1 and iL2 increase linearly. Without being forward biased, the four rectifier diodes D1 , D2 , D3 , and D4 are all turned off. Part of the load currents are thus provided by the clamping capacitors C1 and C2 through

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(+)

(−)

C1 −S1 −Lk1 −R−S4 −Lk4 −C1 (−) S2 −Lk2 −C2 , respectively.

(+)

and C2 −S3 −Lk3 −R−

Stage 2 [t1 −t2 ] At t1 , Q1 is turned off, and the body diode of Q3 is turned on. The inductor current iL2 increases continuously as (1), and the inductor current iL1 decreases linearly as (2) Vi (t − t0 ) L2 Vc−Vi iL1 (t) = iL1 (t1 ) − (t − t1 ). L1

iL2 (t) = iL2 (t0 ) +

(1) (2)

The voltage across the transformer primary winding P1 is clamped to VC3 , which is the sum of the input voltage and the inductor voltage vL1 . The input power is transferred to the load via transformer windings S1 and S2 during the time interval t1 −t3 , i.e., Ttransfer . In addition to providing the load current, part of the input power is used to charge the clamping capacitors C1 and C2 through S4 −S2 −Lk2 −D2 −D1 −C1 −Lk4 −S4 and S3 −C2 −D2 −D1 −S1 −Lk1 −Lk3 −S3 , respectively. The average voltage across C1 or C2 is thus clamped to VO because the voltages across S1 −S4 or S2 −S3 cancelled each other. Moreover, the voltage stresses of D3 and D4 are equal to VO due to the turning on of D1 and D2 . At t2 , the current through Q3 reaches zero and starts to reverse. Q3 should be turned on before t2 so that zero-voltage switching operation can be achieved.

Fig. 5.

Key current waveforms of the rectifier circuit.

Fig. 6.

Current ripple distribution during the t1 −t2 time interval.

Stage 3 [t2 −t3 ] During this time interval, the operation is identical to that in [t1 −t2 ] except that the current through Q3 is negative until Q1 is turned on at t3 . The operations of the following stages during another half period are analogous. At t0 , Q2 is turned on again to start another switching cycle. As mentioned, the proposed rectifier circuit has an output current ripple cancellation mechanism so that the rectifier output current Io1 is a continuous waveform. Several key current waveforms are thus shown in Fig. 5 to explore the criteria. Accordingly, (3) has to be met ΔIs1 + ΔIs3 = 0

ΔIs2 + ΔIs4 = 0.

(3)

On the other hand, the current ripples during switch tran+ sients, for instance t− 1 to t1 , are shown in Fig. 6(a) and (b), and the diode current ripple ΔID can be derived as ΔID = |ΔIs1 | + |ΔIs4 | = |ΔIs2 | + |ΔIs3 |.

(4)

To simplify the analysis, the equivalent series resistance of the clamping capacitors and output filter capacitor can be ignored without making significant deviations. Consequently, the current ripple distribution among the secondary windings S1 , S2 , S3 , and S4 can be calculated as follows: Lk4 |ΔIs1 | = ΔID Lk1 + Lk4 Lk3 |ΔIs2 | = ΔID Lk2 + Lk3

(5) (6)

Lk2 Lk2 + Lk3 Lk1 |ΔIs4 | = ΔID . Lk1 + Lk4

|ΔIs3 | = ΔID

(7) (8)

Applying (5)–(8) into (3), the criteria to achieve a continuous rectifier output current can be obtained as Lk1 Lk4 = . Lk3 Lk2

(9)

Practically, one of the conditions, namely, LK1 = LK4 and LK2 = LK3 , can be approximately realized by using the sandwich winding scheme in the transformer construction. Moreover, the currents Ia , Ib , and Ic labeled in Fig. 5 can be calculated by applying the ampere–second balance

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on the clamping capacitor under Lk1 = Lk4 and Lk2 = Lk3 assumptions Io 2 Io (2 · D − 1) Ib = 4(1 − D)

Ia =

Ic =

Io (2D − 3) . 4(D − 1)

(10) (11) (12)

+ − − During t+ 0 < t < t1 or t2 < t < t3 time interval, the output current Io1 is calculated as

Io1 = 2 · Ia = Io .

TABLE I M AIN PARAMETERS OF DIBCRC

voltage Vo . Thus, the clamping capacitors C1 and C2 can be expressed as

(13) C1,2 >

+ − − During t+ 1 < t < t2 or t3 < t < t0 time interval, the output current Io1 is calculated as

Io1 = Ic − Ib = Io .

(14)

Therefore, the current Io1 is equal to the load current Io all the time. Ideally, the output capacitor Co can be eliminated. III. D ESIGN C ONSIDERATIONS It can be seen from the previous descriptions that there are two phases, namely, Tcharge and Ttransfer , within each half of the switching cycle. The Tcharge and Ttransfer time intervals are given as follows: 1 Tcharge = D − (15) · Ts 2 Ttransfer = (1 − D) · Ts .

(16)

From the volt–second balance, the converter voltage gain can be derived as Vo 2N = Vin (1 − D)

(17)

where the duty cycle of each switch must be greater than 50% and the turns ratio of the transformer can be calculated as N=

(1 − Dmax ) · Vo . 2 · Vin,min

(18)

Because S1 −S4 and S3 −S2 have the same number of turns with opposite polarities, the average voltage across the clamping capacitors C1 and C2 can be calculated as follows: VC1 = Vo − VS1 + VS4 = Vo

(19)

VC2 = Vo − VS3 + VS2 = Vo .

(20)

Moreover, the voltage ripples of the clamping capacitors C1 and C2 are defined to be smaller than 1% of the steady-state

25 · (2Dmax − 1) · TS Rmin

(21)

where Rmin is the full-load resistance. IV. E XPERIMENTAL R ESULTS The presented dual-inductor boost converter with output current ripple cancellation is implemented with 24–36-V input voltage and 200-V/3-A output power operated at 150-kHz switching frequency. The main parameters of the DIBCRC are listed in Table I, and the experimental results are shown as follows. Fig. 7 shows the key voltage waveforms of the DIBCRC captured at low-line and full-load operation condition. VDS1 and VDS2 are clamped to Vo /N (200/2.4 = 83.3 V), as shown in Ch3 and Ch4. The Ch5–Ch8 waveforms show the voltage across D1 to D4 which are clamped to the output voltage Vo (200 V). Due to the leakage energy being absorbed by the clamping capacitors C1 and C2 , all the diodes D1 −D4 are free of voltage spikes without adding an RC snubber. Fig. 8 shows the key current waveforms of the DIBCRC operating with low-line and full-load condition. Ch3 and Ch4 are the current waveforms of the input inductors L1 and L2 , respectively. The output current Io1 shown in Ch9 is the sum of Is1 (Ch5) and Is3 (Ch6). It is a continuous waveform due to the cancellation of the opposite pulsating current ripples of Is1 and Is3 . Because the leakage inductance criteria are not met well, a small current ripple can be seen. Moreover, the current waveform of the output filter capacitor ICo is shown in Ch10. Due to the output current ripple cancellation mechanism, the output capacitor can be significantly reduced. Finally, the measured efficiency of the proposed converter with different input voltage and load conditions is shown in Fig. 9. The DIBCRC has lower efficiency at low-line conditions because of the higher conduction loss in the primary. As shown, a maximum 95.58% efficiency is achieved at 36-V-input and 200-V/1.6-A-output operating condition. V. C ONCLUSION By integrating the proposed rectifier circuit with a conventional current-fed boost converter, a dual-inductor current-fed

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IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 58, NO. 4, APRIL 2011

Fig. 7. Key experimental voltage waveforms under low-line and full-load operation condition.

boost converter with 150 kHz, 24–36 V input, and 200 V/600 W output prototype has been implemented and tested. In addition to having a high voltage gain (8.3 = 200 V/24 V) with a

Fig. 8. Key experimental current waveforms under low-line full-load operation condition.

small-turns-ratio (2.4 = 24/10) transformer, several additional advantages over the conventional current-fed boost converter can be obtained.

LEU et al.: NOVEL DIBCRC FOR HIGH-VOLTAGE-GAIN APPLICATIONS

Fig. 9.

Measured efficiency of the DIBCRC power stage.

First, the secondary leakage energies are absorbed and recovered so that the rectifier diodes are free of voltage spikes without adding an RC snubber circuit. Next, the rectifier output current is a continuous waveform due to having a built-in current ripple cancellation mechanism. Therefore, the number of required output filter capacitors can be reduced, resulting in increasing the power density performance. Moreover, two series-connected 250-V Schottky diodes can be used to replace one 500-V fast recovery diode due to having built-in voltage-clamping and voltage-sharing features. As a result, the conduction loss is reduced, and a maximum 95.58% efficiency is obtained. These features make the proposed rectifier circuit desirable for high-voltage-gain applications. R EFERENCES [1] L. Tang and G.-J. Su, “An interleaved reduced component count multivoltage bus DC/DC converter for fuel cell powered electric vehicle applications,” IEEE Trans. Ind. Appl., vol. 44, no. 5, pp. 1638–1644, Sep./Oct. 2008. [2] L. Zhu, “A novel soft-commutating isolated boost full-bridge ZVS-PWM DC–DC converter for bidirectional high power applications,” IEEE Trans. Power Electron., vol. 21, no. 2, pp. 422–429, Mar. 2006. [3] K. C. Tseng and T. J. Liang, “Novel high-efficiency step-up converter,” Proc. Inst. Elect. Eng.-Elect. Power Appl., vol. 151, no. 2, pp. 182–190, Mar. 2004. [4] S.-J. Jang, C.-Y. Won, B.-K. Lee, and J. Hur, “Fuel cell generation system with a new active clamping current-fed half-bridge converter,” IEEE Trans. Energy Convers., vol. 22, no. 2, pp. 332–340, Jun. 2007. [5] E.-H. Kim and B.-H. Kwon, “High step-up resonant push–pull converter with high efficiency,” IET Power Electron., vol. 2, no. 1, pp. 79–89, Jan. 2009. [6] X. Kong and A. M. Khambadkone, “Analysis and implementation of a high efficiency, interleaved current-fed full bridge converter for fuel cell system,” IEEE Trans. Power Electron., vol. 22, no. 2, pp. 543–550, Mar. 2007. [7] P. Thounthong, P. Sethakul, S. Rael, and B. Davat, “Design and implementation of 2-phase interleaved boost converter for fuel cell power source,” in Proc. 4th IET Conf. PEMD, Apr. 2–4, 2008, pp. 91–95. [8] M. H. Todorovic, L. Palma, and P. N. Enjeti, “Design of a wide input range DC–DC converter with a robust power control scheme suitable for fuel cell power conversion,” IEEE Trans. Ind. Electron., vol. 55, no. 3, pp. 1247–1255, Mar. 2008. [9] R.-J. Wai, C.-Y. Lin, R.-Y. Duan, and Y.-R. Chang, “High-efficiency power conversion system for kilowatt-level stand-alone generation unit with low input voltage,” IEEE Trans. Ind. Electron., vol. 55, no. 10, pp. 3702–3714, Oct. 2008.

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[10] R.-J. Wai, C.-Y. Lin, R.-Y. Duan, and Y.-R. Chang, “High-efficiency DC–DC converter with high voltage gain and reduced switch stress,” IEEE Trans. Ind. Electron., vol. 54, no. 1, pp. 354–364, Feb. 2007. [11] S. Jalbrzykowski and T. Citko, “Current-fed resonant full-bridge boost DC/AC/DC converter,” IEEE Trans. Ind. Electron., vol. 55, no. 3, pp. 1198–1205, Mar. 2008. [12] T.-F. Wu, Y.-S. Lai, J.-C. Hung, and Y.-M. Chen, “Boost converter with coupled inductors and buck–boost type of active clamp,” IEEE Trans. Ind. Electron., vol. 55, no. 1, pp. 154–162, Jan. 2008. [13] R. Sharma and H. Gao, “Low cost high efficiency DC–DC converter for fuel cell powered auxiliary power unit of a heavy vehicle,” IEEE Trans. Power Electron., vol. 21, no. 3, pp. 587–591, May 2006. [14] J. Wang, F. Z. Peng, J. Anderson, A. Joseph, and R. Buffenbarger, “Low cost fuel cell converter system for residential power generation,” IEEE Trans. Power Electron., vol. 19, no. 5, pp. 1315–1322, Sep. 2004. [15] J. L. Duarte, M. Hendrix, and M. G. Simoes, “Three-port bidirectional converter for hybrid fuel cell systems,” IEEE Trans. Power Electron., vol. 22, no. 2, pp. 480–487, Mar. 2007. [16] K. Jin and X. Ruan, “Hybrid full-bridge three-level LLC resonant converter—A novel DC–DC converter suitable for fuel-cell power system,” IEEE Trans. Ind. Electron., vol. 53, no. 5, pp. 1492–1503, Oct. 2006. [17] R.-J. Wai, L.-W. Liu, and R.-Y. Duan, “High-efficiency voltage-clamped DC–DC converter with reduced reverse-recovery current and switchvoltage stress,” IEEE Trans. Ind. Electron., vol. 53, no. 1, pp. 272–280, Feb. 2006. [18] S. C. Kim, S. H. Nam, S. H. Kim, D. T. Kim, and S. H. Jeong, “High power density, high frequency, and high voltage pulse transformer,” in Proc. Dig. Tech. Papers PPPS, 2001, vol. 1, pp. 808–811. [19] M. A. Perez, C. Blanco, M. Rico, and F. F. Linera, “A new topology for high voltage, high frequency transformers,” in Proc. Appl. Power Electron. Conf. Expo., Mar. 5–9, 1995, vol. 2, pp. 554–559. [20] C.-S. Leu, “Low voltage stress power converters,” U.S. Patent 7 515 439, Apr. 7, 2009.

Ching-Shan Leu (M’96) received the B.S. degree from the Department of Electrical Engineering, National Taiwan Ocean University, Keelung City, Taiwan, in 1974, the M.S. degree from the Department of Electrical and Computer Engineering, The Ohio State University, Columbus, in 1985, and the Ph.D. degree from the Department of Electrical and Computer Engineering, Virginia Polytechnic Institute and State University, Blacksburg, in 2006. From 1978 to 2003, he was an Assistant Researcher with the Chung-Shan Institute of Science and Technology, Longtan, Taiwan. Since 2006, he has been with the Department of Electrical Engineering, National Taiwan University of Science and Technology, Taipei, Taiwan, where he is currently an Assistant Professor. His research is focused on high-frequency power conversion technology for renewable energy applications.

Pin-Yu Huang (S’10) was born in Taipei, Taiwan, in 1985. He received the B.S. degree from the Department of Electrical Engineering, Tamkang University, Taipei, in 2007. He is currently working toward the M.S. degree in the Department of Electrical Engineering, National Taiwan University of Science and Technology, Taipei. His research is focused on high-frequency power conversion technology for renewable energy applications.

Ming-Hui Li was born in Taoyuan, Taiwan, on December 21, 1981. He received the B.S. degree from the Department of Electrical Engineering, National United University, Miaoli, Taiwan, in 2003, and the M.S. degree from the Department of Electrical Engineering, National Taiwan University of Science and Technology, Taipei, Taiwan, in 2009. He is currently with the Skynet Electronics Company, Ltd., Taipei. His research interests include switching-mode dc–dc power converters.