A Novel Metal-Mountable Electrically Small Antenna for ... - IEEE Xplore

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IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 62, NO. 11, NOVEMBER 2014

A Novel Metal-Mountable Electrically Small Antenna for RFID Tag Applications With Practical Guidelines for the Antenna Design Jun Zhang and Yunliang Long, Senior Member, IEEE

Abstract—A novel dual-layer electrically small radio-frequencyidentification (RFID) tag antenna is proposed for metallic object applications. With a proximity-coupled feed method, two rotationally symmetric loaded via-patches are fed through an embedded dual-element planar inverted-F antenna (PIFA) array. With this 0.04 configuration, the antenna volume is reduced to 0.08 0.007 while the measured antenna gain is 0.08 dBi at the frequency of 923 MHz. Studies demonstrate that the proposed antenna is a good candidate for ultra-high-frequency RFID tags to be mounted on metallic surfaces, especially in size-constrained scenarios. Meanwhile, a figure of merit, namely, NBG, is presented, with which a comparison among electrically small tag antennas is carried out. Finally, several guidelines are given out to facilitate the miniaturization of RFID tag antennas for metallic object applications. Index Terms—Dual layer, electrically small antenna (ESA), figure of merit (FoM), loaded via-patch, metallic surface, radio-frequency identification (RFID), rotationally symmetric, tag antenna, ultra-high frequency (UHF).

I. INTRODUCTION

R

ADIO-FREQUENCY identification (RFID) has been receiving more attention recently because of its long-readrange and low-manufacturing-cost characteristics. The key issues for long reading distances are the gain of the antenna, the efficiency of the rectifier, and the power consumption of the chip [1]. Together, with the power sensitivity of the microchip, the tag antenna plays a significant role in tag performance, such as overall size, read range, and compatibility with tagged objects [2]. The design goal of the ultra-high frequency (UHF) RFID tag antenna is to reduce its volume, enlarge its bandwidth, and improve its gain, while enough budgets should be reserved for the reliability and robustness of the system. As we know, in most RFID applications, the tag should be directly attached to a tagged object. Since the incident electromagnetic (EM) wave is totally reflected from the metallic surface with a phase reversal, it will change the antenna’s radiation Manuscript received June 08, 2013; revised December 20, 2013; accepted August 25, 2014. Date of publication September 04, 2014; date of current version October 28, 2014. This work was supported by the Natural Science Foundation of China (61172026, 41376041). The authors are with the Department of Electronics and Communication Engineering, Sun Yat-sen University, Guangzhou 510006, China, and also with the SYSU-CMU Shunde International Joint Research Institute, Foshan 528300, China (e-mail: [email protected]; [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TAP.2014.2354412

pattern, input impedance, and resonant frequency. This change depends on the material and size of the object and the distance from it [3], [4]. Therefore, a major challenge to the widespread adoption of RFID technology in the UHF band is the antenna efficiency degradation caused by the nearby object, especially an object made of metallic material [5], [6]. One obstacle in designing the RFID tag antenna is to alleviate the influence of the tagged object. Compared with the general definition of bandwidth based on the voltage standing-wave ratio (VSWR), a more comprehensive definition of bandwidth in the RFID application is based on the realized gain [2]. Generally, 3-dB variations in either chip sensitivity or impedance matching can change the tag range by about 40% [7]; thus, proper impedance matching between the antenna and microchip is of paramount importance. In fact, a tag antenna must be designed to meet the operating band specified in at least one country/region (e.g., 866 to 869 MHz in Europe, 920 to 925 MHz in China, 902 to 928 MHz in the U.S., or 916 to 924 MHz in Japan). Consequently, broadening the bandwidth is an effective means to overcome the operating frequency shift or impedance variation caused by tagged objects or manufacturing errors [8]. To broaden the bandwidth of a UHF RFID tag antenna for metallic object applications, one choice is to introduce a lossy substrate (e.g., FR4) in the antenna design, thus improving the bandwidth easily [9]. However, this solution is less worthy because it is at the expense of antenna gain [10], and studies on meander-line antennas [11], [12] have demonstrated this. In addition, an antenna with multimode characteristics could improve the bandwidth [13]. However, the evaluation of the eigenvalue is hard, and most important, it is not easy to excite the desired mode in an ESA because of size limitations [14], [15]. Another obstacle in designing the RFID tag antenna is to improve the radiation efficiency and, thus, the antenna gain, or the reading performance. The study in [16] showed that gain, bandwidth, and other radiation properties of microstrip antennas could be enhanced by using parasitic elements placed between the radiating patch and ground plane. Furthermore, [17] proved that radiation enhancement would occur when the surface currents of all radiating elements were made in the same phase. However, due to size miniaturization, current cancellation is hard to avoid, and so the improvement in radiation efficiency is not prominent compared with single-layer ones [18], [19]. To improve antenna gain, one choice is to degenerate the multiresonant modes into one, but as mentioned before, the design

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ZHANG AND LONG: NOVEL METAL-MOUNTABLE ELECTRICALLY SMALL ANTENNA

procedure is hard and time-consuming. Metamaterials can be used to improve the radiation efficiency, but they are always too complex or too expensive to apply [20]. Considering that the achievable radiation efficiency is proportional to the electrical size of the antenna, it is best to try to make use of the available space as much as possible [21]. Then, a 3-D antenna [22] was designed to enhance the antenna gain; however, this meander-line topology was bulky and hard for mass production, especially in the size and cost-limited RFID applications. In addition, substrates with high-permittivity, low-loss characteristics were used to increase the electrical size of the antennas; however, they always suffered poor bandwidths and high costs [23], [24]. Recently, [25] studied a dual-element planar-inverted-F antenna (PIFA) based on a high impedance surface (HIS) unit to lower the profile of the conventional PIFAs. However, since microchip impedance is always designed with a high quality factor to increase the read range, this type of antenna has some difficulty in impedance matching. Therefore, [26] tried to improve the matching performance based on a loaded bar. In [27], a conductive layer was inserted into the antenna and made it function as a capacitance to further minimize the antenna size; however, low efficiency was a byproduct while the impedance was still hard to tune. Besides, [10] implemented a stacked shunt capacitor to adjust the antenna impedance, and similarly, [28] used a loop feed method to overcome the impedance tuning problem. Nevertheless, since the latter two antennas have only used one-half of the available space, there is still ample scope for us to reduce the antenna size while maintaining radiation efficiency, or vice-versa. The loaded via-patch is often found [29] in the small antenna design, and this patch not only functions as a capacitance to reduce the size, but also changes the current flows and affects the radiation efficiency. Naturally, the proximity-coupled feed method can be used in this stacked antenna to feed the loaded patch, and [30] used this feed method to minimize an antenna size to . It was also used in the design of an RFID tag antenna for metallic object applications [31]; however, the volume of this antenna is still cumbersome. Anyway, the capacitive feed via-patch loaded antenna has the benefit of small size and large bandwidth, but more often than not, the current property of the feed method often leads to radiation cancellation. Today, there is an increasing demand for tag antenna-based sensors because of their low cost [32] (e.g., in the field of structural health monitoring (SHM) [33]), where (resonant) frequency shift is a good choice to be used in pervasive sensing applications [34]; hence, antennas with narrow bandwidths and high-radiation-efficiency characteristics are more favorable [35]. Generally, there is a tradeoff between efficiency and bandwidth for a given antenna size [36]. This paper is dedicated to guiding the design of ultra small, metal-mountable tag antennas with a gain-enhanced characteristic. Inspired by the loaded via-patch structures and HIS-based topologies, we propose a dual-layer antenna with a feed network embedded in the middle to capacitively drive the radiation of two loaded via-patches at the top. By carefully designing the structural configuration, we successfully eliminate the current cancellation, but increase the electrical size of the via-patches. Consequently, radiation

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Fig. 1. Structural configuration of the proposed antenna: (a) top radiator, (b) feed radiator, and (c) side view.

enhancement occurs in the proposed antenna. We would like to organize our paper as follows. At first, we propose a via-patch loaded, dual-layer antenna structure based on the proximity coupled feed method to minimize the antenna size. Next, we analyze the impedance and radiation performance of this antenna on the metallic surface. After that, a new figure of merit (FoM) is presented to assess the performances of differently sized electrically small antennas (ESAs) and, ultimately, several guidelines are given out to facilitate the miniaturization of RFID tag antennas for metallic object applications. II. ANTENNA STRUCTURE AND DESIGN A. Antenna Structure The structural configuration of this antenna is illustrated in Fig. 1(a)–(c), the substrate of which consists of two different layers. The upper layer is polytetrafluorethylene (PTFE), with a thickness of 0.8 mm, a relative permittivity of 2.65, and a dielectric loss tangent of 0.0016, while the lower layer is FR4, with a thickness of 1.6 mm, a relative permittivity of 4.4, and a dielectric loss tangent of 0.01. The total volume of this antenna is 26 mm 14 mm 2.4 mm. Two patches shorted to the ground plane, namely, a feed radiator, are printed on the FR4 substrate with a small gap in between, and a differential feed is imposed in the center of this gap, which is shown in Fig. 1(b). Since this configuration is hard to tune the impedance, two loaded via-patches, namely, top radiator, are printed on the PTFE substrate with a distance of 0.8 mm above the feed radiator, which is shown in Fig. 1(a). Besides, for test convenience, we move the feed point in the feed radiator to the upper layer. After that, a compact dual-layer tag antenna is constructed. It is worth noting that the cost-effective, low-loss PTFE, and high-loss FR4 substrates chosen here are a tradeoff between antenna efficiency and bandwidth.

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Fig. 2. Fabricated antenna with 8.5 mm, 4 mm, 0.8 mm.


26 mm, 0.5 mm,

14 mm, 2 mm, 2 mm, 1.6 mm,

B. Antenna Design To improve the gain of a lossy antenna on metallic surfaces, the most important thing is to enhance its radiation efficiency. In the proposed antenna, we choose a dual-element PIFA array to function as a feed network. Since the electrical size of this structure is too small to draw the interrogating power, it only functions as an impedance-matching network. However, a little radiation occurs near the small gap [37], and that is why we call it the feed radiator. At the same time, we choose two loaded via-patches to act as the top radiator because slotting is not a valid solution in the design of a compact, high-efficiency antenna [9]. Naturally, the top radiator is fed by the underneath network through the proximity coupled feed method. In order to validate the design procedure, a prototype antenna is modeled and simulated using a full-wave simulator Ansoft HFSS. The antenna is designed for conjugate impedance matching to the passive microchip of NXP UCODE G2XM [38], the nominal read sensitivity of which is 15 dBm while the input impedance is at 920 MHz. Therefore, the tag antenna should have an input impedance near the conjugate value of for sufficient power delivery to the tag chip. Besides, to improve the tag performance on metallic surfaces, an environment and tag co-design methodology [39] has been used by modeling the antenna on a 100 mm 100-mm sized perfect electrical conductor (PEC). After parameter optimization, the antenna was manually fabricated through the photolithography and etching technique, which is shown in Fig. 2. The maximum activation distance of the tag along the direction, under the hypothesis of light sight propagation between reader and tag antennas [40] is governed by (1) where is the tag gain, is the wavelength, is the power transmitted by reader, is the gain of the transmitting antenna, th is the minimum threshold power to activate the tag chip, and is the polarization mismatch between the reader and tag antennas. The power transmission coefficient , which accounts for the impedance mismatch between the chip

Fig. 3. Simulated and measured input impedances and reflection coefficients of the proposed tag antenna on a ground-plane-sized 100 mm 100 mm: (a) . input impedance and (b) reflection coefficient of

( by

) and the antenna (

), is given

(2) where

is the conjugate value of the antenna impedance. III. ANTENNA CHARACTERISTICS AND ANALYSIS

A. Impedance Characteristics on Metallic Surfaces Since the tag microchip includes an energy-storage stage, its input reactance is strongly capacitive. Consequently, most of the RFID microchips available in the UHF band exhibit an input reactance roughly ranging from 100 to 400 , or more, while the real part is about an order of magnitude smaller, or less [41]. In this design, the impedance measurement was carried out using an Agilent E5071C vector network analyzer (VNA). Moreover, a port-extension technique proposed in [39] was also used to measure the input impedance of this differential feed antenna. Fig. 3(a) shows the simulated and measured input impedances of the proposed antenna as well as the conjugate impedance of


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Fig. 4. Input impedances of the proposed antenna mounted on a conductive ground plane that is 100 mm 100 mm in size with a variation of tcw, where R_chip and X_chip represent the real and imaginary parts of the conjugate chip impedance.

the specified chip. As indicated, the measured resistance and reactance of the antenna vary around the conjugate value of the microchip at 925 MHz. Based on the impedances shown in Fig. 3(a), the reflection coefficients of are calculated and available in Fig. 3(b). It could be observed that the measured half-power bandwidth is 16 MHz (917–933 MHz). It is important to point out that the impedance measurement is a hard task for most differential-feed ESAs, since various errors could be induced in the implementation of the measurement system. In Fig. 3, we can observe considerable discrepancies between the simulated and measured results, which may be found in the differential-feed asymmetric antennas [28], especially in the electrically small ones [24], [42], [43], because they are more sensitive to parameter changes. In this design, a small gap between the two substrate layers was introduced in the soldering process, which has shifted the resonance to higher frequency. However, this shift could be tuned back easily by changing the area of the capacitive cover (tcw), which indeed validates the feasibility of the following impedance tuning method. The imperfectness of the test fixture, such as the inescapable discontinuity between the fixture and antenna [44], the inefficient calibration [45], or the radiation disturbance from the conductive shielding layer [46] will also greatly affect the measurement accuracy. Further simulation shows that the size of the test fixture has a significant influence on the accuracy of the impedance measurement, and more discussions about this topic can be found in [11]. The adjustability of the antenna input impedance is given in Fig. 4. Actually, there are many factors that could be selected to act as a tuning parameter; however, the subunit chosen for tuning here is the area of the capacitive cover (tcw) in the top radiator. It is obvious that by adjusting the tcw in the loaded viapatches, the resonant frequency can be adjusted flexibly, which makes the conjugate impedance matching possible. Besides, it is worth mentioning that the antenna quality factor decreases a little with the increase of tcw.

Fig. 5. Current distributions of the proposed tag antenna on a conductive ground plane that is 100 mm 100 mm in size: (a) top radiator, (b) feed radiator, and (c) ground plane (in part).

B. Radiation Characteristics on Metallic Surfaces The radiation behaviors of the antenna are directly related to the current distributions; hence, Fig. 5(a)–(c) is presented to depict the simulated surface current distributions of the proposed antenna at 920 MHz. We can see that the interspersion of the via-holes helps to achieve a dominating horizontal current distribution in the feed radiator [47]. Meanwhile, the current flows in the feed network are greatly affected by the radiator above through the proximity-coupled method. Furthermore, we can also observe that the current magnitude in the top radiator is much larger than that of the feed radiator, which implies the former is the primary radiator. Together, with the ground plane, the distribution of the viaholes in the top radiator helps to increase the electrical length of the proposed antenna. Meanwhile, the rotationally symmetric configuration helps to turn the current flows of the top radiator either in the same direction (negative ) or in the perpendicular (negative ) direction, which can be easily observed in Fig. 5(a). These current distributions not only create radiation simultaneously in the four edges, but also remarkably enhance the

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Fig. 6. Simulated and measured radiation efficiencies and realized gains of the proposed tag antenna on a ground plane that is 100 mm 100 mm in size, where the “dash line” represents the simulated values with a loss tangent of an FR4 substrate ranging from 0.00 to 0.02 in a step of 0.01 while the “solid line” represents the measured values with a loss tangent of 0.01.

efficiency compared to the conventional meander line antennas [9], [48]. It is worth mentioning that the impedance bandwidth can be broadened by inducing a slight difference in the physical length of the two via-patches, resulting from splitting the single resonance into two adjacent ones, whereas this is always at the expense of lowering the radiation efficiency. The radiation efficiency, together with the realized gain of the antenna, is disclosed in Fig. 6, from which we can find that the simulated radiation efficiency is about 40% while the simulated realized gain almost reaches 0 dBi when the loss tangent of the FR4 substrate is 0.01. At the same time, we can see that efficiency and realized gain decrease with an increase in the loss tangent of the FR4 substrate; however, the effect of loss tangent is strong in radiation efficiency but weak in realized gain. This implies that the increase in loss tangent would improve matching performance. As is already known, the most important tag characteristic is the read range and usually 2.5 m is a critical requirement for general applications in 4-W effective isotropic radiated power (EIRP) [40]. To study the reading performance of the proposed tag on metallic surfaces, the read ranges are presented in Fig. 7, where the antenna is directly attached to four ground planes that are 100 mm 100 mm, 200 mm 200 mm, 400 mm 400 mm, and 500 mm 500 mm, in size, respectively. It can be observed that the 2.5-m range-width [2] is more than 20 MHz in all cases. Meanwhile, we can also find some fluctuations in the read range when the size of the ground plane increases; however, this phenomenon will disappear when the ground-plane size exceeds 500 mm 500 mm, which is practical for most metallic object applications. In order to measure the radiation parameters of the proposed antenna, a method based on the measurement of the minimum tag turn on power was used [49], the mechanism and feasibility of which have been studied in [50] and [51]. Fig. 8 shows the test setup for this measurement in an anechoic chamber, where

Fig. 7. Simulated and measured read ranges of the proposed tag, where the “dash line” represents the simulated values in the ground plane that is sized M1, M2, M3, and M4, respectively, corresponding to 100 mm 100 mm, 200 mm 200 mm, 400 mm 400 mm, and 500 mm 500 mm, while the “solid line” represents the measured value in the ground-plane size of M1.

Fig. 8. Test setup for the measurement of minimum tag turn on power.

the distance between the two antennas is fixed at 0.9 m. Due to the firmware limitation, the measurement was only carried out under the operating frequency of 900–928 MHz or less. The measured minimum tag turn on power was 13.6 dBm at the center frequency of 923 MHz while the measured half-power bandwidth of the realized gain was about 8 MHz. At the same time, the measured radiation efficiency, realized gain, and read range in the valid measurable frequency band have been separately displayed in Figs. 6 and 7, from which we can see that the maximum radiation efficiency is close to 23%, the maximum realized gain is nearly 1.4 dBi, and the maximum read range is almost 5.5 m. Furthermore, we can discover that the simulated and measured values in the realized gain and read range agree well except for a little shift in the center frequency. However, a big difference can be observed between the simulated and measured radiation efficiencies, which is probably caused by the measurement errors in the minimum tag turn on power pattern, such as a slight misalignment of the reader and tag antennas because of the high directivity of the tag antenna.


IV. PERFORMANCE COMPARISONS OF RFID TAG ESAS FOR METALLIC OBJECT APPLICATIONS A. Studies in the Figure of Merit for ESAs As we know, the physical limitations of antennas first investigated by Wheeler [52] and Chu [53] govern the tradeoffs among antenna size, bandwidth, and gain. For an arbitrary antenna, when its size satisfies the requirement of 0.5, it is called an electrically small antenna, where is the free-space wave number ( ), and is the radius of an imaginary sphere circumscribing the maximum physical dimension of the antenna. Conventionally, the most acceptable lower bound of the quality factor for a lossy ESA is [36], [54]

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So far as we know, the realized gain is of paramount importance in the tag antenna design, and should be taken into consideration in the FoM. Therefore, [11] defined a 3-dB realized gain bandwidth to permit the comparisons of different tags whatever may be the reader’s transmitting power and the microchip’s sensitivity, whereas similar definitions could be found in the works of [2] and [40] as well. Indeed, a much earlier work [60] proposed a so-called supergain-bandwidth FoM, which is Gain

(8)

is the normal gain defined by Harwhere rington [61]. Furthermore, the bandwidth gain product of conventional microstrip antennas is given out by way of example [62]

(3) Gain where 1 for a single-mode (TE or TM) antenna and 2 for a dual-mode (TE and TM) antenna. Up to now, many works have been done to study the physical limitations of antennas. Reference [55] presented the maximum possible ratio of gain to quality factor for a directional ESA

(4) Reference [56] found the relationship between the bandwidth and quality factor of an antenna (5) is the fractional matched bandwidth of where VSWR at an angular frequency , is the maximum-allowable VSWR, and the tuned antenna quality factor can be obtained directly from the antenna impedance (6) where the prime denotes the frequency derivative. To compare the performance of ESAs, a convenient FoM is the ratio of quality factor . At the same time, since the bandwidth efficiency product is directly related to the channel capacity for a given transmitter power and propagation path [57], the antenna performance can be best described by the product of gain and bandwidth [58]. As a result, [59] gave out a closed-form bandwidth efficiency product of for a quarter-wavelength antenna. Recently, [36] summarized the performance limits of various ESAs, where the general bandwidth efficiency product of was used as the FoM, that is (7)

(9)

where is a constant number, is the length of radiation edge, is the length of the nonradiation edge, is the thickness of the substrate, and is the wavelength in free space. Above all, to unambiguously evaluate the merits of an arbitrary electrically small RFID tag antenna designed for metallic object applications, a new FoM, called normalized bandwidth gain product (NBG) is defined as follows: (10) which indicates the proximity of the antenna performance to its is the realized gain bandwidth, is the limit. Here, maximum realized gain, is the maximum-allowable gain variation while 3-dB is always set to represent a 40% variation in the read range, and is the maximum ratio of gain to quality factor for a directional lossy antenna. It is worth mentioning that the introduction of the factor in (10) is to remove the effect of the substrate thickness, such as in (9). However, the here includes not only the substrate thickness but also the gap from the antenna to the metallic surface, if any. The maximum ratio of gain to quality factor for a directional lossy antenna could be expressed as (11) is equal to the ratio of the directional That is to say, lossless antenna mentioned in (4). Nevertheless, due to the profile limitation in the practical antenna design, there is a lot of space left empty in the minimum sphere enclosing the antenna and its image (with a ground plane); hence, many nonpropagating energies are stored in this sphere, which makes the bound of (11) higher than its exact value [58]. Besides, for the sake of contrast, we normalize (7), and it becomes (12) It is worth noting that the value of NBE is always less than one while that of NBG could be greater than unity. Yet for both values, the larger the better.

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TABLE I CHARACTERISTICS OF ELECTRICALLY SMALL TAG ANTENNAS PROPOSED FOR METALLIC OBJECT APPLICATIONS.

B. Parameter Extraction Methods for RFID Tag Antennas In order to evaluate the antenna performance by the proposed FoM, first we should extract the fractional bandwidth. For each antenna, we record the measured center frequency and its 3-dB bandwidth either from the plot of or the plot of antenna gain, where we assume these two bandwidths are equal to each other. To unify the diversity of the bandwidth definition, we will transform the 10-dB bandwidth to the 3-dB one by a multiple factor of , which can be inferred from (5). As those articles have already provided the measured realized gains, we use the given out gains directly. However, since gain information is seldom available while the measured read ranges are often accessible, we will revalue the realized gain from the read range with the help of (1), where it reads (13) is the EIRP, is the wavelength at the measured Here, center frequency, and is the typical reading sensitivity of the specified tag chip, which is 14 dBm for the Higgs-2 tag chip from Alien Technologies [63] and 13 dBm for the tag chip from Texas Instruments [64]. Since the tag antenna is always linearly polarized (LP), the polarization mismatch factor is set to 1 for an LP reader antenna but to 0.5 for a circularly polarized (CP) reader antenna, which now means a 3-dB polarization mismatch exists between the reader and the tag antennas. As (2) indicates, the power transmission coefficient can be calculated directly from the reflection coefficient of , and then we can easily extract the antenna gain of from the realized gain of . After that, the antenna efficiency of can be obtained from (14) is the upper bound of antenna gain. It is well known where that there is no bound to the antenna gain obtained from currents confined in an arbitrary small volume [65]; moreover, it is also believed that the antenna gain could be larger than its normal

while the quality factor approximates to its lower gain bound [55]. However, for a practical antenna, the maximumallowable gain is always limited [66]. Furthermore, since the condition of equal excitation of and spherical modes must be satisfied to maximize the antenna gain while and spherical modes are almost the only modes that can be excited in an ESA [67], we set the maximum gain of an arbitrary directional ESA to 3, just as (4) indicates. C. Comparisons of ESAs for Metallic Object Applications After the completion of the process that was just given, we extracted the antenna parameters of eight types ( and are of the same type) of electrically small RFID tags designed for metallic object applications, which are displayed in Table I. For each antenna developed by the research group ( , , , and ) of the Electronics and Telecommunications Research Institute (ETRI), owing to the absence of polarization information, we assumed that the read range was measured with an LP reader antenna, which might be inferred from [68]. Meanwhile, a loss tangent of 0.002 is assumed for the low-loss ceramic substrate used in the antenna design [72] due to the absence of loss tangent information. For the same reason, we have assumed a loss tangent of 0.1 for the high-loss PVC substrate used in the antenna presented by Park [73], [74]. The overall performance comparisons are carried out in Fig. 9, where the two FoMs of NBG and NBE are given out together. Compared with the antenna presented by , we can observe that the bandwidth improvement in is due to the increase in the substrate thickness (or aspect ratio), which motivates the introduction of the . However, even though the utilization of a low-loss, high-dielectric substrate can improve reading performance, it is not a good choice because of its high cost; moreover, the robust operation also leads to an increase in the antenna profile, which will further limit the applications of this antenna type. We find that no matter what the antenna topology is, the new FoM proposed in this paper can unambiguously assess its performance. That is to say, the


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The topology choice is to improve space utilization as much as possible. As [36] indicates, the best performance of an antenna can be achieved if the aspect ratio is close to unity. Many conventional metal-mountable RFID tag antennas suffer poor performance because of their critical profile limitations, and undoubtedly, the multilayer-type antenna has demonstrated improvements in its performance [75]. However, this type of antenna has some negative effects on its cost since it involves more complex manufacturing than the single-layer one. V. CONCLUSION

Fig. 9. Overall performance comparisons of electrically small RFID tag antennas for metallic object applications based on the NBG and NBE.

NBG here could be used to act as a rule of thumb in designing metal-mountable tag antennas. As far as we know, we cannot optimize the gain and bandwidth at the same time, and there exists a tradeoff between gain degradation and bandwidth expansion, where efficiency is the key factor. For an ESA, the substrate, even having only a little loss, could profoundly affect the radiation efficiency [66]. To make things worse, the existence of the ground plane reduces the antenna’s radiation resistance, and this reduction will become severe when the antenna profile decreases. That is to say, the major challenge for the miniaturization of metal-mountable RFID tag antennas is to maintain the realized gain. Based on the aforementioned analysis, we summarize three guidelines to facilitate the miniaturization of RFID tag antennas for metallic object applications, which are listed as follows. The antenna optimization is to maximize the realized gain and its bandwidth, but not just the impedance bandwidth. Due to the limitations of the tags’ size and cost, it is easy to depress the radiation efficiency of the tag antenna to improve the impedancematching performance. However, poor efficiency always results in poor antenna gain, which will further lead to degradation in the realized gain even if perfect conjugate impedance matching is achieved between the tag antenna and the tag chip. This means that to improve the reading performance of an ESA in a specific region/country, it is better to increase the radiation efficiency while enough bandwidth budget is reserved. However, if the concept of range width is used, then the low efficiency could be compensated for by the evolution of chip sensitivity. The substrate selection involves choosing a material with low dielectric permittivity, low loss, and low-cost characteristics, and not a material with only high-loss (low-cost) characteristics. As [66] indicates, for an antenna with a given size, there exists a bound in the antenna gain, and this bound increases with the increase in the “loss merit factor.” Meanwhile, to achieve the best performance, the dielectric constant of the antenna substrate should be chosen as low as possible [36]. In addition, to promote the widespread adoption of RFID technology in the UHF band, the tag cost should be kept low.

In this paper, a narrow-bandwidth, gain-enhanced, electrically small RFID tag antenna with 0.29 has been proposed for metallic object applications. The antenna is simulated to have good conjugate impedance matching with one commercial tag chip and is fabricated to prove the validity of the design process. Both the simulated and the measured results demonstrate that the reading performance of the proposed antenna is reliable; therefore, this type of metal-mountable tag antenna is an attractive candidate in sensing applications, especially in size-constrained scenarios. It is also a representative example in the tradeoffs between performance and cost, between bandwidth and radiation efficiency. Meanwhile, the behavior assessment of ESAs has been studied, and a new FoM, namely, NBG is presented. After that, a comparison between the metal-mountable electrically small tag antennas has been carried out based on the NBG. Even though the study in this paper mainly focuses on ESAs, the NBG here functions as a rule in designing RFID tag antennas for metallic object applications, whether they are electrically small or not. Finally, several practical guidelines have been given out to guide the miniaturization of RFID tag antennas for metallic object applications. ACKNOWLEDGMENT The authors would like to express their gratitude to the three anonymous reviewers for their valuable suggestions and to X. Liu, Director of the Guangzhou IoT Testing and Technology Service Center, for his help in the measurements. REFERENCES [1] H. E. Nilsson, J. Siden, T. Olsson, P. Jonsson, and A. Koptioug, “Evaluation of a printed patch antenna for robust microwave RFID tags,” IET Microw. Antennas Propag., vol. 1, pp. 776–781, 2007. [2] G. Marrocco, “The art of UHF RFID antenna design: Impedance matching and size-reduction techniques,” IEEE Antennas Propag. Mag., vol. 50, no. 1, pp. 66–79, Feb. 2008. [3] L. Ukkonen, L. Sydanheimo, and M. Kivikoski, “Effects of metallic plate size on the performance of microstrip patch-type tag antennas for passive RFID,” IEEE Antennas Wireless Propag. Lett., vol. 4, pp. 410–413, 2005. [4] J. D. Griffin, G. D. Durgin, A. Haldi, and B. Kippelen, “RF tag antenna performance on various materials using radio link budgets,” IEEE Antennas Wireless Propag. Lett., vol. 5, pp. 247–250, 2006. [5] J. C. E. Sten, A. Hujanen, and P. K. Koivisto, “Quality factor of an electrically small antenna radiating close to a conducting plane,” IEEE Trans. Antennas Propag., vol. 49, no. 5, pp. 829–837, May 2001. [6] J. W. Lee and B. Lee, “Design of high- UHF radio-frequency identification tag antennas for an increased read range,” IET Microw. Antennas Propag., vol. 2, pp. 711–717, Oct. 2008. [7] P. V. Nikitin, K. V. S. Rao, R. Martinez, and S. F. Lam, “Sensitivity and impedance measurements of UHF RFID chips,” IEEE Trans. Microw. Theory Tech., vol. 57, no. 5, pt. 2, pp. 1297–1302, May 2009.

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[8] T. DeJeruyelle, P. Pannier, M. EgeJs, and E. Bergeret, “An RFID tag antenna tolerant to mounting on materials,” IEEE Antennas Propag. Mag., vol. 52, no. 4, pp. 14–19, Aug. 2010. [9] G. A. Mavridis, D. E. Anagnostou, and M. T. Chryssomallis, “Evaluation of the quality factor, , of electrically small microstrip-patch antennas [Wireless corner],” IEEE Antennas Propag. Mag., vol. 53, no. 4, pp. 216–224, Aug. 2011. [10] H. W. Son, “Design of RFID tag antenna for metallic surfaces using lossy substrate,” Electron. Lett., vol. 44, pp. 711–713, 2008. [11] K. Mohammadpour-Aghdam, S. Radiom, R. Faraji-Dana, G. A. E. Vandenbosch, and G. G. E. Gielen, “Miniaturized RFID/UWB antenna structure that can be optimized for arbitrary input impedance,” IEEE Antennas Propag. Mag., vol. 54, no. 2, pp. 74–87, Apr. 2012. [12] J. Zhang and Y. Long, “A miniaturized via-patch loaded dual-layer RFID tag antenna for metallic object applications,” IEEE Antennas Wireless Propag. Lett., vol. 12, pp. 1184–1187, 2013. [13] J. Zhang and Y. Long, “A dual-layer broadband compact UHF RFID tag antenna for platform tolerant application,” IEEE Trans. Antennas Propag., vol. 61, no. 9, pp. 4447–4455, Sep. 2013. [14] R. F. Harrington and J. R. Mautz, “Theory of characteristic modes for conducting bodies,” IEEE Trans. Antennas Propag., vol. 19, no. 5, pp. 622–628, Sep. 1971. [15] M. Cabedo-Fabres, E. Antonino-Daviu, A. Valero-Nogueira, and M. F. Bataller, “The theory of characteristic modes revisited: A contribution to the design of antennas for modern applications,” IEEE Antennas Propag. Mag., vol. 49, no. 5, pp. 52–68, Oct. 2007. [16] L. Ukkonen, M. Schaffrath, D. W. Engels, L. Sydanheimo, and M. Kivikoski, “Operability of folded microstrip patch-type tag antenna in the UHF RFID bands within 865–928 MHz,” IEEE Antennas Wireless Propag. Lett., vol. 5, pp. 414–417, 2006. [17] H.-D. Chen and Y.-H. Tsao, “Low-profile PIFA array antennas for UHF band RFID tags mountable on metallic objects,” IEEE Trans. Antennas Propag., vol. 58, no. 4, pp. 1087–1092, Apr. 2010. [18] H. W. Son, G. Y. Choi, and C. S. Pyo, “Open-ended two-strip meanderline antenna for RFID tags,” ETRI J., vol. 28, pp. 383–385, Jun. 2006. [19] A. E. Abdulhadi and R. Abhari, “Design and experimental evaluation of miniaturized monopole UHF RFID tag antennas,” IEEE Antennas Wireless Propag. Lett., vol. 11, pp. 248–251, 2012. [20] R. W. Ziolkowski and A. Erentok, “Metamaterial-based efficient electrically small antennas,” IEEE Trans. Antennas Propag., vol. 54, no. 7, pp. 2113–2130, Jul. 2006. [21] T. Bjorninen, A. Z. Elsherbeni, and L. Ukkonen, “Low-profile conformal UHF RFID tag antenna for integration with water bottles,” IEEE Antennas Wireless Propag. Lett., vol. 10, pp. 1147–1150, 2011. [22] A. Galchdar, D. V. Thiel, and S. G. O’Keefe, “Design methods for 3D RFID antennas located on a conducting ground plane,” IEEE Trans. Antennas Propag., vol. 57, no. 2, pp. 339–346, Feb. 2009. [23] J. S. Kim, W. Choi, and G. Y. Choi, “UHF RFID tag antenna using two PIFAs embedded in metallic objects,” Electron. Lett., vol. 44, pp. 1181–1182, 2008. [24] W. Choi, J. S. Kim, J. H. Bae, G. Y. Choi, and C. S. Pyo et al., “RFID tag antenna coupled by shorted microstrip line for metallic surfaces,” ETRI J., vol. 30, pp. 597–599, Aug. 2008. [25] S.-L. Chen and K.-H. Lin, “A slim RFID tag antenna design for metallic object applications,” IEEE Antennas Wireless Propag. Lett., vol. 7, pp. 729–732, 2008. [26] K.-H. Lin, S.-L. Chen, and R. Mittra, “A looped-bowtie RFID tag antenna design for metallic objects,” IEEE Trans. Antennas Propag., vol. 61, no. 2, pp. 499–505, Feb. 2013. [27] S.-L. Chen, “A miniature RFID tag antenna design for metallic objects application,” IEEE Antennas Wireless Propag. Lett., vol. 8, pp. 1043–1045, 2009. [28] P. H. Yang, L. Yan, L. J. Jiang, W. C. Chew, and T. T. Ye, “Compact metallic RFID tag antennas with a loop-fed method,” IEEE Trans. Antennas Propag., vol. 59, no. 12, pp. 4454–4462, Dec. 2011. [29] C. Y. Chiu, K. M. Shum, and C. H. Chan, “A tunable via-patch loaded PIFA with size reduction,” IEEE Trans. Antennas Propag., vol. 55, no. 1, pp. 65–71, Jan. 2007. [30] R. L. Li, G. DeJean, M. M. Tentzeris, and J. Laskar, “Development and analysis of a folded shorted-patch antenna with reduced size,” IEEE Trans. Antennas Propag., vol. 52, no. 2, pp. 555–562, Feb. 2004. [31] H.-W. Son and S.-H. Jeong, “Wideband RFID tag antenna for metallic surfaces using proximity-coupled feed,” IEEE Antennas Wireless Propag. Lett., vol. 10, pp. 377–380, 2011. [32] R. Bhattacharyya, C. Floerkemeier, and S. Sarma, “Low-cost, ubiquitous RFID-tag-antenna-based sensing,” Proc. IEEE, vol. 98, no. 9, pp. 1593–1600, Sep. 2010.

[33] X. H. Yi, C. H. Cho, J. Cooper, Y. Wang, and M. M. Tentzeris, “Passive wireless antenna sensor for strain and crack sensing—electromagnetic modeling, simulation, and testing,” Smart Mater. Struct., vol. 22, Aug. 2013. [34] R. Bhattacharyya, “Low-cost, passive UHF RFID tag antenna-based sensors for pervasive sensing applications,” Ph.D. dissertation, Mass. Inst. Technol., Cambridge, MA, USA, 2012. [35] S. Manzari and G. Marrocco, “Modeling and applications of a chemical-loaded UHF RFID sensing antenna with tuning capability,” IEEE Trans. Antennas Propag., vol. 62, no. 1, pp. 94–101, Jan. 2014. [36] D. F. Sievenpiper, D. C. Dawson, M. M. Jacob, T. Kanar, and K. Sanghoon et al., “Experimental validation of performance limits and design guidelines for small antennas,” IEEE Trans. Antennas Propag., vol. 60, no. 1, pp. 8–19, Jan. 2012. [37] S. Manzari, C. Occhiuzzi, S. Nawale, A. Catini, C. Di Natale, and G. Marrocco, “Humidity sensing by polymer-loaded UHF RFID antennas,” IEEE Sens. J., vol. 12, no. 9, pp. 2851–2858, Sep. 2012. [38] NP Semiconductors, “SL3ICS1002/1202 UCODE G2XM and G2XL,” Data sheet, Mar. 2011. [39] X. M. Qing, C. K. Goh, and Z. N. Chen, “Impedance characterization of RFID tag antennas and application in tag co-design,” IEEE Trans. Microw. Theory Tech., vol. 57, no. 5, pt. 2, pp. 1268–1274, May 2009. [40] K. V. S. Rao, P. V. Nikitin, and S. F. Lam, “Antenna design for UHF RFID tags: A review and a practical application,” IEEE Trans. Antennas Propag., vol. 53, no. 12, pp. 3870–3876, Dec. 2005. [41] G. Marrocco, “RFID antennas for the UHF remote monitoring of human subjects,” IEEE Trans. Antennas Propag., vol. 55, no. 6, pt. 2, pp. 1862–1870, Jun. 2007. [42] A. S. A. Jalal, A. Ismail, A. R. H. Alhawari, M. F. A. Rasid, and N. K. Noordin et al., “Miniaturized metal mount Minkowski fractal RFID tag antenna with complementary split ring resonator,” Prog. Electromagn. Res. C, vol. 39, pp. 25–36, 2013. [43] W. Choi, J. Kim, J. H. Bae, and G. Choi, “A small RFID tag antenna using proximity-coupling to identify metallic objects,” Microw. Opt. Technol. Lett., vol. 50, pp. 2978–2981, Nov. 2008. [44] S. Krevsky, “A note concernig the precise measurements of dipole antenna impedance,” IRE Trans. Antennas Propag., vol. 8, pp. 343–344, 1960. [45] K. D. Palmer and M. W. van Rooyen, “Simple broadband measurements of balanced loads using a network analyzer,” IEEE Trans. Instrum. Meas., vol. 55, no. 1, pp. 266–272, Feb. 2006. [46] T. Fukasawa, T. Yanagi, H. Miyashita, and Y. Konishi, “Extended S-parameter method including radiation pattern measurements of an antenna,” IEEE Trans. Antennas Propag., vol. 60, no. 12, pp. 5645–5653, Dec. 2012. [47] M. Hirvonen, P. Pursula, K. Jaakkola, and K. Laukkanen, “Planar inverted-F antenna for radio frequency identification,” Electron. Lett., vol. 40, pp. 848–850, Jul. 2004. [48] H. K. Ryu and J. M. Woo, “Miniaturisation of rectangular loop antenna using meander line for RFID tags,” Electron. Lett., vol. 43, pp. 372–374, 2007. [49] EPCglobal, “Static test method for applied tag performance testing,” May 2008. [50] Y.-H. Lee, M.-Y. Tsai, C.-F. Yang, and I. Lin, “A 3D RFID static test system using a spherical near-field antenna measurement chamber,” IEEE Trans. Antennas Propag., vol. 61, no. 1, pp. 362–370, Jan. 2013. [51] E. Koski, T. Bjorninen, L. Ukkonen, and L. Sydanheimo, “Radiation efficiency measurement method for passive UHF RFID dipole tag antennas,” IEEE Trans. Antennas Propag., vol. 61, no. 8, pp. 4026–4035, Aug. 2013. [52] H. A. Wheeler, “Fundamental limitations of small antennas,” Proc. IRE, vol. 35, pp. 1479–1484, 1947. [53] L. J. Chu, “Physical limitations of omnidirectional antennas,” J. Appl. Phys., vol. 19, pp. 1163–1175, 1948. [54] J. S. McLean, “A re-examination of the fundamental limits on the radiation of electrically small antennas,” IEEE Trans. Antennas Propag., vol. 44, no. 5, p. 672, May 1996. [55] G. Y. Wen, “Physical limitations of antenna,” IEEE Trans. Antennas Propag., vol. 51, no. 8, pp. 2116–2123, Aug. 2003. [56] A. D. Yaghjian and R. S. Best, “Impedance, bandwidth, and of antennas,” IEEE Trans. Antennas Propag., vol. 53, no. 4, pp. 1298–1324, Apr. 2005. [57] P. Hansen and R. Adams, “The minimum for spheroidally shaped objects: Extension to cylindrically shaped objects and comparison to practical antennas,” IEEE Antennas Propag. Mag., vol. 53, no. 3, pp. 75–83, Jun. 2011.


[58] G. Y. Wen, “Optimization of the ratio of gain to ,” IEEE Trans. Antennas Propag., vol. 61, no. 4, pt. 2, pp. 1916–1922, Apr. 2013. [59] A. Krall, J. McCorkle, J. Scarzello, and A. Syeles, “The omni microstrip antenna: A new small antenna,” IEEE Trans. Antennas Propag., vol. AP-27, no. 6, pp. 850–853, Nov. 1979. [60] L. P. Ivrissimtzis, M. J. Lancaster, and M. Esa, “Miniature superconducting coplanar strip antennas for microwave and MM-wave applications,” presented at the Int. Conf. Antennas Propag., Eindhoven, The Netherlands, 1995. [61] R. F. Harrington, “Effect of antenna size on gain, bandwidth, and efficiency,” J. Res. Nat. Bur. Stand., vol. 64-D, pp. 1–12, 1960. [62] D.-G. Fang, Antenna Theory and Microstrip Antennas. Boca Raton, FL, USA: CRC, 2009. [63] Alien Technology, “Higgs-2 EPC Class 1 Gen 2 RFID Tag IC,” Jul. 2008, data sheet. [64] Texas Instruments, Inc., “UHF Gen2 Strap RI-UHF-STRAP-08,” Oct. 2006, data sheet. [65] R. F. Harrington, Time Harmonic Electromagnetic Fields. Hoboken, NJ, USA: Wiley, 1961. [66] A. Arbabi and S. Safavi-Naeini, “Maximum gain of a lossy antenna,” IEEE Trans. Antennas Propag., vol. 60, no. 1, pp. 2–7, Jan. 2012. [67] R. L. Fante, “Quality factor of general ideal antennas,” IEEE Trans. Antennas Propag., vol. AP-17, no. 2, pp. 151–155, Mar. 1969. [68] J. S. Kim, W. K. Choi, and G. Y. Choi, “Small proximity coupled ceramic patch antenna for UHF RFID tag mountable on metallic objects,” Prog. Electromagn. Res. C, vol. 4, pp. 129–138, 2008. [69] W. S. Chen, J. C. Chen, and B. Y. Lee, “A compact UHF RFID tag antenna design for metallic objects,” presented at the IEEE AP-S Int. Symp., Toronto, ON, Canada, 2010. [70] J. Y. Park and J. M. Woo, “Miniaturised dual-band s-shaped RFID tag antenna mountable on metallic surface,” Electron. Lett., vol. 44, pp. 1339–1341, 2008. [71] J. S. Kim, W. Choi, G. Y. Choi, C. S. Pyo, and J. S. Chae, “Shorted microstrip patch antenna using inductively coupled feed for UHF RFID tag,” ETRI J., vol. 30, pp. 600–602, Aug. 2008. [72] J. Baker-Jarvis, R. G. Geyer, J. H. Grosvenor, Jr., and M. D. Janezic, “Dielectric characterization of low-loss materials a comparison of techniques,” IEEE Trans. Dielectr. Electr. Insul., vol. 5, no. 4, pp. 571–577, Aug. 1998. [73] D. K. Ghodgaonkar, V. V. Varadan, and V. K. Varadan, “A free-space method for measurement of dielectric constants and loss tangents at microwave frequencies,” IEEE Trans. Instrum. Meas., vol. 38, no. 3, pp. 789–793, Jun. 1989. [74] S. Chandran and S. Narayanankutty, “Dielectric behavior of polyaniline/polyvinyl chloride/nylon fiber composites at microwave frequencies,” Polym.-Plast. Technol. Eng., vol. 48, pp. 82–89, 2009.

5829

[75] S. Latif, L. Shafai, and C. Shafai, “Gain and efficiency enhancement of compact and miniaturised microstrip antennas using multi-layered laminated conductors,” IET Microw. Antennas Propag., vol. 5, pp. 402–411, 2011.

Jun Zhang was born in Zhejiang, China, in 1985. He received the B.Sc. degree in information engineering from the South China University of Technology (SCUT), Guangzhou, China, in 2008, and the Ph.D. degree in radio physics from Sun Yat-sen University (SYSU), Guangzhou, in 2013. His current research interests include electrically small antenna theory, microstrip antenna theory, and the design of novel ultra-high-frequency radio-frequency identification tag antennas for metallic object applications.

Yunliang Long (M’01–SM’02) received the B.Sc., M.Eng., and Ph.D. degrees in electronic engineering from the University of Electronic Science and Technology of China (UESTC), Chengdu, China, in 1983, 1989, and 1992, respectively. From 1992 to 1994, he was a Postdoctoral Research Fellow. After that, he was an Associate Professor and then a Full Professor with the Department of Electronics and Communication Engineering, Sun Yat-sen University, Guangzhou, China. From 1998 to 1999, he was a Visiting Scholar with IHF, RWTH University of Aachen, Aachen, Germany. From 2000 to 2001, he was a Research Fellow with the Department of Electronics Engineering, City University of Hong Kong, Hong Kong, China. From 2004 to 2013, he was the Head of the Department of Electronics and Communication Engineering. Currently, he is the Vice Dean of the School of Information Science and Technology, Sun Yat-sen University. He has authored and coauthored more than 180 academic papers. His research interests include antennas and propagation theory, electromagnetic theory in inhomogeneous lossy medium, computational electromagnetics, and wireless communication applications. Prof. Long is a member of the Committee of Microwave Society of CIE and is on the editorial board of the Chinese Journal of Radio Science.