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Abstract—This paper proposes a quad-mode digitally con- trolled oscillator (DCO) for multi-band frequency synthesizer application. The DCO consists of an ...
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A Quad-Mode DCO for Multi-standard Communication Application Bo Jiang, Tian Xia School of Engineering, University of Vermont

Abstract—This paper proposes a quad-mode digitally controlled oscillator (DCO) for multi-band frequency synthesizer application. The DCO consists of an LC-tank using three inductors and two varactor arrays. Four different inductances and capacitances are obtained by tuning corresponding varactor values, which result in four resonances to increase the frequency tuning range. The simulation results show that the proposed DCO output frequency covers the frequency ranges from 1.5GHz to 2.5GHz and from 4.3GHz to 6.2GHz, which are suitable for multistandard applications. Keywords—DCO, multi-standard, phase noise, phase locked-loop, tuning range.

I.

I NTRODUCTION

arrays (V ar1 , V ar2 ). By alternatively turning on two pairs of switches SW1N and SW1P , or SW2N and SW2P the circuit can be converted into two different structures as shown in Fig. 1 (b) and (c), respectively. In structure-I, L2 , V ar2 branch features capacitive or inductive through V ar2 tuning. In structure-II, L1 , V ar1 shunt shows capacitive or inductive by setting V ar1 values. Thus, there are two different modes in each structure producing two different frequency bands. Combing these two structures and four operation modes, the frequency ranging from 1.5GHz to 2.5GHz and from 4.3GHz to 6.2GHz can be obtained, which cover GPS, Bluetooth, WiFi 802.11a/b/g frequency bands. Moveover, as the proposed DCO eliminates the need of switches in the LC tank, the resistive loss and phase noise effects are greatly alleviated.

R

ECENTLY, digitally controlled oscillator (DCO) has become an essential function unit in frequency synthesizer in multi-standard wireless communication systems [1-3]. In such applications, the DCO must have wide frequency tuning range, low phase noise and low power consumption. LC tank DCO is widely adopted in a digital phase locked loop (PLL) frequency synthesizer. Comparing to the ring oscillator [4-5], LC tank DCO has superior phase noise performance. However, LC resonator typically has limited frequency tuning range due to limited varactor tuning capability. How to improve LC-DCO tuning scope has been an active research subject. In reference [6], authors present a tunable active inductor to obtain a wide frequency coverage. However, the active inductor usually doesnt result in low phase noise performance. In reference [7], authors present a dual-mode LC tank oscillator, where the frequency ranging from 3.14GHz to 6.44GHz is achieved from the core LC tank oscillator. While the lower frequency, from 0.25MHz to 3.22GHz, is achieved by a frequency divider chain. Although such design can achieve a wide frequency range, it is not power efficient. Switched inductor LC oscillator [8] has been proposed to increase frequency tuning capability. However, switch devices employed in the LC tank degrade the phase noise performance. Although large size switches can be utilized to reduce their resistive loss, the DCO frequency tuning range will be decreased due to their large parasitic capacitance. In reference [9], authors present a triple-mode oscillator using three coupled inductors. Each inductor has vertical dimensions to save chip area. However, vertical dimensions result in extra resistance loss and the quality factor for inductors is degraded [7]. In this paper, a method to achieve multi-band frequency DCO is proposed. The proposed DCO circuit schematic is shown in Fig. 1, which uses three inductors and two varactor

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Fig. 1. DCO models for (a) proposed DCO; (b) Structure-I, (c) Structure-II.

This paper is organized as follows. Section II analyzes the quad-mode DCO. Simulation results are presented in section III. Section IV contains the concluding remarks. II. Q UAD - MODE DCO MODEL The proposed DCO schematic is shown in Fig. 1 (a). The DCO consists of two pairs of switches SW1 including SW1N , SW1P and SW2 including SW2N , SW2P , three inductors and two varactor arrays. When SW1 is off and SW2 turns on, the DCO is converted to structure-I whose abstract model is shown in Fig. 1 (b). In this structure, by changing V ar2 value, L2 and V ar2 branch impedance can become either inductive or capacitive, which results in two different operating modes, named mode-I and mode-II. When SW1 is on and SW2 turns off, the DCO model can be transferred to structure-II as shown in Fig. 1 (c). L1 , V ar1 shunt can be capacitive or inductive through V ar1 tuning in mode-III and mode-IV.

2

A. Structure-I The DCO structure-I model is shown in Fig. 1 (b) when SW1 is off and SW2 is on. Assuming the capacitances of two varactors V ar1 and V ar2 are Cvar1 and Cvar2 respectively, the tank impedance can be calculated as: 1 1 Z(ω) = jωL1 || ||(2jωL2 + ) (1) jωCvar1 jωCvar2 The corresponding tank resonant frequency is fosc . The resonant frequency fs for series connected L2 and V ar2 subbranch is: p fs = 1/(2π L1 Cvar1 ) (2) By tuning the varactor V ar2 , fosc can be made lower than or higher than DCO output frequency fs . 1) Mode-I: The L2 , V ar2 branch is inductive when fosc > fs . The equivalent inductance is: 1 jωosc Cvar2

1 = 2L2 − 2 2 jωosc 4π fosc Cvar2 where ωosc is DCO radial frequency. In mode-I, the DCO output frequency fosc equals: q fosc1 = 1/[2π (L1 ||L02 ) · Cvar1 ] L02 = 2L2 +

1) Mode-III: L1 and V ar1 branch impedance is capacitive when fosc > fs0 . The equivalent capacitance is: C10 =

1/jωosc 1 = Cvar1 − 2 2 1 4π fosc L1 jωosc L1 || jωosc Cvar1

(11)

The resulting DCO output frequency fosc is: s L1 fosc3 = 1/[2π (2L2 − 2 2 ) · Cvar2 ] (12) 4π fosc L1 Cvar1 − 1

(4)

2) Mode-IV: L1 and V ar1 branch impedance becomes inductive when fosc < fs0 through V ar1 tuning. The equivalent inductance is:

Cvar2 1/jωosc = (5) 2 L C 1 − 8π 2 fosc (2jωosc L2 + jωosc1Cvar2 ) 2 var2

In this mode, the DCO output frequency fosc is: q fosc2 = 1/[2π L1 · (Cvar1 + C20 )]

DCO output frequency fosc (a) Mode-I; (b) Mode-II.

(3)

2) Mode-II: The L2 , V ar2 branch is capacitive when fosc < fs . The equivalent capacitance is: C20 =

Fig. 2.

(6)

By manipulating equations (4) and (6), Equations (7) and (8) can be obtained, which characterize DCO output frequency in each mode. Apparent to see that the DCO can produce higher output frequency in mode-I than in mode-II. Fig. 2 shows DCO output frequency in mode-I and mode-II when Cvar1 is changed from 0.5pF to 2.5pF . Other circuit parameters are: L1 = 2.5nH, L2 = 1.25nH, Cvar2 equals 2.5pF in mode-I, and 0.5pF in mode-II. From Fig. 2, it can be observed that two different frequency bands are produced in these two operating modes. In mode-I, DCO frequency band spans from 3.3GHz to 6.5GHz, while in mode II, DCO frequency band spans from 1.8GHz to 2.8GHz. B. Structure-II The DCO structure-II model is shown in Fig. 1 (c) when SW1 is on and SW2 is off. The corresponding tank impedance equals: 1 1 )|| (9) Z(ω) = (jω2L2 + ωL1 || jωCvar1 jωCvar2 L2 and V ar2 sub-branch resonant frequency equals: p (10) fs0 = 1/[2π 2L2 Cvar2 ] By tuning the capacitance of V ar1 , fs0 can be made lower than or higher than DCO output frequency fosc .

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L01 =

jωosc L1 || jωosc1Cvar1 jωosc

=

L1 1−

2 L C 4π 2 fosc 1 var1

(13)

The corresponding DCO output frequency is: s L1 ) · Cvar2 ] (14) fosc4 = 1/[2π (2L2 + 2 L C 1 − 4π 2 fosc 1 var1 By manipulating equations (12) and (14), the same equations (7) and (8) are obtained, which imply that DCO output frequency is higher in mode-III than in mode-IV . Fig. 3 shows DCO output frequency in mode-III and modeIV when Cvar2 is changed from 0.5pF to 2.5pF . Other circuit parameters are L1 = 2.5nH, L2 = 1.25nH, Cvar1 equals 2.5pF in mode-III and 0.5pF in mode-IV. As shown in Fig. 3, in mode-III, when Cvar2 is tuned from 0.5pF to 2.5pF , DCO output frequency is produced from 3.3GHz to 5GHz, while in mode-IV, the frequency is from 1.4GHz to 2.8GHz.

Fig. 3.

DCO output frequency fosc (a) Mode-III; (b) Mode-IV.

The DCO output frequency and frequency tuning range in L2 each mode are also dependent on inductor ratio RL = L . 1

3

fosc1,(3)

1 = 2π

fosc2,(4)

1 = 2π

s

s

L1 Cvar1 + L1 Cvar2 + 2L2 Cvar2 +

p

L1 Cvar1 + L1 Cvar2 + 2L2 Cvar2 −

p

(L1 Cvar1 + L1 Cvar2 + 2L2 Cvar2 )2 − 8L1 L2 Cvar1 Cvar2 4L1 L2 Cvar1 Cvar2 (L1 Cvar1 + L1 Cvar2 + 2L2 Cvar2 )2 − 8L1 L2 Cvar1 Cvar2 4L1 L2 Cvar1 Cvar2

Fig. 4 shows DCO output frequency versus different RL in each mode by applying different L2 value. The DCO output frequency is decreased with the raising of RL value in all operation modes. In mode-II and mode-III, the DCO output frequency tuning range is reduced with the increasing of RL value. From Fig. 4, when RL = 0.5, DCO output frequency from 3.3GHz-6.5GHz, 1.8GHz-2.8GHz, 3.3GHz-5GHz and 1.4GHz-2.8GHz can be obtained in mode-I, II, III and IV, respectively.

Fig. 4. Frequency output under different RL . L1 =2.5nH; L2 = 0.625nH; 1.25nH; 2.5nH. (a) Mode-I , (b) Mode-II, (c) Mode-III , (d) Mode-IV

Here the LC tank quality factor in each structure is also characterized. Assuming the quality factors of inductors L1 and L2 , and varactors V ar1 and V ar2 are QL1 , QL2 , QV ar1 and Qvar2 , respectively, thus the tank quality factors in structure I and structure II can be calculated as 1 1 1 + + )−1 (15) QI = ( QL1 Qvar1 QL2 − 2ω2 L2 C12 Qvar2 1

QII = ( QL2 −

1 )−1 var1 L1 2 2L2 (ω C1 − L1 ) 1

( Q1 + Q

+

1 Qvar2

)−1

III.

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(8)

E XPERIMENTAL R ESULTS

To validate the analysis in Section II, the proposed DCO circuit is implemented and simulated in IBM 0.13µm CMOS technology. The simulation tool is Cadence SpectreRF. The circuit operates under a 1.5V supply voltage. Fig. 5 shows the detailed schematic of the proposed multiband DCO. The circuit parameters are: L1 = 2.5nH, L2 = 1.25nH, V ar1 is tuned from 0.4pF -2pF and V ar2 tuning range is 0.66pF -3pF . Transistor sizes are listed in Table. II. Structure selection is achieved by controlling power supply switches SW1 and SW2 . The structure-I is utilized to generate higher frequency that covers WiFi 802.11a frequency band, whose bandwidth is 200MHz spanning from 5.15GHz to 5.35GHz and 5.65GHz to 5.85GHz. To meet such requirement, V ar1 consists of 4-bit coarse tuning and 11-bit fine tuning including 5-bit DAC controlled tuning. The coarse tuning varactor value is 0.38pF -1.85pF , which makes coarse tuning resolution to be 200MHz per step when output frequency is in 5-6GHz. The fine tuning varactor value is 60f F -180f F results in 120KHz frequency resolution in higher frequency band. The structure-II is applied to generate lower frequency that covers GPS, bluetooth and WiFi 802.11b and WiFi 802.11g, whose frequency range is from 1.56GHz to 2.48GHz. The maximum bandwidth among those standards is 80MHz. Therefore, V ar2 is configured with 5-bit and 11-bit fine tuning including 5bit DAC controlled tuning. The coarse tuning varactor value is 0.66pF -3pF , which makes coarse tuning resolution to be 25MHz. The fine tuning varactor value is 60f F -180f F which produces 15KHz resolution in lower frequency band.

(16)

As an example, in mode-I configuration, to achieve 5.6GHz oscillation frequency, the inductance values of L1 and L2 are set to be 2.5nH and 1.25nH, and the capacitance values of V ar1 and V ar2 are 0.7pF and 2.5pF. Their quality factor values are 6, 8.5, 50 and 50 respectively. Utilizing Eq. (15), the tank quality factor is calculated to be 3.3. The tank quality factor values at other specific frequencies in different modes are listed in Table I.

(7)

Fig. 5.

Detailed schematic of proposed DCO.

4

Fig. 6 shows the transient signal of DCO output at 5.6GHz (Mode-I) and 2.4GHz (Mode-II). Fig. 7 shows the output frequency with the control code and phase noise. The circuit parameters and performances of each mode are summarized in Table I.

Fig. 6.

Transient signal of DCO output, (a) 5.6GHz; (b) 2.4GHz.

published wideband VCOs/DCOs. In addition, the simulation shows that the DCO power consumption is within the range from 7.6 mW to 11.5 mW. IV. C ONCLUSION A quad-mode DCO, formed by two power supply switches, three inductors and two varactor arrays is presented in this paper. Switching between structures is obtained by controlling power supply switches. Mode selection is obtained by controlling the LC sub-branch inductive or capacitive characteristics. This design eliminates the need of using switch components in LC tank. Therefore, phase noise effect and the resistive loss associated with switches are alleviated. Two wide frequency bands, 1.5GHz∼2.5GHz and 4.3GHz∼6.2GHz are produced, which are suitable for applications of multi-standard wireless applications. R EFERENCES [1]

[2] [3]

Fig. 7. (a) Output frequency with control code in Mode-I, (b) Phase noise at frequency 5.6GHz and 2.4GHz.

[4]

[5] TABLE I. L1 /L2

2.5nH/ 1.25nH MOS. (W/L)

P ROPOSED DCO

CIRCUIT PARAMETERS .

Var. Mode Freq. Phase Noise Q KDCO pF GHz dBc/Hz@1MHz Hz V ar1 I 4.3-6.24 [email protected] 3.3 120K =0.4-2 II 2.21-2.56 [email protected] 2.4 35K V ar2 III 4.91-5.48 -114@5GHz 6.2 50K 0.66-3 IV 1.49-2.52 [email protected] 8.1 15K P1/P2/N3/N4: 100µm/0.13µm; P3/P4/P5/P6: 70µm/0.13µm N1/N2: 40µm/0.13µm; N5/N6: 30µm/0.13µm

[6]

[7]

[8] TABLE II.

C OMPARISON BETWEEN WIDEBAND VCO S /DCO S [9]

Ref. NO. [6] [7] [13] [14]

Tech. µm 0.18 0.18 0.13 0.18

Power mW 6-28 7.1-16.3 4.4-9.4 4.6-6

This work

0.13

7.6-11.5

Phase Noise dBc/Hz@1MHz [email protected] [email protected] [email protected] [email protected] [email protected] [email protected] [email protected]

DCO Freq. Range GHz % 0.5-3 143 3.14-6.44 69 1.3-6 128 2.4-2.52 4.9 4.65-5.12 9.6 1.49-2.56 53 4.3-6.24 37

F OMT dBc/Hz -180 -197 -201 -187 -191 -197 -187

[10]

[11]

F OMT is calculated to evaluate frequency tuning range along with phase noise [12]. fosc · F T R P ) + 10log( ) 10fof f set 1mW (17) where L(fof f set ) is phase noise at offset frequency fof f set . P is power consumption. F T R is frequency tuning range in percentage. Table II summarizes the comparison against other F OMT = L(fof f set ) − 20log(

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[12]

[13]

[14]

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