applied sciences Article
A Single-Stage High-Power Factor Converter with Synchronized Self-Excited Technique for LED Lighting Yong-Nong Chang 1 , Shun-Yu Chan 2 and Hung-Liang Cheng 3, * 1 2 3
*
ID
Department of Electrical Engineering, National Formosa University, Yunlin County 63201, Taiwan;
[email protected] Department of Electrical Engineering, Cheng Shiu University, Kaohsiung 83347, Taiwan;
[email protected] Department of Electrical Engineering, I-Shou University, Kaohsiung 84001, Taiwan Correspondence:
[email protected]; Tel.: +886-7-657-7711 (ext. 6634)
Received: 16 July 2018; Accepted: 16 August 2018; Published: 20 August 2018
Abstract: This paper proposes a single-stage, high power-factor light-emitting diode (LED) driver with a self-excited control scheme for the power switches. The self-excited mechanism is accomplished by fetching the driving voltages from a center-tapped transformer. The frequency of the driving voltages is exactly the same as the resonant frequency of the resonant converter, thus synchronizing the resonant frequency with the switching frequency and achieving zero-voltage switching (ZVS) and zero-current switching (ZCS) of power switches. The circuit topology is mainly composed of a half-bridge LC resonant converter, along with a boost-type power-factor corrector (PFC) to fulfill the single-stage structure, meaning that the presented LED driver possesses high power-factor features and low switching loss. Finally, a 40 W prototype circuit is implemented and tested, and the experimental results exhibit a satisfactory performance. Keywords: light-emitting diode (LED); power-factor correction; self-excited; single stage; soft switching
1. Introduction Light-emitting diode (LED) lighting has many advantages, such as high lighting efficiency, a compact size, and fast response. When compared with fluorescent lamps, LEDs are more environmentally friendly because they do not use mercury. As such, with these features, the usage of LED lighting has become an irresistible trend. Among the various advantages of LED, the most prominent still focuses on its high lighting efficiency, although not all questions about the interaction between humans and light emitted by the LEDs have been completely solved [1–3]. Nowadays, in pursuit of things such as high lighting quality, special lighting atmospheres, and/or energy-saving capacities, many LED lights have incorporated a dimming function. However, because there are many people who do not require higher-quality lighting, LED lighting without the dimming function is still widely used, such as in household or street lighting. For such lights, the key factor of lighting equipment selection depends on the cost–performance ratio of the product [4–6]. Thus, the development of a simple and low-cost LED driver is likely to greatly enhance the popularity of LED lighting. In the highly-competitive lighting market, the lighting drive circuit, using a self-excited control mechanism, has been widely used in many low-price products—especially in the applications of electronic ballast [7–10]. The purpose of a self-excited circuit is to drive the power switches repeatedly
Appl. Sci. 2018, 8, 1408; doi:10.3390/app8081408
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by using the polarity changes of the voltage across windings of a magnetic component, meaning that it does not need to use an integrated circuit (IC), and the components used to build a direct-current (dc) voltage for the IC can be saved. In the conventional self-excited fluorescent lamp electronic ballast, the circuit topology possesses a half-bridge (inductor-capacitor-capacitor) (LCC) resonant circuit by introducing a saturable transformer connected in series with a resonant circuit to generate the self-excited driving voltage that can drive bipolar junction transistors (BJT) [11–13]. Regarding driving a BJT, when it is used as a power switch, it requires a base current sufficiently high enough to drive the BJT to operate in the saturation region. However, driving a BJT into the saturation region will influence the charge carrier behavior in the (positive-negative) (PN) junction, thus further affecting the switching time thereof. In general, there are many such uncontrollable factors, including things like saturation point of an iron core, storage time carriers, ambient temperature, and humidity—and all of them hinder the precise control of the resonant frequency and voltage gain of the resonant circuit. On the other hand, active power-factor correctors (PFCs) are widely applied to the lighting circuit due to the increasingly strict regulations on power quality. Although the power factor can be effectively improved by adding a PFC circuit to the original lighting circuit, the number of circuit components then need to be increased, resulting in an uneconomical solution. Thus, aiming to find a cost-effective solution, researchers proposed some single-stage lighting circuits with self-excited control schemes by integrating a PFC and dc-to-dc resonant circuit [14–16]. However, using BJT as power switches invokes further problems [17], because among these single-stage circuits, two power BJTs could suffer from uneven current and lead to unequal conduction time between the two BJTs, thus making the simplification of single-stage PFC impossible. In this paper, a novel single-stage LED driver with a self-excited mechanism through integration of a boost-type PFC and a half-bridge series resonant converter is proposed. No additional integrated circuit is required. The driving voltages of the self-excited mechanism are taken directly from the center-tapped transformer, which is cascaded by a full-wave diode rectifier to obtain the dc output voltage. As compared with the traditional self-excited mechanism, it does not need to use an easily saturable transformer—thus, the number of components can be reduced. Two metal-oxide-semiconductor field-effect transistors (MOSFETs) serve as the power switches. The MOSFET is a voltage-driven component. Unlike the BJT-based circuit topology, the core saturation and carrier storage time problems can be avoided, thus making the circuit design easier. Moreover, since MOSFETs are used in the single-stage PFC circuit, the incidence of unequal currents in power switches is far less, meaning the uneven conduction times between the switches can be ignored. Therefore, the developed self-excited method is quite suitable for single-stage PFC application. In addition, the switching frequency of the MOSFETs can be identical to the resonant frequency of the resonant converter. The resonant converter operates at the resistive mode, thus accomplishing both the zero-voltage switching (ZVS) and zero-current switching (ZCS) performance of power switches [18–21]. Consequently, the developed self-excited single-stage converter will possess the advantages of having both a high power factor and low switching losses, and is also suitable for LED drive application. 2. Proposed Circuit Topology and Operation Analysis In our particular study, a half-bridge inductor-capacitor (LC) resonant converter, a boost-typed PFC, a starter circuit, and the developed self-excited circuit were integrated to constitute the novel single-stage structure, as shown in Figure 1. A list of symbols is as follows. S1 , S2 : power switches; DS1 , DS2 : intrinsic body diodes of the power switches, LPFC : inductance of the boost-typed PFC, T1 : center-tapped transformer, Np : turns of the primary winding, NS1 , NS2 : turns of the secondary windings, NSE1 , NSE2 : turns of the self-excited windings, D1 , D2 : full-wave rectifier diodes, Lr : inductance of the resonant inductor, Cr : capacitance of the resonant capacitor, Lm : inductance of the low-pass filter, Cm : capacitance of the low-pass filter, Cbus : capacitance of the dc-bus capacitor, and Co : capacitance of the output capacitor.
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3 of 16 vin : input line voltage, iin : input line current, iLPFC : PFC inductor current, Vcbus : dc-bus voltage, icbus : dc-bus current, Vo : output voltage, vgs1 : gate-to-source voltage of S1 , vgs2 : gate-to-source voltage iS1: drain-to-source current of S1, iS2: drain-to source current of S2, ir: resonant current of the resonant of S2 , iS1 : drain-to-source current of S1 , iS2 : drain-to source current of S2 , ir : resonant current of the circuit, vAB: input voltage of the resonant circuit, vp: voltage across the primary winding, fr: resonant resonant circuit, vAB : input voltage of the resonant circuit, vp : voltage across the primary winding, fr : frequency of the resonant circuit, fs: switching frequency of power switches, Mvr: voltage gain of the resonant frequency of the resonant circuit, fs : switching frequency of power switches, Mvr : voltage resonant circuit, Q: quality factor of the resonant circuit, s: switching angular frequency, r: resonant gain of the resonant circuit, Q: quality factor of the resonant circuit, s : switching angular frequency, angular frequency, Rp: equivalent load resistance across the primary winding, RLED: equivalent r : resonant angular frequency, Rp : equivalent load resistance across the primary winding, RLED : resistance of the LED string, and D: duty ratio of the power switches. equivalent resistance of the LED string, and D: duty ratio of the power switches. MOSFETs S1 and S2 acted as power switches, and the antiparallel diodes DS1 and DS2 as their MOSFETs S1 and S2 acted as power switches, and the antiparallel diodes DS1 and DS2 as their intrinsic body diodes, respectively. The center-tapped transformer T1 and diodes D1 and D2 formed a intrinsic body diodes, respectively. The center-tapped transformer T1 and diodes D1 and D2 formed full-wave rectifier. The turn ratio of the primary winding to the secondary windings was n: 1:1. In a full-wave rectifier. The turn ratio of the primary winding to the secondary windings was n: 1:1. addition, T1 had two more windings which were employed to provide power switches with selfIn addition, T1 had two more windings which were employed to provide power switches with excited driving voltages. Besides, T1 could provide galvanic insulation with respect to the input line. self-excited driving voltages. Besides, T1 could provide galvanic insulation with respect to the input The low-pass filter (Lm and Cm) was used to remove the high-frequency current of the inductor current, line. The low-pass filter (Lm and Cm ) was used to remove the high-frequency current of the inductor because in this way, the input line current would be sinusoidal with the same frequency as the input current, because in this way, the input line current would be sinusoidal with the same frequency as line voltage. For simplification of the circuit analysis, the capacitances of Cbus and Co were assumed the input line voltage. For simplification of the circuit analysis, the capacitances of Cbus and Co were to be large enough that the dc-bus voltage Vcbus and the output voltage V0 could be regarded as assumed to be large enough that the dc-bus voltage Vcbus and the output voltage V 0 could be regarded constant. As shown in Figure 1, the power switch S2 should carry both currents of the PFC converter as constant. As shown in Figure 1, the power switch S2 should carry both currents of the PFC converter and the resonant converter. The voltage across S2 is equal to the dc-bus voltage when it is off. Hence, and the resonant converter. The voltage across S2 is equal to the dc-bus voltage when it is off. Hence, S2 is the crucial component of the proposed circuit and its voltage and current ratings will determine S2 is the crucial component of the proposed circuit and its voltage and current ratings will determine the power range. the power range.
Half-Bridge LC Resonant Converter
iin vin
Lm
LPFC
vgs1
iLPFC Cm
iS1 + DS1 vAB
S2
Cbus Vcbus+_
vgs2
Boost-Typed PFC
vp N p _
_ iS2 DS2
Starter Circuit
Co + Vo _
+
Sync hronize d Self-Excited Circ uit vgs1 vgs2
NS1
LED Strings
S1
T1 D n:1:1 iD1 1
Cr
Lr
ir
icbus
NS2 D2 iD2 NSE1 NSE2
Figure 1. The developed, synchronized, self-excited, single-stage, high power-factor light-emitting Figure 1. The developed, synchronized, self-excited, single-stage, high power-factor light-emitting diode (LED) driver. diode (LED) driver.
2.1. 2.1.Self-Excited Self-ExcitedDriving DrivingVoltages Voltages The The center-tapped center-tapped transformer transformerhas has two two additional additional windings—the windings—theturn turnratio ratioof of the the primary primary winding to the additional windings is m: 1:1. The self-excited driving voltages were obtained by winding to the additional windings is m: 1:1. The self-excited driving voltages were obtained stepping down the high-frequency square wave via the transformer. A pair of symmetrical square by stepping down the high-frequency square wave via the transformer. A pair of symmetrical wave voltages (vgs1, vgs2(v ) were generated by the self-excited windings, as depicted in Figure 2. Their square wave voltages gs1 , vgs2 ) were generated by the self-excited windings, as depicted in Figure 2. frequency fs was inf synchronism with the resonant frequency fr which was created by the combination Their frequency s was in synchronism with the resonant frequency fr which was created by the of resonant inductor L r and resonant capacitor Cr, as can be expressed by Equations (1) and (2). combination of resonant inductor L and resonant capacitor C , as can be expressed by Equations (1) r
and (2).
r
1
fr = 1 f r = 2π√ Lr C r 2π Lr Cr fs = fr
The voltage on the primary winding vp of the transformer can be achieved by Equation (3),
(1) (1) (2)
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fs = fr
(2)
The voltage on the primary winding vp of the transformer can be achieved by Equation (3), Appl. Sci. 2018, 8, x FOR PEER REVIEW
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v p = DVcbus Mvr v p = D V cb u s M
(3)
(3)
vr
where Mvr is the voltage gain of the resonant circuit, and D is the duty ratio of the lower-arm power is the voltage gain of theand resonant D is the duty ratio of the lower-arm powerto: Mvr control switchwhere S2 . The voltages vgs1 vgs2 circuit, may beand approximated as square waves, equal switch S2. The control voltages vgs1 and vgs2 may be approximated as square waves, equal to: − v p− v p
v gs1v gs= −v 2 = = 1 = − v gsgs2
m
(4)
m
(4)
the negative indicatesthe thephase phase relationship between vgs1 and . v . wherewhere the negative signsign indicates relationship between vgs1vgs2 and gs2
vAB
Vcbus
vp
0.5Vcbus
t
-0.5Vcbus vgs2
t vgs1 t Figure Schematic waveforms of of thethe self-excited mechanism. Figure 2.2.Schematic waveforms self-excited mechanism.
LC Resonant Converter 2.2. LC2.2. Resonant Converter Since the self-excited driving voltages were directly obtained from the resonant circuit, the
Since the self-excited driving voltages were directly obtained from the resonant circuit, switching frequency of the power MOSFETs was, of course, synchronous with the resonant voltage the switching frequency of the power MOSFETs was, of course, synchronous with the resonant voltage waveform, thus making the switching frequency and the resonant frequency identical. The voltage waveform, thus making the switching frequency themagnitude resonantoffrequency gain Mvr of the resonant circuit is defined as the ratioand of the T1’s primaryidentical. voltage to The the voltage gain M circuit is and defined the ratioas:of the magnitude of T1 ’s primary voltage to the magnitude of the input voltage, can beasexpressed vr of the resonant magnitude of the input voltage, and can be expressed as: Vp 1 M vr =
Mvr
V AB
=
ω s 1ω r Vp = = r1 + Q hω r − ω s VAB ωs 1+ Q ω − r
2
(5) ωr ωs
Lr Cr Q= √ RLpr /Cr
Q=
(5)
i2 (6)
(6)
R
where Q is the quality factor of the resonant circuit, ωs pdesignates the switching angular frequency, r isisthe and Rp represents equivalent load resistance between whereωQ theresonant qualityangular factorfrequency, of the resonant circuit, ωthe the switching angularthefrequency, s designates primary winding. At steady-state operation, R p can be expressed as [22]: ω is the resonant angular frequency, and R represents the equivalent load resistance between the r
p
primary winding. At steady-state operation, Rp can expressed as [22]: 8n2 Rbe LED Rp =
Rp =
π2
8n2 R LED π2
(7)
(7)
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where R RLED representsthe theequivalent equivalentimpedance impedanceofofthe theLED LEDstring. string.Figure Figure33depicts depictsthe therelationship relationship where LEDrepresents where R LED represents the equivalent impedance of the LED string. Figure 3 depicts the relationship between voltage gain M and frequency ratio f /f based on Equations (5) and (6). It can be seen between voltage gain Mvrvrand frequency ratio fs/frs based on Equations (5) and (6). It can be seen that r between voltage gain M vr and frequency ratio f s /f r based on Equations (5) and (6). It can be seen thatcircuit the circuit exhibits capacitive characteristics when switching frequency smallerthan thanthat the the exhibits capacitive characteristics when thethe switching frequency is issmaller the the circuit exhibits The capacitive characteristics whenthe the switching smaller than the resonant frequency. will feature of when 1.1.As Assoon soonas asthe theswitching switchingfrequency frequency and and the the The The power switchesare will possess ZVS when fs/fr > a1. As soon feature as the switching frequency and can the resonant frequency are equal, the circuit circuit possesses a resistive resistive feature and the the power power switches can resonant frequency equal, the possesses and switches resonant are the circuit possesses a resistive feature and the of power switches can operateat atfrequency both ZCS ZCS and andequal, ZVS. Figure Figure shows the ideally ideally schematic waveforms of the the input input voltage operate both ZVS. 44 shows the schematic waveforms voltage operate at both ZCS and ZVS. Figure 4 shows the ideally schematic waveforms of the input voltage ofthe theresonant resonantcircuit circuitvAB vAB and resonant current ir when fr , that i.e., the thatvoltage the voltage switches to of and thethe resonant current ir when fs = fsr, =i.e., switches to zero of the resonant circuit v AB and the resonant current ir when fs = fr, i.e., that the voltage switches to zero zero the moment the resonant current reaches zero. This makes the power switch undergo no extra the moment the resonant current reaches zero. This makes the power switch undergo no extra the moment the resonant current reaches zero. This makes the power switch undergo no extra switching loss. switching loss. switching loss. 1.0 1.0 0.9 0.9 0.8 ZCS region 0.8 ZCS region 0.7 0.7 Mvr 0.6 0.6 Mvr 0.5 0.5 0.4 0.4 0.3 0.3 0.2 0.2 0.1 0.1 0 0.5 0 0 0.5 0
ZVS region ZVS region
1.0 1.0
1.5
fs1.5 /fr fs/fr
2.0 2.0
2.5 2.5
3.0 3.0
Figure 3. Voltage gain versus frequency ratio. Figure 3. 3. Voltage Voltage gain gain versus versus frequency Figure frequency ratio. ratio.
vAB vAB ir ir
t t
Figure 4. Voltage and current waveforms of the resonant circuit (fs = fr). Figure and current current waveforms waveforms of of the the resonant resonant circuit circuit (f(fs = = fr). Figure 4. 4. Voltage Voltage and s fr ).
2.3. Boost-Typed Power-Factor Corrector 2.3. Boost-Typed Power-Factor Corrector As shown in Figure 1, the boost converter, which is composed of LPFC, S2, DS1, and Cbus, performs As shown in Figure Figure 1,the the boostconverter, converter, whichiscurrent iscomposed composed of LPFC,the ,SS2 ,2PFC ,DDS1 S1, and and C Cbus bus,, performs in 1, boost which LPFC performs the function of PFC. Figure 5 shows the PFC inductor iLPFCofwhen circuit is designed the function of PFC. Figure 5 shows the PFC inductor current i LPFC when the PFC circuit is designed function PFC. Figure 5conduction shows the PFC inductor iLPFC when the of PFC circuit is designedto to to operate at of discontinuous mode (DCM).current Since the rising slope iLPFC is proportional to operate at discontinuous conduction mode (DCM). Since the rising slope of i LPFC is proportional to operate at discontinuous conduction mode (DCM). Since the rising slope of i is proportional to the LPFC the line voltage vin, the envelope of iLPFC would be sinusoidal. The alternating current (ac)/dc converter the voltage in, the envelope iLPFCwould wouldbebesinusoidal. sinusoidal.The Thealternating alternatingcurrent current(ac)/dc (ac)/dc converter lineline voltage vin , vthe envelope iof LPFC was supplied from the ac line of voltage. was supplied from the ac line voltage. voltage. v in ( t ) = V m sin ( 2 π f L t ) (8) v in ( t ) = V m sin ( 2 π f L t ) (8) vin (t) = Vmofsin f L voltage t) (8) (2π the line source, respectively. In practice, where fL and Vm are the frequency and amplitude L and Vm are the frequency and amplitude of the line voltage source, respectively. In practice, where f fL is much lower than the switching frequency, fs, of the active switches. It is reasonable to consider where fL and Vm are the and amplitude of the voltage source, In consider practice, fL isrectified much lower than thefrequency switching frequency, fs, of the line active switches. It 6isrespectively. reasonable the input voltage as a constant over a high-frequency cycle. Figure shows iLPFC intoone highfthe is much lower than the switching frequency, f , of the active switches. It is reasonable to consider L rectified input voltage as a constant over a high-frequency Figure 6the shows iLPFC in frequency cycle where ton is the conduction time ofs switch S2 andcycle. tf represents fall time of one iLPFC.highThe the rectified input voltage constant over high-frequency Figure shows in. The one frequency cycle where tas: on is as theaconduction timeaof switch S2 and tfcycle. represents the6fall timeiLPFC of iLPFC duty ratio D is defined duty ratio D is defined as: t D = ton (9) D = Ton s (9) Ts
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high-frequency cycle where ton is the conduction time of switch S2 and tf represents the fall time of iLPFC . The duty ratio D is defined as: ton D= (9) Appl. 6 6ofof1616 Appl.Sci. Sci.2018, 2018,8,8,x xFOR FORPEER PEERREVIEW REVIEW Ts where Ts is the period of the switching frequency. At DCM operation, the peak value of the inductor where s is whereTT s isthe theperiod periodofofthe theswitching switchingfrequency. frequency.At AtDCM DCMoperation, operation,the thepeak peakvalue valueofofthe theinductor inductor current can be expressed as: current can be expressed as: current can be expressed as:
((
))
sin −−VV sin V(m2(π sin (Vcbus ( 2(π2fπfLftt)|)t )⋅·t⋅ontt VVcbus )t ⋅ tf⋅ t (tt)))t · t f (t) sin sin 2πf L(ft2π sin (2π |V VV m m− L ) fL f ( ) i PK (itPK )iPK= ( t()t )==m m LPFCL L onon=== cbus m LL LL L PFC PFC PFC
(10) (10) (10)
PFC PFC
From bebe described asas Equation (11). From Equation(10), (10),relationship relationshipbetween betweentonton onand and f can described Equation (11). FromEquation andtf ttcan described as Equation (11). f canbe
V ( 2πfLLfLt)t)) t m sin (( 2π t ()t )==V Vm−msin t f (ttft)(f t= sin π2πf LfftL)tt))ontonon V sin((2(2π Vcbus −−VVV sin cbus mmm cbus V sin 2π f t
y 1
iPK (t)(t) iPK
vinvin
y
(11) (11) (11)
iLPFC iLPFC
1
x
ππ
O O
vgs2 vgs2
x
2π2π
-1 -1
Figure 5.5. Schematic waveform ofof a apower-factor Figure Schematic waveform corrector (PFC) inductor current. Figure 5. Schematic waveform of a power-factor power-factorcorrector corrector(PFC) (PFC)inductor inductorcurrent. current.
iPK iPK
ton ton ton ton+t+tf f
TTs s
tt
Figure inductor current Figure6.6. 6.PFC PFC inductor currentinin inone onehigh-frequency high-frequencycycle. Figure PFC inductor current one high-frequency cycle.
The average value of iLPFC over The overa ahigh-frequency high-frequencycycle cycleisisgiven, given,and andcan canbebeexpressed expressedasas[23]: [23]: The average average value value of of iiLPFC LPFC over a high-frequency cycle is given, and can be expressed as [23]:
((
))
2 tonton+ +t ft f(t()t )⋅ i⋅PK iPK( t()t ) DDV2m Vmsin sin( 2(π2πf LftL)t ) 11 iLPFC t = = iLPFC( ()tt on ) =+ t f (t)2T· iPK (t) =D2 V2mL|sin(f2π f L t⋅)|⋅ 1 1 1 2sTs 2PFC LPFC sf s 1 1−· − ⋅ sin i LPFC (t) = = ⋅ sin( 2(π2πf Lft )t ) k1 k− 1 · |sin(L2π f L t)| 2Ts 2L PFC f s
(12) (12) (12)
k
where the index k is defined as: where as: where the the index index kk is is defined defined as:
VVVcbus cbus kkk≡≡ ≡ cbus Vm VV mm
(13) (13) (13)
The input current is equal to the average value of the PFC inductor current, and can be The Theinput inputcurrent currentisisequal equaltotothe theaverage averagevalue valueofofthe thePFC PFCinductor inductorcurrent, current,and andcan canbebeexpressed expressed expressed as: as: as: D2 Vm sin(2π f L t) 1 iin (t) = · (14) 22 1 2L DDV sin PFC V sin(f2(sπ2πf LftL)t ) 1 − k · |1sin 1 (2π f L t)| mm iiniin( t()t = ⋅⋅ ) = 2L f (14) 1 (14) 2PFC LPFC sf s 1 1−− 1⋅ ⋅sin sin( 2(π2πf LftL)t ) kk
The Theinput inputpower powercan canbebeobtained obtainedby bytaking takingthe theaverage averageofofthe theproduct productofofthe theinput inputvoltage voltageand and current currentover overone oneline-frequency line-frequencycycle. cycle.From FromEquations Equations(8) (8)and and(14), (14),the theinput inputpower powercan canbebederived derived asas[24]: [24]:
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The input power can be obtained by taking the average of the product of the input voltage and Appl. Sci. 2018, 8, x FOR PEER REVIEW 7 of 16 current over one line-frequency cycle. From Equations (8) and (14), the input power can be derived as [24]: Z 1 1π π D2VVmm22 Pin =Pin = VmVmsin (15) sin(2π 2πf Lf Lt)t )· ⋅iiinin ((tt))dd(2π 2πf Lf Ltt))= = ⋅ ·yy ( ( (15) π π0 0 LPFC 22L PFCffs s where where yy is is expressed expressedas: as: 2 3 2 2 π −1 1 sin2sin θ θ dθ = kk3 1 + 2 sin = y 0 1 1 dθ = √ k22 − 1 1 +π 2 sin−k 11− k −−kπ2 ⋅−k 2 · k (16) y= (16) π k π 0 1 −1 −sinsin θ θ k −1 k k PFC can From Equations Equations (15) (15) and and (16), (16), the the inductance inductance of of LLPFC From can be be calculated calculated by by using using Equation Equation (17). (17).
Z π
η D 2V 2 LPFC =ηD2 Vmm2 ⋅ y L PFC = 2P f · y 2Po of s s
(17) (17)
where Po and η represent the output power and circuit efficiency, respectively. where Po and η represent the output power and circuit efficiency, respectively. 2.4. Starter Starter Circuit Circuit 2.4. In the the self-excited self-excited circuit, circuit, aa circuit circuit starter starter is is indispensable indispensable to to initialize initialize the the operation. operation. Figure Figure 77 In shows the starter circuit used in the developed LED drive. The starter circuit is connected in parallel shows the starter circuit used in the developed LED drive. The starter circuit is connected in parallel with the dc-bus capacitor C bus with a simple resistor-capacitor (RC) circuit. Prior to the action of the with the dc-bus capacitor Cbus with a simple resistor-capacitor (RC) circuit. Prior to the action of starter, the dc-bus voltage VCbus charges towardtoward Cstart viaCRstart. With the time constant being represented the starter, the dc-bus voltage VCbus charges start via Rstart . With the time constant being by Equation (18), when the exponentially increasing voltage on the starting on capacitor Cstart reaches the represented by Equation (18), when the exponentially increasing voltage the starting capacitor breakover voltage of DIAC, the DIAC conducts and the power switch is triggered. As soon as S2 C start reaches the breakover voltage of DIAC, the DIAC conducts and the power switch is triggered. commences conduct, the resonantthe process begins. However, to avoid VtoCbus continually charging As soon as S2to commences to conduct, resonant process begins. However, avoid VCbus continually C start, a bypass diode, Dbp, is connected across the power switch, S2. Of course, the bypass current charging Cstart , a bypass diode, Dbp , is connected across the power switch, S2 . Of course, the bypass throughthrough Dbp is drastically quantitatively smaller than the resonant preventTo theprevent upper and current Dbp is drastically quantitatively smaller than thecurrent. resonantTocurrent. the lower switches from shooting through, the switching frequency needs to be far greater thanthan the upper and lower switches from shooting through, the switching frequency needs to be far greater reciprocal of the time constant, as shown in Equation (19). the reciprocal of the time constant, as shown in Equation (19). τ = R start C start τ = Rstart Cstart
(18) (18)
1 f s >>1 f s >> ττ
(19) (19)
Starter Circuit Dbp Rstart Cbus
+ _
Vcbus Cstart
+ _
S2 Vcstart
+
DIAC
vgs2
_
Figure 7. Starter circuit. Figure 7. Starter circuit.
3. Circuit Circuit Analysis Analysis and and Operation Operation Modes Modes 3. Figure 88illustrates illustratesthe theschematic schematicvoltage voltage and current waveforms of some components. It Figure and current waveforms of some key key components. It can caninspected be inspected the PFC inductor current, , and the resonant current, ir, flow through be that that bothboth the PFC inductor current, iLPFCi,LPFC and the resonant current, ir , flow through the the power switch S 2. When the input square voltage crosses zero, the resonant current becomes zero power switch S2 . When the input square voltage crosses zero, the resonant current becomes zero as well. well. as The input voltage of the resonant circuit is a square wave, and can be described by the Fourier expansion. 2V v AB ( t ) = cbus sin ( 2 nπ f s t ) n nπ
n = 1, 3, 5...
(20)
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The input voltage of the resonant circuit is a square wave, and can be described by the Fourier expansion. 2Vcbus v AB (t) = ∑ sin(2n π f s t) n = 1, 3, 5 . . . (20) nπ Appl. Sci. 2018, 8, x FOR PEER REVIEW 8 of 16 n When the quality factor of the resonant circuit is high enough, almost all the harmonic contents When the quality factor of the resonant circuit is high enough, almost all the harmonic contents of v will be filtered out by the resonant circuit. Only the fundamental current at the switching of vABABwill be filtered out by the resonant circuit. Only the fundamental current at the switching frequency is present in the load resonant converter [22]. At steady-state operation, the voltage across frequency is present in the load resonant converter [22]. At steady-state operation, the voltage across the secondary winding is equal to the output voltage, and the load (including the output capacitance the secondary winding is equal to the output voltage, and the load (including the output capacitance Co) is fed by a rectified sinusoidal current source. The impedance across the secondary winding is Co) is fed by a rectified sinusoidal current source. The impedance across the secondary winding is equal to the resistance of the LED string (RLED ). Figure 9 shows the equivalent circuit of the resonant equal to the resistance of the LED string (RLED ). Figure 9 shows the equivalent circuit of the resonant circuit, where v represents the fundamental voltage of v . From Equation (20), the root-mean-square circuit, where v11represents the fundamental voltage of vABAB . From Equation (20), the root-mean-square (rms) value of the fundamental voltage is equal to: (rms) value of the fundamental voltage is equal to: √ 2V cbus 2Vcbus V1V= (21) (21) 1= π π Theimpedance impedanceof ofthe theresonant resonantcircuit circuitcan canbe beexpressed expressedas: as: The 11 Z AB = =RR 2πf sf sLLr r−− Z AB p p++j j 2π 2π f C 2π ss rr
(22) (22)
Underthis thiscircumstance, circumstance,the theresonant resonantcurrent currentisissinusoidal, sinusoidal,and andcan canbe beexpressed expressed as: as: Under
2Vcbus 2V sin((22ππf sfts )t) sin ir (itr)( t=) = cbus πZ π ZAB AB
(23) (23)
FromEquation Equation(23), (23),the theresonant resonantcurrent currentwould wouldtheoretically theoreticallybe bezero zeroininthe thecase caseofoffailure failureand and From theload loadwould wouldgo goopen opencircuit. circuit.However, However,ininpractical practicalterms, terms,aasmall smallcurrent currentisisthere thereififthe themutual mutual the inductance of of the the transformer transformer is in in thethe case of failure where the load goes inductance is considered. considered.On Onthe thecontrary, contrary, case of failure where the load short circuit, the primary winding and the excited windings will all become short-circuited. Thus, goes short circuit, the primary winding and the excited windings will all become short-circuited. both power switches will not energized and the will remain turned off. Besides, the LED Thus, both power switches willbe not be energized andcircuit the circuit will remain turned off. Besides, the current is not in the circuit sincesince a self-excited control mechanism is used. LED current is regulated not regulated inproposed the proposed circuit a self-excited control mechanism is used.
Figure Figure8.8.Schematic Schematicvoltage voltageand andcurrent currentwaveforms waveformsofofsome somekey keycomponents. components.
Lr
v1
Cr
ir
vp ZAB
+
Rp _
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Figure 8. Schematic voltage and current waveforms of some key components.
Cr
Lr
v1
ir
vp
+
Rp _
ZAB
Figure Figure9. 9.Equivalent Equivalentcircuit circuitto tothe theload loadresonant resonant circuit. circuit.
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At At a steady state, thethe circuit operation cancan be divided intointo three modes in every high-frequency a steady state, circuit operation be divided three modes in every high-frequency cycle. TheThe current loops of each operation mode areare shown in in Figure 10,10, where vrecvrepresents thethe cycle. current loops of each operation mode shown Figure where rec represents rectified voltage of the input line.line. As soon as theaslower-arm powerpower switchswitch S2 starts the rectified voltage of the input As soon the lower-arm S2 successfully, starts successfully, circuit operation initiates. The operating modesmodes are described below.below. the circuit operation initiates. The operating are described Cr
Lr
D1 Co
RL ED
LPFC vre c
S2 vgs2
Cbu s
vgs2
(a)
Cr
Lr S1 LPFC vre c
Co
RL ED
Co
RL ED
vgs1 vgs1
Cb us
D2 (b)
Cr
Lr S1 vgs1 vgs1
D2
(c) Figure 10. Equivalent circuit in different operation modes: (a) Mode I, (b) Mode II, and (c) Mode III. Figure 10. Equivalent circuit in different operation modes: (a) Mode I, (b) Mode II, and (c) Mode III.
3.1. Operation Mode I (t0 − t1) In this mode, S2 conducts, S1 is off, and the rectified input voltage vrec is across the boost inductor LPFC. The PFC current iLPFC flows through S2 and increases linearly, as written in Equation (24).
iLPFC ( t ) =
Vm sin ( 2π f L t ) LPFC
( t − t0 )
(24)
In addition, the dc-bus voltage Vcbus is across the input of the resonant circuit and supplies a
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3.1. Operation Mode I (t0 − t1 ) In this mode, S2 conducts, S1 is off, and the rectified input voltage vrec is across the boost inductor LPFC . The PFC current iLPFC flows through S2 and increases linearly, as written in Equation (24). i LPFC (t) =
Vm |sin(2π f L t)| ( t − t0 ) L PFC
(24)
In addition, the dc-bus voltage Vcbus is across the input of the resonant circuit and supplies a nearly sinusoidal resonant current, ir . The ir is positive and energizes the center-tapped transformer, as indicated in Figure 10a. 3.2. Operation Mode II (t1 − t2 ) The equivalent circuit of Mode II is shown in Figure 10b. This mode starts as soon as the resonant current, ir (via the center-tapped transformer) decreases to zero, and the primary winding then induces a back electromotive force (emf) and result in the conduction of power switch S1 , along with the turn-off of switch S2 . The current iLPFC reaches a peak value at the beginning of Mode II. Since S2 is off, the current iLPFC diverts from S2 to DS1 to charge the dc-bus capacitor Cbus . The voltage across LPFC is equal to Vcbus minus vrec , which is negative, meaning that iLPFC decreases linearly. i LPFC (t) =
V − Vm |sin(2π f L t)| Vm |sin(2π f L t)| DTs − cbus ( t − t1 ) L PFC L PFC
(25)
In this mode the current ir is negative. Simultaneously, the resonant tank releases energy to the load. 3.3. Operation Mode III (t2 − t3 ) The equivalent circuit of Mode III is shown in Figure 10c. This mode starts when the current iLPFC decreases to zero. In this mode, S1 keeps conducting. The resonant current continually supplies energy via the center-tapped transformer to the output. 4. Circuit Implementation Based on the above-mentioned circuit analysis and mathematical derivations, a 40 W, 110 Vac/95 Vdc half-bridge, self-excited, single-stage LED driver was implemented and realized. Table 1 outlines the circuit specification. The design criteria are briefly described as follows. 1.
2.
3.
4.
The PFC current iLPFC is designed to operate in boundary conduction mode (BCM) when the input voltage gets near the peak value. By this way, the PFC converter would operate at DCM over the whole input line cycle. By using this self-excited control mechanism, the duty ratio D of both MOSFETs is 0.5. The dc-bus voltage Vcbus should be sufficiently high to ensure the PFC converter operate at DCM. Theoretically, Vcbus should be higher than two times of the amplitude of the input line voltage at 0.5 duty ratio. In this illustrative example, VCbus is designed to be 320 V. Due to the self-excited mechanism, the driving voltages are fetched from the center-tapped transformer by means of the resonant current. The switching frequency is in synchronism with the resonant frequency. The resonant frequency is controlled by the selection of the resonant inductor and the resonant capacitor as is designed with 36 k Hz resonant frequency. The higher resonant frequency will detrimental to self-excited signal response. It will increase the rise and fall times of switching current, thus leading to extra switching loss. Following determination of the resonant frequency fr , only one single resonant capacitor is used with capacitance 10 nF. Accordingly, the resonant inductance can be achieved by Equation (1).
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5.
6.
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Substituting input voltage Vin , dc-bus voltage VCbus and switching frequency fs into Equations (13), (16) and (17), the PFC inductor LPFC can be obtained by the aid of numerical computation. Since all the transferred power goes through the resonant capacitor, it is suggested to use a capacitor with low equivalent series resistance (ESR). Table 1. Circuit Specification. Input Voltage vin
110 Vrms , 60 Hz
DC-bus voltage Vcbus Output voltage Vo Output Current Io Switching frequency fs
320 Vdc 95 V 0.42 A 36 kHz
5. Experimental Results A prototype circuit was built and tested. The circuit parameters are shown in Table 2, and the prototype circuit is shown in Figure 11. The experimental apparatuses consisted of an oscilloscope (Lecroy HDO4034, New York, NY, USA), voltage probes (Cybertek DP6150A, Shenzhen, China), Appl. Sci. 2018, 8, x FOR PEER REVIEW 11 of 16 current probes (Cybertek CP8030A, Shenzhen, China), and a power quality analyzer (Precision International Corp.(Cybertek PA-750B,CP8030A, New Taipei City, Taiwan). 12 quality shows analyzer the measured voltage current probes Shenzhen, China), andFigure a power (Precision International PA-750B, New Taipei City, Taiwan). Figure 12ofshows the measured transformer. voltage waveform on the Corp. primary winding and the self-excited windings the center-tapped thethe primary windingvoltage and the self-excited windings the center-tapped transformer. It vp . It canwaveform be seen on that self-excited vgs1 and vgs2 are inofphase with the primary voltage can be seen that the self-excited voltage v gs1 and vgs2 are in phase with the primary ◦ voltage v p. Obviously, the driving voltages vgs1 and vgs2 are symmetrically displaced by 180 . Hence, the Obviously, the driving voltages v gs1 and vgs2 are symmetrically displaced by 180°. Hence, the switching switching frequency of the power switch could be the same as the resonant frequency of the resonant frequency of the power switch could be the same as the resonant frequency of the resonant converter. converter. Figure 13 shows the input voltage of the resonant converter vAB and the resonant current ir . Figure 13 shows the input voltage of the resonant converter vAB and the resonant current ir. It It demonstrates that ir is in phase with vAB . This marks the equality of the resonant frequency and demonstrates that ir is in phase with vAB. This marks the equality of the resonant frequency and switching frequency, i.e.,i.e., fr =frf=s . fs. switching frequency, Table Table2.2.Circuit Circuit parameters. parameters. 1.81.8 mH mH 0.47 μF 0.47 µF 100 μF 100 µF 50 μF 50 µF 1.31.3 mH mH 1.81.8 mH mH 1010 nFnF PP == 6464 turns, NN S1 S1 =N NN turns, =S2N=S240= turns 40 turns Turns of the transformer Turns of center-tapped the center-tapped transformer = SE2 NSE2 6 turns SE1 =N = 6 =turns NN SE1 Power Switches S1 , SS21, S2 IRF840 Power Switches IRF840 Low-Pass Inductor Low-Pass Inductor Lm Lm Low-pass capacitor Cm Low-pass capacitor Cm DC-bus capacitor Cbus DC-bus capacitor Cbus Output capacitor Co Output capacitor Co Boost inductor Boost inductor LPFCLPFC Resonant inductor Resonant inductor Lr Lr Resonant capacitor Resonant capacitor Cr Cr
Figure 11. Prototype Circuit.
Figure 11. Prototype Circuit.
Appl. Sci. 2018, 8, 1408
Figure 11. Prototype Circuit.
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Figure 12. Voltage waveforms of the center-tapped transform. (vp: 100 V/div, vgs1, vgs2: 20 V/div, time: Figure 12. Voltage waveforms of the center-tapped transform. (vp : 100 V/div, vgs1 , vgs2 : 20 V/div, time: Appl.10 Sci.μs/div). 2018, 8, x FOR PEER REVIEW 12 of 16 10 µs/div).
Figure 13. Voltage and current waveforms of the resonant converter. (vAB : 100 V/div, ir : 0.2 A/div, Figure Voltage and current waveforms of the resonant converter. (vAB: 100 V/div, ir: 0.2 A/div, time: time: 1013. µs/div). 10 μs/div).
Figure 14 shows the waveforms of the input line voltage and the PFC inductor current iLPFC . Figure 14 shows the waveforms of the input line voltage and the PFC inductor current iLPFC. It It reveals that the envelope of iLPFC is in phase with the low-frequency input line voltage. Figure 15 reveals that the envelope of iLPFC is in phase with the low-frequency input line voltage. Figure 15 shows the waveforms of iLPFC and the voltage across the power switch S2 . It indicates that the PFC shows the waveforms of iLPFC and the voltage across the power switch S2. It indicates that the PFC converter operates at BCM when the input line voltage is at the peak value, while it operates at DCM converter operates at BCM when the input line voltage is at the peak value, while it operates at DCM elsewhere. The waveforms of the input line voltage and current are shown in Figure 16. It can be seen elsewhere. The waveforms of the input line voltage and current are shown in Figure 16. It can be seen that the input current is in phase with the line voltage. The measured unity power factor and total that the input current is in phase with the line voltage. The measured unity power factor and total current harmonic distortion (THDi) is 0.98 and 10.2%, respectively. It is seen that the input current current harmonic distortion (THDi) is 0.98 and 10.2%, respectively. It is seen that the input current is is close to a triangle waveform. In this proposed circuit, the power switches are operated at a fixed close to a triangle waveform. In this proposed circuit, the power switches are operated at a fixed switching frequency and fixed duty ratio (D = 0.5). Since the boost-typed PFC operates at DCM, the switching frequency and fixed duty ratio (D = 0.5). Since the boost-typed PFC operates at DCM, the peak value of the inductor current is proportional to the rectified input voltage, i.e., the envelope of the peak value of the inductor current is proportional to the rectified input voltage, i.e., the envelope of inductor current would be a sinusoidal waveform. Theoretically, the average value of the rising portion the inductor current would be a sinusoidal waveform. Theoretically, the average value of the rising of the inductor current within a high-frequency cycle is also sinusoidal. Nevertheless, the duration portion of the inductor current within a high-frequency cycle is also sinusoidal. Nevertheless, the of the inductor current dropping from peak to zero is not constant. The higher the input voltage, duration of the inductor current dropping from peak to zero is not constant. The higher the input the longer the declining duration. When the PFC operates at the time when the input voltage is at the voltage, the longer the declining duration. When the PFC operates at the time when the input voltage peak of a sinusoidal waveform, the average value of the inductor current would be much higher than is at the peak of a sinusoidal waveform, the average value of the inductor current would be much higher the peak. This is why the current waveform looks triangular. than the peak. This is why the current waveform looks triangular.
vin
the inductor current would be a sinusoidal waveform. Theoretically, the average value of the rising portion of the inductor current within a high-frequency cycle is also sinusoidal. Nevertheless, the duration of the inductor current dropping from peak to zero is not constant. The higher the input voltage, the longer the declining duration. When the PFC operates at the time when the input voltage is at the a sinusoidal waveform, the average value of the inductor current would be much higher Appl. Sci. peak 2018, 8,of1408 13 of 17 than the peak. This is why the current waveform looks triangular.
vin
iLPFC
Time: 5 ms/div
Figure 14. of thethe input voltage and and the PFC inductor current. (vin: 100(vV/div, iLPFC: 1.0 A/div, Figure 14.Waveforms input voltage the PFC inductor current. in : 100 V/div, iLPFC :13 of 16 Appl. Sci. 2018, 8,Waveforms x FOR PEER of REVIEW time: 5 ms/div). 1.0 A/div, time: 5 ms/div).
vDS2
Time: 2 ms/div
iLPFC
BCM
Time: 10 us/div
(a)
vDS2
Time: 2 ms/div iLPFC
Time: 10 us/div
(b) Figure 15. Waveforms of the PFC inductor current and the voltage across S2 at (a) the peak value, and Figure 15. Waveforms of the PFC inductor current and the voltage across S2 at (a) the peak value, (b) near 135 degrees of the input line voltage. (vDS2: 100 V/div, iLPFC: 0.5 A/div). and (b) near 135 degrees of the input line voltage. (vDS2 : 100 V/div, iLPFC : 0.5 A/div).
vin
iin
Time: 10 us/div
(b) Waveforms of the PFC inductor current and the voltage across S2 at (a) the peak value, and Appl. Sci.Figure 2018, 8,15. 1408 14 of 17 (b) near 135 degrees of the input line voltage. (vDS2: 100 V/div, iLPFC: 0.5 A/div).
vin
iin
Figure Waveformsof of the the input and current. (vin: (v 100 :V/div, iin: 0.5 A/div, time: 5 ms/div). Figure 16.16.Waveforms inputline linevoltage voltage and current. in 100 V/div, iin : 0.5 A/div, time: 5 ms/div).
Figure 17 shows the voltage and current waveforms of the power switches. Since the switching frequency is identical to the resonant frequency, S1 can be turned off at zero current. Besides, as Figure 17 shows the voltage and current waveforms of the power switches. Since the switching mentioned in operation Mode II, the PFC inductor current diverts from S2 to DS1 as soon as S2 is turned frequency is identical to the resonant frequency, S1 can be turned off at zero current. Besides, off. In this way, the voltage across S1 is clamped to near zero volts. It helps the power switch S1 to be as mentioned in operation Mode II, the PFC inductor current diverts from S2 to DS1 as soon as turned on at zero voltage. Theoretically, the power switch S2 should be turned on at zero voltage S2 is turned off. In this way, the voltage across S1 is clamped to near zero volts. It helps the power when the resonant circuit presents a resistive characteristic. Nevertheless, it is shown that a spike switch S1 to be turned on at zero voltage. Theoretically, the power switch S2 should be turned on at current happens when S2 is turned on, meaning that S2 does not achieve ZVS operation. zero voltage when the resonant circuit presents a resistive characteristic. Nevertheless, it is shown that a Appl. spikeSci. current when S2 is turned on, meaning that S2 does not achieve ZVS operation.14 of 16 2018, 8, happens x FOR PEER REVIEW vDS1
ZVS ZCS
iS1 vDS2
iS2
Figure17.17.Voltage Voltageand andcurrent current waveformsofofthe the power switches.(v(vDS1,, vvDS2:: 200 V/div, iS1, iS2: 1.0 Figure waveforms power switches. DS1 DS2 200 V/div, iS1 , iS2 : A/div, time: 10 μs/div). 1.0 A/div, time: 10 µs/div).
At the output side of the developed circuit, the ZCS performance has been accomplished by At the output side of the developed circuit, the ZCS performance has been accomplished by inspecting the relationship between diode voltage waveforms and diode current waveforms on both inspecting the relationship between diode voltage waveforms and diode current waveforms on both D1 and D2. Figure 18 obviously shows that the current of the output diodes naturally fall to zeros D1 and D2 . Figure 18 obviously shows that the current of the output diodes naturally fall to zeros prior to the moment that the transformer secondary windings switch. It means that diodes D1 and D2 prior to the moment that the transformer secondary windings switch. It means that diodes D1 and D2 can operate at ZCS, resulting in a zero-recovery current and low turning-off losses. Figure 19 shows can operate at ZCS, resulting in a zero-recovery current and low turning-off losses. Figure 19 shows the waveforms of the output voltage and current, and as displayed, there are both high-frequency the waveforms of the output voltage and current, and as displayed, there are both high-frequency and low-frequency ripple components in the voltage and current. The measured average voltage and and low-frequency ripple components in the voltage and current. The measured average voltage average current are 95 V and 0.42 A, respectively. The circuit efficiency is 92.2%. The measured values and average current are 95 V and 0.42 A, respectively. The circuit efficiency is 92.2%. The measured are consistent with the theoretical prediction, and the peak-to-peak voltage ripple is 560 mV, while values are consistent with the theoretical prediction, and the peak-to-peak voltage ripple is 560 mV, the peak-to-peak current ripple is 46 mA. The ripple factors of the voltage and current are 0.6% and while the peak-to-peak current ripple is 46 mA. The ripple factors of the voltage and current are 0.6% 10.9%, respectively. and 10.9%, respectively. vD1
iD1 ZCS
the waveforms of the output voltage and current, and as displayed, there are both high-frequency the waveforms of the output voltage and current, and as displayed, there are both high-frequency and low-frequency ripple components in the voltage and current. The measured average voltage and and low-frequency ripple components in the voltage and current. The measured average voltage and average current are 95 V and 0.42 A, respectively. The circuit efficiency is 92.2%. The measured values average current are 95 V and 0.42 A, respectively. The circuit efficiency is 92.2%. The measured values are consistent with the theoretical prediction, and the peak-to-peak voltage ripple is 560 mV, while are consistent with the theoretical prediction, and the peak-to-peak voltage ripple is 560 mV, while the Sci. peak-to-peak current ripple is 46 mA. The ripple factors of the voltage and current are 0.6% and Appl. 2018, 8, 1408 current 15 of 17 the peak-to-peak ripple is 46 mA. The ripple factors of the voltage and current are 0.6% and 10.9%, respectively. 10.9%, respectively.
vD1 vD1
iD1 iD1 ZCS ZCS
iD2 iD2
vD2 vD2 ZCS ZCS
Figure 18. Voltage and current waveforms of the output diodes. (vD1, vD2: 100 V/div, iD1, iD2: 0.5 A/div, Figure Voltageand andcurrent currentwaveforms waveforms output diodes. , vD2 : 100 iD1 V/div, iD1A/div, , iD2 : D1 Figure 18. 18. Voltage of of thethe output diodes. (vD1, (v vD2 : 100 V/div, , iD2: 0.5 time: 10 μs/div). 0.5 A/div, time: 10 µs/div). time: 10 μs/div).
Vo = 95 V Vo = 95 V
Io = 0.42 A Io = 0.42 A
Time: 20 ms/div Time: 20 ms/div Figure 19. Waveforms of the output voltage and current. (vAB: 50 V/div, ir: 0.2 A/div, time: 20 ms/div). Figure19. 19.Waveforms Waveformsofofthe theoutput outputvoltage voltage and current.(v(v 50V/div, V/div,iirr:: 0.2 A/div, A/div, time: 20 ms/div). Figure and current. : :50 ms/div). ABAB
6. Conclusions In this research, a synchronized, self-excited control for a single-stage LED driver has been developed. The proposed circuit topology was derived by integrating a boost converter which functioned as a PFC, as well as a series resonant converter outputting a dc voltage to drive an LED string, which helps to save the number of power switches and their corresponding control circuits. In using this novel, self-excited driving approach, the proposed circuit does not need to use an integrated circuit. In addition, the components used to build a dc voltage for an integrated circuit could be saved. Because the driving voltages of the power switches are synchronized with the resonant current of the resonant circuit, the switching frequency and resonant frequency are identical, meaning that the power switch of the upper arm can operate at both ZVS and ZCS. However, the power switch of the lower arm cannot operate at ZCS, since it is shared by the PFC converter to help both the PFC inductor current and the resonant current to flow. Besides, the rectifying diodes at the output side possess the zero current cut-off characteristics. The high power factor and low THDi allow, by design, the boost-typed PFC converter to operate at DCM. The measured power factor and THDi are 0.98 and 10.2%, respectively, and the overall performance efficiency is 92.2%. Author Contributions: Y.-N.C. conceived and designed the circuit; H.-L.C. performed circuit simulations and designed parameters of the circuit components; S.-Y.C. wrote the paper and H.-L.C. revised it for submission. Funding: The research received no external funding. Acknowledgments: This work was supported by the Ministry of Science and Technology, R.O.C. under Grant MOST 104-2622-E-150-005-CC3. Conflicts of Interest: The authors declare no conflict of interest.
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