Title: A Single-Stage Single-Switch Soft-Switching Power-Factor-Correction LED Driver Authors: Behzad Poorali, Student Member, IEEE, Ehsan Adib, Member, IEEE, and Hosein Farzanehfard, Member, IEEE Affiliation: The authors are with the Department of Electrical and Computer Engineering, Isfahan University of Technology, Isfahan, 84156-83111, Iran. Corresponding Author: Assoc. Prof. Ehsan Adib Postal Address: Department of Electrical and Computer Engineering, Isfahan University of Technology, Isfahan, 84156-83111, Iran Voice Telephone Number: +98-313-3915465 email:
[email protected] This manuscript has not been presented at a conference. “The answers to the reviewers’ comments are added at the end of the manuscript”
Abstract: This paper proposes a new isolated single-stage single-switch soft-switching powerfactor-correction (S6-PFC) driver for supplying light-emitting-diodes (LEDs) without electrolytic capacitors. In the proposed LED driver, all the semiconductor devices are soft-switched without employing any extra active switch. Leakage inductance of the transformer acts as a resonant component so that the leakage energy is recycled. Soft-switching operation of the semiconductor devices together with absorption of the leakage energy improve the proposed driver efficiency. Furthermore, input current of the driver has very low harmonics resulting in high power-factor. Operating principles of the proposed S6-PFC LED driver as well as its design guidelines are presented. Also, experimental results of a laboratory prototype for supplying a 21W/30V LED module from 220V/50Hz AC mains are provided to verify the theoretical analysis. The presented results show that the implemented prototype has an efficiency of 92% under full-load condition and its input current total harmonic distortion (THD) is as low as 2.6%. Keywords: LED driver, power-factor-correction, single-stage, single-switch, soft-switching
1. Introduction Advantages of light-emitting-diodes (LEDs), such as high light efficacy, long lifetime, low maintenance requirements, small sizes, and being mercury-free, have resulted in increasing their utilization in lighting applications [1], [2]. Some of the applications in which LEDs are recently being used include street lighting, traffic lighting, automotive lighting, decorative lighting, etc. Approval of some regulations (such as IEC 61000-3-2) for limiting input current harmonics of power supplies has influenced on power-factor (PF) requirements of AC-DC power converters [3]. Thus, power-factor-correction (PFC) techniques are normally applied to improve the input current harmonic contents of power supplies. In preliminary approaches, a basic DC-DC converter (such as boost or flyback converters) is employed as AC-DC stage for PF improvement [4]-[6]. In this simple method, however, output capacitor has significant voltage ripple at twice-line-frequency which is normally eliminated using a second stage. This two-stage structure has at least two active switches and requires two distinct control circuits. Therefore, it is not suitable for low power applications. For increasing the efficiency of two-stage PFC converters, parallel structures have been proposed that process the majority of power once and the remainder twice [7]-[9]. Despite of efficiency improvement, using the parallel PFC structures due to relatively high complexity is not suitable for low power circuits. Combining the components between two stages leads to a simpler structure as single-stage PFC converter. On the other hand, integrating the two stages imposes high current stress to the converter switch. However, owing to low complexity and cost, it is an appropriate choice for low power applications. Therefore, many investigations have been performed to improve the performance of single-stage PFC converters [10]-[23]. In [12], a single-stage PFC converter, which integrates buck-boost and buck converters, has
been introduced. Integration of a dual buck-boost converter and a modified bridgeless PFC converter with half-bridge LLC resonant converter have been suggested for street-lighting driver in [13] and [14], respectively. These converters are not economically suitable for low power applications due to employment of two active switches. An integrated buck-flyback converter has been proposed for power supply and LED driver applications [16], [17]. Also, [20] presents an integrated double-buck converter. Although these converters have simple structures, zero-crossing distortion exists in their input current which increases the current harmonics. Another single-stage LED driver has been proposed based on class-E converter which has one switch and very low input current harmonics [22]. However, all of its semiconductor devices are not soft-switched. An integrated SEPIC-flyback converter has been suggested in [23] for LED driver application. Despite the switch of this converter turns on under ZCS condition, switching losses at turn-off instants can limit the efficiency. In this paper, a new isolated single-stage single-switch soft-switching PFC (S6-PFC) converter as an LED driver is introduced which does not use any electrolytic capacitors. Switch of the proposed driver turns on and off under zero-current-switching (ZCS) and zero-voltage-zerocurrent-switching (ZVZCS) conditions, respectively. Soft-switching operation without using any extra active switch results in low switching losses. Furthermore, employment of the transformer leakage inductance causes the leakage energy to be recycled. Moreover, the proposed LED driver has very low input current harmonics which makes it suitable for high PF applications. Other sections of the paper have been organized as follows: Section 2 introduces the proposed LED driver and presents its operating principles. Its analysis and design guidelines are discussed in section 3 and experimental results of an implemented prototype of the proposed driver are provided in section 4. Finally, section 5 presents the conclusions.
2. Proposed LED Driver and its Operating Principles A single-switch soft-switched isolated DC-DC converter was introduced in 2012 which uses the transformer parasitic elements as resonant components [24]. Employment of this converter as an LED driver was also proposed in 2014 [25]. Fig. 1 illustrates the derivation of the proposed S6PFC LED driver by integrating a buck-boost converter (PFC cell) with the mentioned single-switch soft-switched resonant DC-DC converter. Fig. 1(a) shows the rearranged topologies of buck-boost converter and the resonant converter proposed in [25]. The resonant converter uses a unidirectional switch operating under ZCS condition. These converters can be combined such that both of them share a common switch. Since in the integrated structure, currents of both converters flow through the common switch, its ZCS turn-off condition is lost. To fix this issue, a regular bidirectional
(a)
(b) Fig. 1. Circuit topology and derivation of the proposed LED driver. (a) Derivation of the proposed topology. (b) Proposed LED driver.
switch can be employed such that current of the resonant converter can continue its resonance in reverse direction. This current reduces the switch total current so that when its current becomes zero, switch can be turned off under zero-current. However, full-cycle resonance results in energy reversal back to the input source and, consequently, lower energy can be transferred to output. This problem can be alleviated by inserting a small capacitor and a parallel diode in series with the resonant inductor which increase the resonant characteristic impedance in reverse direction. The final circuit topology of the proposed LED driver is shown in Fig. 1(b). Components of the buck-boost converter and the modified resonant DC-DC converter are demonstrated in this figure. PFC cell of the proposed driver operates in discontinuous-conduction-mode (DCM), thereby a near-unity PF is attained. The proposed LED driver has nine operating modes within one switching cycle in steadystate condition. Some assumptions are made in order to explain the operating principles: All the semiconductor devices are assumed ideal; resonant capacitors Cr1 and Cr2 are identical and equal to Cr; capacitors CB and Co are large enough that their voltage ripples are negligible; and switching frequency fs is much larger than line frequency fl such that the input voltage is considered constant during each switching period Ts. Equivalent circuit topologies and theoretical waveforms of the proposed LED driver during a switching cycle are illustrated in Figs. 2 and 3, respectively. For convenience, in the equivalent circuits shown in Fig. 2, a rectified sinusoidal voltage source Vin_rect and a series diode Dr have been employed instead of input AC source Vin, input filter Lf and Cf, and bridge diodes Dr1 to Dr4. Also, the transformer T has been modeled by a magnetizing inductance Lm, a leakage inductance Llk, and an ideal transformer with turns ratio of n equal to Np/Ns.
(a)
(b)
(c)
(d)
(e)
(f)
(g)
(h)
(i) Fig. 2. Equivalent circuit topology of each operating mode. (a) Mode 1. (b) Mode 2. (c) Mode 3. (d) Mode 4. (e) Mode 5. (f) Mode 6. (g) Mode 7. (h) Mode 8. (i) Mode 9.
Fig. 3. Theoretical waveforms of the proposed LED driver during a switching cycle.
Before mode 1, it is assumed that all the semiconductor devices are off and voltages of the resonant capacitors Cr1 and Cr2 are equal to -V1 and V2, respectively. Mode 1 [t0-t1]: This mode begins when the switch S turns on. The voltage Vin_rect is applied to the inductor L, so iL starts increasing linearly from zero. Also, a resonance occurs between Lr, Lm, Cr1, and Cr2 which makes vCr2 reduce to zero. Smooth increase of the currents through the inductors ensures ZCS turn-on condition of switch S. If Lm is assumed large enough, its current will be much smaller than the resonant current iLr and it can be neglected. Therefore, resonant capacitors Cr1 and Cr2 would be in series. Accordingly, the effect of Lm on resonant currents and voltages within other operating modes is ignored as well. During this mode, iLr and vCr2 can be expressed as i Lr t
V 1 V 2 V B
v Cr 2 t V 2
sin
2 Z r1
2 r 1 t t 0
V 1 V 2 V B 1 cos 2
,
(1)
2 r 1 t t 0
(2)
where ωr1 and Zr1 are the resonant frequency and characteristic impedance of Lr and Cr, respectively. According to (1) and (2), voltage across Cr2 will become zero at the same time that iLr has reached to its peak, if V2 is equal to V1+VB. Under this condition, iLr, vCr1 and vCr2 are simplified to i Lr (t )
2V2 sin Z r1
v Cr 1 (t ) V 2 1 cos
v Cr 2 t V 2 cos
2 r 1 t t 0 ,
2 r 1 t t 0 V 1 ,
2 r 1 t t 0
.
(3) (4) (5)
This mode ends when the voltage vCr2 reaches zero. Mode 2 [t1-t2]: At the beginning of this mode, D2 turns on under zero-voltage-switching (ZVS) condition. As a result, the resonance between Lr and Cr1 continues and vCr1 increases. When vCr1 reaches nVo, voltage of the output diode becomes zero and this mode ends. During this mode,
iLr and vCr1 are given by i Lr t
2V2 cos r 1 t t1 , Z r1
(6)
v Cr 1 t V B 2 V 2 sin r 1 t t1 .
(7)
Mode 3 [t2-t3]: The output diode starts conducting under ZVZCS condition and Llk enters the resonance. By solving the circuit’s equations, iLr is obtained as
Leq
nV o V B Leq
Lr
r 3 L r 2
i Lr t 1
i Lr t 2
sin r 3 t t 2
Leq Lr
i Lr t 2 cos r 3 t t 2
nV o V B L r n 2 L lk
t t 2
(8)
where Leq is the equivalent inductance of parallel connection of Lr and n2Llk, and ωr3 is the resonant frequency of Leq and Cr. If the equivalent inductance of Llk transferred to primary-side of the transformer is much larger than Lr (n2Llk >> Lr), resonant current given in (8) would be simplified to i Lr t
nV o V B Z r1
sin r 1 t t 2 i Lr t 2 cos r 1 t t 2
nV o V B t t 2 . n 2 Llk
(9)
This equation represents that under the condition n2Llk >> Lr, iLr consists of a resonant component caused by the resonance between Lr and Cr1, and a linear component which is the current of Llk transferred to primary-side of the transformer. If n is large enough, linear component of iLr can be ignored. On the other hand, voltage across the capacitor Cr1 is calculated as v Cr 1 t V B Z r 1i Lr t 2 sin r 1 t t 2 nV o V B cos r 1 t t 2 .
(10)
This mode continues until the resonant current iLr reaches zero. Mode 4 [t3-t4]: Direction of the current iLr is inversed and D2 turns off under ZVZCS condition. So, the resonance continues through Cr2. By solving the circuit’s equations during this mode and using the assumption n2Llk >> Lr, iLr is obtained as i Lr t
v Cr 1 t 3 V B 2 Z r1
sin
2 r 1 t t 3
i Llk t 3 2n
cos
2 r 1 t t 3
i Llk t 3 2n
cos r 2 t t 3 2
(11)
where ωr2 is the resonant frequency of n2Llk and Cr. According to (11), iLr has two frequency
components. However, the second and third terms of (11) are negligible by assuming n to be large enough. Therefore, the resonant currents and voltages can be approximately expressed as i Lr t
v Cr 1 t V B v Cr 2 t
v Cr 1 t 3 V B
sin
2 Z r1
v Cr 1 t 3 V B 2
v Cr 1 t 3 V B 2
2 r 1 t t 3
1 cos
1 cos
,
(12)
(13)
2 r 1 t t 3 ,
(14)
2 r 1 t t 3 ,
i Llk t i Llk t 3 cos r 2 t t 3 . 2
(15)
The negative current of iLr results in reduction of current through the switch. When magnitude of iLr becomes equal to iL, the switch current reaches zero and this mode ends. Mode 5 [t4-t5]: When magnitude of iLr becomes greater than iL, body diode of the switch DS turns on. The resonance between Lr, Cr1, Cr2, and Llk continues until the current of DS becomes zero. Resonant currents and voltages follow the equations given in previous mode. During this mode, switch S can be turned off under ZVZCS condition. Mode 6 [t5-t6]: Faster reduction of iLr compared to iL provides ZCS turn-on condition for D1. At the end of this mode, iLr reaches zero and vCr2 increases to the voltage V2. Mode 7 [t6-t7]: During this mode, vCr1 decreases due to the resonance of Cr1 and Llk. The resonance continues until iLlk reaches zero and the output diode turns off under ZCS condition. Resonant equations within this mode are given by n nV o v Cr 1 t 6 i Llk t sin r 2 t t 6 i Llk t 6 cos r 2 t t 6 , Z r2
v Cr 1 t nV o
Z r 2i Llk t 6 n
sin r 2 t t 6 nV o v Cr 1 t 6 cos r 2 t t 6
(16) (17)
where Zr2 is the resonant characteristic impedance of n2Llk and Cr. Mode 8 [t7-t8]: The current iL reduces linearly until it reaches zero. On the other hand, Cr1 is charged by current of the transformer magnetizing inductance which makes vCr1 rise linearly. This
mode ends when iL becomes zero and D1 turns off under ZCS condition. Mode 9 [t8-t9]: All the semiconductor devices are off during this mode and vCr1 still rises linearly until it reaches -V1.
3. Analysis and Design Considerations In this section, the proposed LED driver is analyzed in detail and design guidelines are also presented. The line voltage is assumed to be a sinusoidal waveform with amplitude Vm and angular frequency ωl. Considering that the input filter eliminates the high frequency harmonics of the PFC cell’s input current, the line current Iin is the average of input current pulses over each switching period as I in (t )
1 1 I Lp t t rL Ts 2
(18)
where ILp is the peak current of inductor L in each switching cycle and trL is the rise-time of iL, i.e., the time duration of modes 1 to 5. ILp can be written as I Lp t
V m t rL sin(l t ) . L
(19)
According to Fig. 3, trL is slightly greater than on-time duration of the switch. However, for simplicity, trL can be assumed equal to DTs, where D is the switch duty cycle. Therefore, Iin is obtained using (18) and (19) as I in (t )
D 2 Ts V m sin(l t ) . 2L
(20)
According to (20), the line current will be a sinusoidal waveform in-phase with the line voltage, if D is constant. Input average power Pin can be calculated using the line current equation given in (20). The inductance L must be designed such that the required input power can be
provided with desired switching frequency and duty cycle. By neglecting the converter losses, input average power is equal to the constant output power Po. Therefore, the inductance L can be approximately obtained using the equation L
D 2 V m2 4 Po f s
.
(21)
In order to provide the ZVZCS turn-off condition for the switch, it must be turned off within mode 5 when its body diode is conducting. Thus, on-time duration of the switch can be expressed in terms of resonant period Tr1 of Lr and Cr as DT s
1 T r1 1 1 T r1 T r1 . 4 2 4 4 2
(22)
According to (22), Tr1 (or ωr1) is obtained using desired DTs. For uniquely determining the Lr and Cr, it is essential to specify Zr1 in addition to ωr1. Since the resonant characteristic impedance effects on the resonance peak current, Zr1 must be designed such that magnitude of the negative peak current of iLr during mode 5 be greater than the maximum value of ILp which is equal to VmDTs/L in accordance with (19). In each switching cycle, iLr discharges Cr2 within mode 1 and charges it within modes 4 to 6. Current balance of Cr2 implies that integration of iLr over [t3,t6] must be equal to that over [t0,t1] which is V2Cr according to (3). Assuming iLr to be a half-sinusoidal waveform with frequency of 2 r 1 and integration value equal to V2Cr, magnitude of the iLr negative peak in mode 5 is obtained
as I Lr _ np
V2 2 Z r1
.
(23)
On the other hand, the power transferred from CB to the load by output stage is equal to Po. Therefore, average of the iLr on each switching cycle is Po/VB. Noting that magnitudes of integration of iLr over [t0,t1] and [t3,t6] are equal, it can be written that
1 Ts
t3
t1
i Lr (t ) dt
Po VB
.
(24)
By averaging iLr given in (6) and comparing the result with (24), V2 is calculated as Po T s
V2
.
2 Cr V B
(25)
Magnitude of the iLr negative peak can be stated by substituting V2 from (25) into (23) as I Lr _ np
Po T s r 1 2V B
.
(26)
Average of iLlk on each switching period is equal to the output current Io. Since current flowing through the Cr1 is sum of the currents iLr, ILm, and iLlk/n, average current of Cr1 can be expressed as 1 Ts
i
T s Cr 1
(t ) dt
Po I I Lm o VB n
.
(27)
Considering that average current of Cr1 is equal to zero due to its current balance, magnetizing inductance current ILm is obtained as I Lm
I o Po n VB
.
(28)
Since Cr1 is parallel with the magnetizing inductance of the transformer, average of vCr1 must be zero on each Ts due to voltage balance of Lm. Integrating vCr1 during mode 1 given in (4) yields t1
t v Cr1 (t ) dt
V 2 V 1
0
2 2 r 1
V2 2 r 1
.
(29)
Noting that average of the voltage across Llk is zero, integration of vCr1 over [t2,t7] can be calculated as
t7
t2
v Cr 1 (t ) dt n V o T r 2 2
(30)
where Tr2 is the resonant period of n2Llk and Cr1. Due to discharge of the Cr1 by constant current ILm within the modes 8 and 9, integration of vCr1 over [t7,t9] is obtained as
t9
t7
v Cr 1 (t ) dt
T c I Lm T c 2V 1 2 Cr
(31)
where Tc is the time duration of modes 8 and 9 in which Cr1 linearly charges and it can be approximately stated as Tc T s
T r1
Tr 2 . 4 2 2
(32)
As mentioned, summation of vCr1 integrations given in (29) to (31) must be zero. Therefore, by neglecting the time interval of mode 2 since its duration is so short, a second-order equation in terms of VB can be extracted as T T 2I T 2PT 1 T r1 T c V B2 nV o T r 2 c o V B r 1 2 T c c o s 0 . 2 nCr T s 2C r 4 2 2
(33)
For given parameters, (33) can be numerically solved for different values of Cr. Then, magnitude of iLr negative peak is attained using (26) which can be plotted versus Cr. Thereby, an appropriate value for Cr is selected such that I Lr _ np V m DT s L
(34)
would be valid which ensures soft-switching condition for the switch turn-off. After determination of Cr, inductance Lr is obtained using Lr 1 (r 12 C r ) .
(35)
4. Experimental Results In this section, experimental results of a laboratory prototype of the proposed LED driver are presented. First, the proposed driver is designed for supplying an LED string with 21W consisting of nine 3.3V white LEDs from 220V/50Hz utility. Then, control approach of the proposed driver is discussed and the employed control circuit is shown. Also, some results obtained from testing the implemented prototype are provided.
A. Design Procedure In order to design the proposed driver for specified output power and input voltage, switching frequency and operating duty cycle are selected to be 50kHz and 0.25, respectively. Approximate value of inductance L is obtained using (21) as 1.5mH. As mentioned, L must be selected larger. In accordance with (22), resonant period Tr1 is determined equal to 8.3µs which corresponds to ωr1 of 760krad/s. In analysis of the proposed driver, it was assumed that equivalent inductance of Llk transferred to primary-side of the transformer is much larger than Lr. In order to realize this assumption, turns ratio of the transformer must be sufficiently large. In addition, it was also assumed that nVo is greater than VB. The driver is designed such that VB is lower than 200V in order to reduce the switch voltage stress. Noting that Vo is 30V, n is selected to be 7. Furthermore, it is essential that Lm is large enough. Llk is obtained by measuring the leakage inductance seen from secondary-side of the transformer. According to (32), Tc depends on Tr2 and, consequently, on Cr. Substituting the parameters into (33) yields a second-order equation of VB which is numerically solved for different values of Cr in the range of 1nF to 10nF. Using (12), magnitude of the iLr negative peak is obtained which is plotted in Fig. 4. Maximum value of ILp is also illustrated in the plot. As it can be observed, Cr must be selected greater than 6nF according to (34). Furthermore, obtained values of VB is also plotted in Fig. 4 which shows that VB reduces by increasing Cr. In order to keep VB lower than 200V, Cr is selected as 8.2nF, thereby Lr is attained 220µH using (35). Components used in the implemented prototype are listed in Table I. Since the capacitances are small enough, film capacitors can be used instead of electrolytic capacitors. Fig. 5 shows the photo of the implemented prototype in which size and components of the driver are demonstrated.
Fig. 4. Plot of iLr negative peak and VB versus different values of Cr. TABLE I COMPONENTS OF THE IMPLEMENTED PROTOTYPE Component Switch S Bridge diodes Dr Diode D1 Diodes D2 and Do Inductor Lf Inductor L Inductor Lr1 Transformer T (Lm, Llk, n) Capacitor Cf Capacitor CB Capacitors Cr1 and Cr2 Capacitor Co
Fig. 5. Photo of the implemented prototype.
Value 2SK2611 UF4007 UF5408 MUR460 1mH 2mH 300µH 5mH, 15µH, 7 220nF / 400V 2×4.7µF / 250V 8.2nF / 630V 2×4.7µF / 100V
B. Control Circuit Noting that the input cell of the proposed driver is a DCM buck-boost converter, the amount of power drawn from the utility and transferred to output can be controlled by changing the switch on-time. On the other hand, to provide ZVZCS turn-off condition for switch, it must be turned off when its body diode is conducting. Since the body diode conducts during a period of time (mode 5), there is a time interval in which the switch can be turned off. Increasing the magnitude of iLr negative peak extends the duration of mode 5 and, consequently, increases the range of duty cycle variation. However, this also results in higher current stress of switch. If output power is approximately constant and variation of the input voltage amplitude is limited, which is the case in LED driver application, duty cycle would vary slightly. Therefore, a regular pulse-widthmodulation (PWM) controller can be employed to control the proposed LED driver. Control circuit of the implemented prototype is shown in Fig. 6 in which output current is sensed and its feedback signal is isolated by PC817 and TL431 devices. Then, UC3842 generates the gate pulses for the circuit’s switch with appropriate duty cycle. The PWM controller has low bandwidth such that the duty cycle remains almost constant in order to keep the input current harmonics low.
Fig. 6. Control circuit of the implemented prototype.
C. Results Experimental waveforms of the implemented prototype are demonstrated in Fig. 7. Fig. 7(a) shows the input current waveform of the driver which is entirely sinusoidal and in-phase with the line voltage. Current and voltage waveforms of the switch S is illustrated in Fig. 7(b) in which ZCS turn-on and ZVZCS turn-off conditions of switch can be observed. Fig. 7(c) shows the diode D2 current and voltage waveforms. Voltage waveform of D2 is also the voltage across the capacitor Cr2. Output current and voltage waveforms of the driver are illustrated in Fig. 7(d) in which current ripple of about 25% is observed.
(a)
(b)
(c)
(d)
Fig. 7. Experimental waveforms of the implemented prototype. (a) Input current and voltage waveforms. (b) Current and voltage waveforms of the switch S. (c) Current and voltage waveforms of the diode D2. (d) Output current and voltage waveforms.
Input current harmonics of the implemented driver have been measured under three line voltages 185Vrms, 220Vrms, and 265Vrms, and the results are plotted in Fig. 8. Limits of the IEC 61000-3-2 class C standard are also illustrated in this figure. As it can be observed, the proposed LED driver complies with the IEC standard and it has very low input current harmonics. Based on measured current harmonics at 220Vrms line voltage, total harmonic distortion (THD) and PF of the implemented PFC LED driver are 2.6% and 0.999, respectively.
Fig. 8. Measured input current harmonics under three different line voltages compared with the IEC 61000-3-2 class C standard.
Fig. 9 shows the efficiency of the implemented prototype under 20% to 100% of the nominal load for three different line voltages. In measuring the efficiency, only the power stage has been considered and losses of the control circuitry have not been taken into account. It can be observed that the driver has higher efficiencies at lower line voltages. Efficiency of the implemented prototype under full-load condition and 220Vrms line voltage is 92%.
Fig. 9. Efficiency of the implemented prototype under different load conditions for three line voltages.
5. Conclusion A new S6-PFC LED driver without electrolytic capacitors was introduced in this paper which applies a buck-boost converter as PFC cell integrated with an isolated resonant converter as DC-DC cell. In the proposed LED driver, soft-switching conditions are provided for all the semiconductor devices without using any extra active switch. Also, the proposed driver utilizes the transformer leakage inductance as a resonant component so that the leakage energy is recycled. Therefore, efficiency of the proposed LED driver is high and it can be a suitable option for low power applications. On the other hand, due to employment of buck-boost converter as the PFC cell in the proposed LED driver, input current harmonics are very low and, consequently, PF is very high.
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