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Abstract— A new converter topology for drives is presented in this paper: a three-phase multilevel converter with separately regulated dc power supplies. The dc ...



A Three-Phase Multilevel Converter for High-Power Induction Motors Nikolaus P. Schibli, Tung Nguyen, and Alfred C. Rufer

Abstract— A new converter topology for drives is presented in this paper: a three-phase multilevel converter with separately regulated dc power supplies. The dc voltages are provided by medium-frequency dc–dc converters. The applications for the converter are especially high-power traction systems, where the voltage applied to the induction motor is bigger than 1 kV. The motor current is of a very high quality, compared to a classical three-phase converter. This allows keeping the switching frequency low by using phase-shifted pulsewidth modulation (PWM) carriers. Different modulation methods have been developed and simulated. Experimental tests have been made on a 12-kW prototype. Index Terms— DC–DC converter, high-speed motor drives, multilevel converter, PWM modulation, three-phase converter.



IGH-POWER induction motor drives using classical three-phase converters have the disadvantage of poor voltage and current qualities. To improve these values, the switching frequency has to be raised which causes additional switching losses. Another possibility is to put a motor input filter between the converter and motor, which causes additional weight. A further inconvenience is the limited voltage that can be applied to the induction motor determined by the blocking voltage of the semiconductor switches. For highpower semiconductors, the switching frequency is limited by the maximal power loss. In this paper, a three-phase multilevel converter is described. The aim is to generate motor currents of high quality by using more semiconducting devices. In particular, the converter is used for high-voltage motor drives ( 2 kV). The series connection allows reaching much higher voltages than the blocking voltage of the semiconductors. The maximal voltage is now limited by the isolation voltage of the medium-frequency transformer. The resulting switching frequency is a multiple of the switching frequency applied to the switches. This allows working with a filter of reduced size or with no filter at all. This paper describes a multilevel converter with regulated dc–dc converters for the dc-voltage supply. The modulation of this kind of converter has been examined and simulated. The next three figures show an overview of three different kinds of three-phase converters: Fig. 1 shows the classical converter, Fig. 2 shows the neutral point clamped converter (NPC), and, finally, Fig. 3 shows the new converter topology scheme. Manuscript received May 7, 1997; revised January 21, 1998. Recommended by Associate Editor, A. Trzynadlowski. The authors are with the Swiss Federal Institute of Technology, 1015 Lausanne, Switzerland. Publisher Item Identifier S 0885-8993(98)06490-4.

II. THREE-PHASE MULTILEVEL CONVERTER The basic element of the converter is the full-bridge converter. The full bridge is supplied with a floating dc voltage, either a dc voltage from a regulated [1] or unregulated dc–dc converter, or by a series connection of a transformer output [3]. The output on the other side of the full bridge can now be a positive or negative output step. With a series connection of several full-bridge cells, the positive and negative dc voltages can be superposed to create a high-voltage output for each phase on the induction motor. In the example in Fig. 3, two cells in series per phase were used. Fig. 4 shows a typical cell for a multilevel converter realized with insulated gate bipolar transistor (IGBT) switches. Each switch has its own snubber, consisting of a capacitor and a resistor in series, with an additional clamping diode [4]. This protects the switches from overvoltages caused by the stray inductances, but also avoids electromagnetic interference (EMI) influences on the driver circuits. The current through the dc sources is not sinusoidal and changes its value instantaneously (see simulation results). So, it is very important to reduce scatter inductance between dc source and the fullbridge converter to limit overvoltage across the IGBT’s ( nH). Experience has shown that the EMI protection is an important point for the realization of a high-power multilevel converter. This demands special construction precautions for the multilevel cells. It is obvious that the dc-voltage supply has to be floating for each cell. This can be made with a transformer with multiple secondary windings and rectifiers [3]. If a bidirectional energy exchange has to be provided between the common dc circuit and the multilevel cells, a dc–dc converter with a medium-frequency transformer is proposed [1], with a switching frequency of 10 kHz (shown in Fig. 5). The dc–dc converter allows reducing the weight by an important factor compared to the low-frequency (50 Hz) transformers. The additional stray inductance allows generating a controlled power flow in both directions. The dc voltage on the converter side can be stabilized and varied for different voltage output on the motor. If only reactive power is taken from the capacitors, no dc–dc converters are needed, for example, for static var generation or active filtering. A modulation algorithm has to assure the equilibration of the voltages on the capacitors [7]. An additional advantage of the multilevel converter is the reduction of voltage stress of the motor caused by high ’s. While classical converters switch on immediately the

0885–8993/98$10.00  1998 IEEE



Fig. 1. Classical three-phase converter for drives.

Fig. 2. Neutral point clamped converter.

Fig. 5. Two-quadrant regulated dc–dc converter.

Fig. 3. Basic scheme of the three-phase multilevel converter. Fig. 6. PWM signals adaptation.

A. Pulsewidth Modulation (PWM)

Fig. 4. Elementary module of the multilevel converter.

whole voltage to the motor, the multilevel converter allows approaching the ideal sine wave. III. MODULATION METHODS For the multilevel converter, different modulation methods can be used to drive an induction motor. The following methods have been developed and tested.

The first method described is a PWM modulation. For one full-bridge converter, two auxiliary triangular functions have to be generated and compared with a sine-wave signal (one for the half bridge generating the positive voltage and the other for the half bridge generating the negative step). So, for the first phase of the multilevel converter, two steps with two triangles per step are taken using the phase angles 0 , 90 , 180 , and 270 . To create shifted pulse patterns for the other two phases, again, phase-shifted triangular signals have to be used. For the second phase, signals with the angles 30 , 120 , 210 , and 300 , and for the third one, 60 , 150 , 240 , and 330 are generated for the modulation. Furthermore, it has to be assured that there is no half bridge causing a positive voltage while another half bridge is causing



Fig. 7. Typical PWM switching signals.


a negative voltage at the same time. This would not cause an output voltage. Like this, a power transfer from a positive cell into a negative without feeding the motor can be avoided. The schematic for this function for one phase is showed in Fig. 6. The logic A and logic B blocks count the number of bits for the switching of positive and negative steps. The combinatorial logic afterwards assures that there are either only positive or negative steps which will be turned on. These output signals can be seen in Fig. 7. The letter represents the number of half bridges in one phase of the converter. The disadvantage of the PWM modulation is especially the high-switching frequency causing additional power losses. Table I shows different power losses for a four-level singlephased converter using IGBT switches. The overall power is 4.8 kW for a converter with the galvanically separated dc voltages at 200 V. The ac voltage amplitude is 800 V. The . current amplitude is 15 A and the power factor is The following loss energies have been taken for the evaluation: IGBT turn off 10 mJ, turn on 7 mJ, Diode turn off 3 mJ, turn on 1 mJ at 600 V/10 A. The on-state values are: diode forward voltage drop 1.4 V, IGBT saturation voltage 2.2 V. The switching losses are considered to depend linearly on the switched current and voltage. Different switching frequencies have been looked at. B. Step Modulation To avoid the high-switching losses, the step modulation method can be used. The functionality is simple, one phase of the converter works like a quantizer of the ideal sine-wave reference voltage. If the current quality has to be improved, a larger number of steps per phase have to be taken. Three different methods were developed to commutate the single-


phase multilevel converter with the step modulation: the first method consist of a normal series turn on and turn off from one cell after the other. Fig. 8 represents the modulation for a half period. In this modulation method, the conduction losses for the first full bridge are the largest, and the rms current is different for each stage. On the other side, the switching losses are not distributed equally. The power losses of each full-bridge cell are resumed in Table II (the same conditions like Table I). To avoid unequal power distribution, the second method uses step modulation with load sharing [6]. With the help of a sequencer in the digital modulator, a selected turn off is chosen, in accordance with the sequence of the turn-on stages (Fig. 9). For this modulation method, the conduction power losses for each stage are better distributed, but the switching losses are still not distributed equally (Table III). The first full bridge takes the important part of the switching losses. This causes a superior stress in the recovery diode and IGBT of the first element. If the frequency of the generated voltage is raised for high-speed operation of motors, the commutation losses take an important part of the total power




Fig. 11. Output-voltage vector positions.




A. Fast Fourier Transform (FFT) Analysis

Fig. 10.

Rotating commutation of steps.

losses, especially in high-power IGBT and gate-turn-off (GTO) thyristor modules. To distribute the switching losses equally over all cells, the third and final step modulation method was implemented with a rotating duty after every period of the fundamental sine wave. The basic principle for four levels is shown in Fig. 10. C. Vector Modulation The upper two methods can also be used for single-phase multilevel converters. For the three-phase version, vector modulation can be used, allowing timing advantages for vector current control. With high-speed processors like DSP, the coordinate transformation can be made for fast sampling times (200 s). The basis for this modulation method can be taken out of Fig. 11, where all the possible output voltages are represented on the complex vector plane (1) for the multilevel converter (1) The converter output voltage is generated with a reference voltage turning in an ideal circle of the complex plane; the nearest point has to be estimated at each sample period. There are 61 different output vector points (classical converter: seven points). With a lookup table programmed on an ROM, the output pattern for the multilevel converter is generated, and the load sharing is generated by sequential logic.

The presented results have been done by MATLAB. The multilevel converter was compared with the classical threephase converter and NPC converter. The dc-link voltage for the NPC converter and the classical converter was chosen at 300 V, while the independent dc voltages of the multilevel converter were chosen at 75 V. The switching frequency is 500 Hz. FFT analysis of typical PWM voltage waveforms has been performed. Fig. 12 shows the phase voltages and its FFT for all converter types. The first graph shows the phase voltage, and it has five different levels for the classical converter. The second graph shows the FFT analysis, with the first harmonics distributed around the 500-Hz switching frequency. The second simulation shows a typical (graph 3, 4) waveform from an NPC three-phase converter. With positive, negative, and zero voltages for each independent phase, nine different voltage levels are generated for the phase voltage. The FFT shows the low-order harmonics around the double-switching frequency due to the fact that phase-shifted triangular carrier signals were used. The last of these analyses shows the multilevel converter, realized with two full bridges per phase. Fifteen different voltage levels can be reached for the phase voltage. The FFT shows very poor harmonic content, beginning at around four times the switching frequency. Table IV compares some of the features of the three different converter types for the same output power. For numerical analysis of the output waveforms of a multilevel converter, the mathematical expressions for the line voltage have been calculated [5]. The following expression represents the voltage for a converter working with step modulation:




Fig. 13. Pulse pattern angles for commutation.

Fig. 12. Phase-voltage waveforms and FFT for the different three-phase converters.


Fig. 14. Resistive load, step modulation.

With (2), the waveform of the harmonic of the order can represents by analyzed by varying the turn-on angles . the voltage of one dc source and is the number of the phase (one, two, and three). The letter represents the number of half bridges per phase. Equation (3) shows the same evaluation for PWM modulation

(3) The letter

corresponds to the switching frequency factor (4)

where angles

is the fundamental frequency. The commutation of a quarter period are found with Fig. 13.

Fig. 15. Induction motor, step modulation.


Fig. 16.

Current vector, step modulation.


Fig. 18. Current vector, PWM modulation.

Fig. 19. The dc current and output voltage.

Fig. 17.

Induction motor, PWM modulation.

The phase angles were calculated by comparing mathematically a triangular function with a sine wave. With the software MATHEMATICA, the angles could be found to evaluate (3). B. SABER Simulation Results A four-level three-phase multilevel converter has been implemented on the SABER simulation tool. Nonideal switches and sources with internal resistance (0.5 ) have been used. The fundamental frequency is 50 Hz, and the switching frequency was chosen at 500 Hz. Twelve phase-shifted triangular carrier signals for the PWM were used. The independent dc voltages were chosen at 150 V. The first results show the

characteristical waves of the converter on a resistive load (14 ). The graphs presented in Fig. 14 are showing line voltage, line–line voltage, motor star-point voltage, and line current. The second simulation result represented in Fig. 15 shows the multilevel converter driving an induction motor , mH, and internal voltage source ( amplitude: 266 V). In Fig. 16, the current phasor for the simulated motor can be seen. Already with the step modulation, the current quality is comparable to the current of a classical converter in PWM modulation. The following simulation results were now made with PWM modulation. A low PWM switching frequency (500 Hz) has been chosen in order to show the harmonics. Fig. 17 shows the motor driven by the multilevel converter: Again, the current phasor is presented in Fig. 18. The highcurrent quality at a relative low-switching frequency of 500 Hz can easily be seen. An interesting point in the simulations was to look at the current in each dc–dc converter. The current waveform has to be taken in consideration for the design of the control circuit and the construction of the multilevel converter. The first graph of Fig. 19 shows a typical output voltage of a full-bridge multilevel cell with PWM modulation, and the second one shows the current taken from the dc supply. The



Fig. 20.

12-kW multilevel converter for experiments and tests.

Fig. 21.

Step modulation, driving induction motor.

current has a sinusoidal output including the current ripple multiplied by the sign of the output voltage of the full bridge (5) ’s in the supplies. The converter switching causes high A multilevel converter with unregulated sources having non’s negligible internal resistance and stray inductance, the will influence the output voltage of the converter. V. EXPERIMENTAL RESULTS For the verification and test of all the modulation circuits described above, a reduced-power model has been realized

in the research lab ( kW). IGBT’s from FUJI (2MBI50L-120 half-bridge modules) were used as switches, with the highest possible switching frequency per full-bridge cell of 10 kHz. Tests have been made on resistive load and on a 4-kW induction motor with nominal values of 120 Hz/9.5 A/390 V. The motor load is a dc-current motor coupled with the induction motor. A current regulator has been implemented on a standard PC Intel 486DX4 processor equipped with a Burr–Brown AD card and a XILINX 4008 gate-array circuit synchronized with the PC for the digital implementation of the modulators (see Fig. 20). In Figs. 21 and 22, an induction motor is driven with a frequency of 120 Hz. It can be seen that the motor which has


Fig. 22.

Measured current vectors with step modulation.


Fig. 24. Measured current vectors with PWM modulation.

nection of steps, a redundant motor drive can be designed. In case of a power failure of one step, the other multilevel steps of the same phase can compensate the missing voltage step. The proposed solution has many advantages, especially for highvoltage motor applications. With the new modulation method, the losses are distributed equally on each full-bridge cell. The simulations and experimental results show the feasibility of such a system. The dc–dc converters provide a bidirectional energy exchange and can be realized in different ways [1], [4]. REFERENCES

Fig. 23.

PWM modulation, motor current.

been used has lower principle and stray inductance than the simulated motor. The current phasor does not have the same waveform like in the simulations. So, the current vector drops to the center at several points. The dashed line shows the current vector with a classical converter to compare the current quality improvement with the new converter topology. Figs. 23 and 24 represent the motor working with PWM modulation The switching frequency per half bridge is chosen ), which provides a resulting frequency at 1.2 kHz (10 of 4.8 kHz. The measurement in Fig. 24 compares the current quality of a classical and ML converter. The resulting switching frequency is four times higher compared to the classical converter due to the shifted carrier signals. VI. CONCLUSIONS A new converter topology has been presented for drive applications in high-power fields. With a high number of semiconducting devices, current quality is improved and weight reduced by avoiding heavy current filters. With the series con-

[1] A. Rufer, N. Schibli, and C. Briguet, “A direct coupled 4-quadrant multilevel converter for 16 23 Hz traction systems,” in PEVD96 Power Electronics and Variable Speed Drives Conf., pp. 448–453. [2] T. Meynard and H. Foch, “Imbricated cells multi-level voltage-source inverters for high power applications,” EPE J., vol. 3, no. 2, pp. 99–106, 1993. [3] A. Rufer, “An aid in the teaching of multilevel inverters for high power applications,” in Power Electronics Specialists Conf. PESC, Atlanta, GA, 1995, pp. 347–352. [4] N. Mohan, M. Undeland, and W. Robbins, Power Electronics. New York: Wiley, 1995, pp. 686–688. [5] M. Bhagwat and R. Stefanovic, “Generalized structure of a multilevel PWM inverter,” IEEE Trans. Ind. Applicat., vol. IA-19, no. 6, pp. 1057–1069, 1983. [6] W. Schminke, “Modulateurs de tr`es grande puissance en technique PSM pour e´ metteurs a` ondes courtes de 500 kW,” Review Brown Boveri, vol. 5, pp. 225–240, 1985. [7] F. Z. Peng, J. Lai, J. W. McKeever, and J. VanCoevering, “A multilevel voltage-source inverter with separate dc sources for static var generation,” IEEE Trans. Ind. Applicat., vol. 32, no. 5, pp. 1130–1138, 1996.

Nikolaus P. Schibli was born in New York City, NY, on November 14, 1970. He received the Master’s degree in electrical engineering from the Federal Institute of Technology, Zurich, Switzerland, in 1996. He is currently working towards the Ph.D. degree in multilevel converters for high-power systems at the Federal Institute of Technology. He joined the Industrial Electronics Laboratory at the Swiss Federal Institute of Technology of Lausanne (EPFL), Switzerland. At EPFL, his research is focused on reversible dc–dc converters and control systems for power electronics.



Tung Nguyen was born in Saigon, Vietnam, in March 1963. He received the Master’s degree in electrical engineering from the Swiss Federal Institute of Technology of Lausanne (EPFL), Switzerland, in 1997, where he worked on three-phase multilevel converters. Since 1997, he has been working on telecommunications at a private company.

Alfred C. Rufer was born in 1951 in Diessbach, Switzerland. He received the Master’s degree from the Swiss Federal Institute of Technology in Lausanne (EPFL), Switzerland, in 1976. In 1978, he joined ABB, Turgi, Switzerland, where he worked in the fields of power electronics and control such as high-power variable-frequency converters for drives. In 1985, he was a Group Leader for power electronics development at ABB. Since 1996, he has been a Professor and Head of the Industrial Electronics Laboratory at EPFL. He has several patents and is the author of publications on modulation and control methods and power electronics.

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