A Three-Port Half-Bridge Converter with Synchronous Rectification for Renewable Energy Application Hongfei Wu, Yan Xing, Runruo Chen, Junjun Zhang Jiangsu Key Lab. Of New Energy Generation and Power Conversion Nanjing University of Aeronautics and Astronautics Nanjing, China
[email protected]
Kai Sun
Hongjuan Ge
State Key Lab. of Power Systems, Department of Electrical Engineering Tsinghua University Beijing, China
[email protected]
Jiangsu Key Lab. Of New Energy Generation and Power Conversion Nanjing University of Aeronautics and Astronautics Nanjing, China
[email protected]
Abstract—A three-port half-bridge converter for stand-alone renewable power system is proposed. The converter, featuring one input port, one bidirectional port and one isolated output port, is derived by integrating half-bridge, forward-flyback and Buck topologies. The magnetizing inductor of the transformer functions as an inductor, by which a power flow path can be configured between the renewable source and the battery, and the synchronous rectification circuit provides a free-wheeling path for the current of inductortransformer. The power flow through the three ports is controlled with the two switches in primary operating independently. Single stage power conversion between any two of the three ports is achieved. The operational states and principles of the proposed converter are analyzed in detail and verified with experimental results. Furthermore, the topology generation and control method of the proposed three-port halfbridge converter is extended and some other novel three-port and multi-port half-bridge converter topologies are proposed.
I.
INTRODUCTION
Renewable sources, such as solar, tide and wind, are intermittent in nature and fuel cell system features a slow transient response. To smoothly supply loads, storage elements like battery or supercapacitor, functioning as an energy buffer, are usually required in a stand-alone renewable power system. To better interface the renewable source, storage elements and loads, a three-port converter (TPC), as shown in Fig. 1, could be good candidate since it could have fewer conversion stages, less component count, more compact and higher efficiency compared with a solution employing several independent converters [1]-[3]. Due to their remarkable merits like low cost and compact structure, unified power management among the ports, and lower cost, many multi-port converters have been proposed recently for various applications, such as hybrid electric vehicles [4-7], fuel-cell and battery systems [8], aerospace power systems [9-10], PV systems with battery backup or hybrid energy storage systems [11-16] and micro-inverter
Three Port Converter
Figure 1. A stand-alone renewable power system with three-port converter.
with power decoupling [17] etc.. Many methods have been invested to provide a multiport interface, such as the timesharing concept, dc-link coupling via a dc-bus, magneticcoupling through a high-frequency transformer, and trimodal operation etc.. However, most of the TPC topologies are much more complicated than conventional two-port DC/DC converters, because additional switches, inductors, capacitors or transformer windings are added to handle the power flow among the three ports. The half-bridge converter (HBC) is one of the basic topologies with isolation and the primary switches are usually operated alternately or in complement while the input capacitors in a HBC can be regarded as voltage sources and their voltage can be both charged higher and discharged lower. This drops a hint that we may build a three port converter from a HBC if we parallel connecting a storage element, such as battery, with the dividing capacitor of the HBC, because battery can either be charged or discharged by controlling the main switches of the HBC. A novel three-port half-bridge converter (TPHBC) along with its control and modulation are proposed in this paper. The devices’ number of the TPHBC proposed is exactly the same as the conversional half-bridge converter. Therefore, the TPHBC has merits of reduced number of devices, high integration, low cost, small size and weight etc., and is suitable for application of standalone renewable power system.
This paper is supported by National Natural Science Foundation of China (51077071) and State Key Lab of Power Systems, Tsinghua University (SKLD10KM02, SKLD10M09). This paper is also supported by Tsinghua University Initiative Scientific Research Program (20101081909)
978-1-4577-0541-0/11/$26.00 ©2011 IEEE
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Vin
C1
D1
S1 Lm
C2
Vin
C1 C2
S1 S2
S2
I in Vin
Vo
Vb
D2
C1
S1
Ib
C2
S2
iP + vDS1 + − L vP m iLm −
Bidirectional switching cell
S3 S4
Vo
iLo
T ∗
S3
∗
∗
Lo
Io
Co
Vo
S4
1: n : n Three-Port Half-Bridge Converter
Vin Vb
C1
C1
S1 Lm
S2
C2
Half-Bridge
Vb
S1 Lm
C2
S2
S3 S4
Vo
Forward-Flyback
Buck
Figure 2. Generation process of three-port half-bridge converter.
II.
TOPOLOGY AND ANALYSIS ON THE THREE-PORT HALF-BRIDGE CONVERTER
vGS1
A. Generation of Three-Port Half-Bridge Converter Considering that there are three power flows in a standalone PV power system: 1) from PV to load, 2) from PV to battery, and 3) from battery to load as in the system in Fig. 1. To derive a TPC, the three power flow paths should be configured firstly. Referring to Fig. 2, where transformer is modeled as a magnetizing inductor, Lm, paralleled with an ideal transformer, the power flow from PV to load is offered by the half bridge topology originally. It is observed that there is a bidirectional switching cell, employing the magnetizing inductance of the transformer, Lm, as its filter inductor, parasitized in the primary side of HBC, so the battery is connected in parallel with one of the input capacitors, C2, to be charged by the PV by operating the bidirectional switching cell as a synchronous rectification Buck (SR-Buck) and then the power flow from PV to battery is built. After that, we find the power flow from the battery to the load is also built via a forward-flyback converter (FFC) [18]. As illustrated in Fig. 2, the three-port half-bridge converter with synchronous rectification (TPHBC-SR) is an integration of HBC, SR-Buck and FFC in fact. It is noted that only one control variable is needed for the conventional HBC, therefore, the two main switches, S1 and S2, operate with equal or complementary duty cycles. However, for a TPC, two control variables are needed to tightly control the power flow through two of the three ports while the third port balances the power in the converter. In the proposed TPHBC-SR, decoupling control of S1 and S2
is achieved thanks to the synchronous rectifier, because a free-wheeling branch across the transformer is built when both S3 and S4 are on. Therefore, the magnetizing inductor current, iLm, and output inductor current, iLo, can freewheel through S3 and S4. As a result, tight control over two of the three ports in the converter and single stage power conversion between any two of the three ports is achieved while the third port provides power balance in the system. B. Operation Mode Analysis Assumptions are made to simplify the analysis as that, C1, C2 and Co are large enough and voltage of the three ports,
t
vGS 4
t
vGS 2
t
vGS 3
t
vDS 1
t
vP
t
iLo
t
iP iLm
t
t0
Mode I Mode II Mode III t3 t2 t1
Figure 3. Key waveforms in DO state.
Vin, Vb and Vo, are constant during steady state. Ignoring the power loss in the conversion, we have
pin = pb + po
(1)
where pin, pb and po are the power flow through the PV, battery and load port, respectively. The TPHBC-SR has three possible work states: (1) dual-output(DO) state, with pin>po, the battery absorbs the surplus solar power and both the load and battery take power from PV; (2) dual-input(DI) state, with pin0, the battery discharges to feed the load along with the PV; and (3) single-input single-output(SISO) state, with pin=0, the battery supplies the load power alone.
When the converter works in DO and DI states, two of the three ports should be tightly controlled. Therefore, duty cycles of S1 and S2, d1 and d2, are controlled independently. The operation modes of the converter in DI state are the same as DO state except that, in DI state,
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I in Vin
C1
S1
Ib
Vb
C2
S2
iLo
iP + vDS 1 + iLm ∗ − vP Lm −
∗
Lo
S3 ∗
iP = iLm + n ⋅ iLo
Io
Vo
Co
(4)
Mode II [t1-t2]: At t1, when S1 and S3 turn off and S2 and S4 turn on, a negative voltage is applied across the transformer’s primary winding [see Fig. 4(b)].
S4
1: n : n
(a) I in Vin
iLo
iP
C1
Ib Vb
C2
S1 S2
+ vDS1 + iLm ∗ − vP Lm −
∗
Lo
S3 ∗
Io Vo
Co
S4
1: n : n
Vb
C1 Ib
C2
S1 S2
iLo
iP + vDS 1 + iLm ∗ − vP Lm −
∗
Lo
S3 ∗
(5)
diLo n(Vin − Vb ) − Vo = dt Lo
(6)
iP = iLm − n ⋅ iLo
(b) Iin Vin
diLm −Vb = dt Lm
Io
Mode III [t2-t3]: At t2, S2 turns off and S3 turns on. The voltage across the primary winding is clamped at zero. Both S3 and S4 are on to freewheel both iLm and iLo during this mode [see Fig. 10(c)].
Vo
Co
S4
1: n : n
(c) Figure 4. Equivalent circuits of each operational modes in DO state: (a) [t0, t1], (b) [t1, t2] and (c) [t2, t3].
vGS 1
(7)
diLm =0 dt
(8)
diLo −Vo = dt Lo
(9)
t
vGS 2
iP = 0
t
iLo
t
iP iLm
t
t0
Mode I
t1
Mode II
t2
Figure 5. Key waveform in SISO state.
the battery is discharged and the average value of ip and iLm is negative while the battery is charged in DO state and average value of ip and iLm is positive. There are three operation modes in one switching cycle. The key waveforms and equivalent circuit of each mode in DO state are shown in Fig. 3 and Fig. 4, respectively.
In SISO state, the battery feeds the load alone, the TPHBC-SR topology is degraded to a FFC, as mentioned before, where S1 and S2 are driven in complementary and S3 and S4 are the synchronous rectifiers. There are only two operation modes (Mode I and Mode II) in one switching cycle and no freewheeling mode (Mode III) anymore. The key waveforms are shown in Fig.5. And the equations in the mode I and II are exactly the same as that in DO state, respectively. Applying the volt-second balance principle on the magnetizing inductor of the transformer, Lm, and the output filter inductor, Lo, respectively, we obtain that: Vin =
Mode I [t0-t1]: Before t0, S3 and S4 are on and S1 and S2 are off while iLo and iLm freewheel through S3 and S4. At t0, S1 turns on and S4 turns off. A positive voltage is applied across the transformer’s primary winding [see Fig. 4 (a)].
diLm Vin − Vb = dt Lm
(2)
diLo n(Vin − Vb ) − Vo = dt Lo
(3)
(10)
( D1 + D2 ) ⋅ Vb D1
Vo = n[ D1 (Vin − Vb ) + D2Vb ] = 2nD2Vb
(11) (12)
where D1 and D2 are the duty cycles of S1 and S2 in steady state, respectively. It can be seen from (11) and (12) that the voltages of the PV source, Vin, can be regulated with D1 for maximum power point tracking (MPPT), taking the battery voltage, Vb, as a constant one. And the output voltage Vo can be further tightly regulated with D2.
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C. ZVS Analysis And Design Consideration The S3 and S4 are operated with ZVS naturally thanks to their body diodes in the synchronous rectification MOSFETs. At the end of the operational Mode I, where S1 is turned off, the energy stored in the transformer leakage inductor is released to charge or discharge the parasitic drain to source capacitors, CDS, of S1 and S2, respectively. As a result, S2 will be operated in ZVS only if the following condition is satisfied. ⎧1 2 2 ⎪ Lk ( I Lm + nI Lo ) > CDS Vb ⎨2 ⎪⎩ I Lm + nI Lo > 0
vin
vb
iin
ib
MPPT +
−
ibr IVR
+ −
vo
TPHBC − SR vGS 1 vGS 2 vGS 3 vGS 4
+ vbr −
BCR
PWM Modulator vC 2 vC1 Max
BVR
−
OVR
+
vC1 vsaw
(13)
vC 2
vGS 4 vGS 1
− +
vGS 3
− +
vGS 2
(b)
As for the semiconductor devices stresses, the TPHBCSR is similar to the traditional HBC. But a key difference between these two converters is, the magnetizing inductor of the transformer, Lm, is functioned as an inductor as well.
vsaw
vC1 vC 2
t
vGS1
t
vGS 3
From (1), in the steady state, we have
t
vGS 2
Vin I in = Vb I b + Vo I o
t
vGS 4
(14)
t
(c)
According to operation mode I, we have I in = D1 ( I Lm + nI o )
Figure 6. Control and modulation for the TPHBC-SR proposed (a) Control diagram, (b) PWM generation (c) Key waveforms of PWM modulator.
(15)
where ILm is the average magnetizing current of the transformer, and then we have I Lm =
I in − nI o D1
(16)
(17)
Then the average transformer magnetizing current, ILm, can also be given by the following equation. I Lm = III.
I b − ( D1 − D2 )nI o D1 + D2
R0 DC Source
Figure 7. Simulation circuit of PV panel in experiment.
On the other hand, the battery current, Ib, equals to the average value of primary winding current, Ip, that is I b = D1 ( I Lm + nI o ) + D2 ( I Lm − nI o )
vor
(a)
(18)
PULSE WIDTH MODULATION AND FEEDBACK CONTROL STRUCTURE
Fig. 6(a) shows the control diagram for the hybrid PV battery system while Fig. 6(b) and (c) show the proposed modulation approach to realize the constant frequency pulse width modulation (PWM), where vsaw is the sawtooth carrier waveform for modulation, vC1 and vC2 are control voltages given by the feedback controllers. In the DO or DI state, as analyzed above, according to (11) and (12) and in steady state, the PV voltage, Vin, can be controlled by the duty cycle
of switch S1, D1. As a result, the power flow through both the PV port and battery port can be controlled with D1. And D1 is modulated by vc1, which presents the output of the only one regulator from the following three ones: PV voltage regulator (IVR) for MPPT, battery voltage regulator (BVR) for maximum voltage charging and battery current regulator (BCR) for maximum current charging, respectively. The output is tightly controlled by duty cycle of switch S2, D2, regulated by the output voltage regulator (OVR). If only pin>0, vC1 is always higher than vC2 to yield D1+D2
