Closed-loop Regulated Power Supply for Ozone ... - IEEE Xplore

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Mar 11, 2008 - Abstract. In this paper the design, implementation and experimental results of a high-voltage power supply for ozone generation are presented.
Closed-loop Regulated Power Supply for Ozone Generation based on Buck Converter and Current-Fed Push-pull Resonant Inverter J. M. Alonso, C. Ordiz, D. Gacio, J. Ribas and A. J. Calleja Universidad de Oviedo, Electrical Eng. Dept. Campus de Viesques, Edificio 3, 33204 Gijón, Spain Phone: +34 985182380 Fax.:+34 985182138 E-mail: [email protected]

Acknowledgements This work has been supported by the regional and national R&D Plans, research grants numbers PC06001 and DPI2007-61267, and the Spanish company RILIZE.

Keywords High voltage power converters, current-fed push-pull resonant inverter, buck converter, high frequency ozone generation, closed-loop operation.

Abstract In this paper the design, implementation and experimental results of a high-voltage power supply for ozone generation are presented. The power stage is formed by a buck converter plus a current-fed parallelresonant push-pull inverter. The buck converter is used to both regulate the output power against the line voltage variation, and to control the power delivered to the ozone generator (OG). The push-pull inverter assures safe operation of the OG by working at the parallel resonant frequency given by the high-voltage transformer plus OG characteristic. In this paper, the closed-loop operation of the proposed converter is analyzed and implemented as a method to compensate variations in the line voltage and in the OG. A low cost commercial UC3872 integrated circuit is used to control both the push-pull inverter and the buck converter, providing OG power control and regulation via buck converter duty cycle. Protections against short circuit and no load operation are also implemented. Experimental results for a 100W prototype are shown and discussed.

Introduction Unlike chlorine, ozone produces much less hazardous by-products and it decays into oxygen within hours. Thus, the use of ozone as a disinfecting and oxidising agent has spread widely in substitution of chlorine, due to the latter’s issues related to smell, unpleasant taste and carcinogenic agents resulting from its use. Ozone finds its application mainly in disinfection, bleaching, water and air purification, chemistry and some pharmaceutical applications [1][2]. However, due to its high instability, ozone can not be stored, so it must be produced on site. Dielectric barrier discharge (DBD) method has proven to be the best method to gnerate ozone. In this approach, dried air or oxygen is forced to pass through a 1-2mm gap formed by two high-voltage electrodes, one of which is covered by a dielectric material so that sparks are prevented from taking place [3]. These devices are preferably supplied by a high-voltage, high-frequency source, typically several tens of kHz. High frequency operation is preferable because high frequencies decrease the necessary voltage to be applied and increase the efficiency in ozone production [2]-[4].

It has been demonstrated that resonant converter topologies are well-suited to supply this kind of OG’s, since they behave as highly capacitive loads [3]-[7]. Note that these OG’s are actually capacitors: they consist of two electrodes with a dielectric material between them. In [3], an open-loop current-fed parallelresonant push-pull inverter was analyzed. Now, the next step is to add the possibility of controlling the power delivered to the OG and allow compensation by closing the loop between the output and the control stage, based on the same UC3872 integrated circuit used in [3][8]. The organization of this paper is as follows. In Section II the proposed power stage is presented. In Section III, the operation of both buck converter and push-pull resonant inverter is illustrated. Section IV deals with the issues related to the implementation of the laboratory prototype, focusing on special matters not treated in previous works, as the design of the filter inductor, high voltage transformer, closed loop regulation, and protections. In Section V, experimental results are provided to validate the proposed design procedure, and evaluate the performance of the proposed topology. Finally, Section VI is devoted to summarize the work with some conclusions.

Proposed Power Stage Fig. 1 illustrates the block diagram of the power stage. It consists of a buck converter and a current-fed push pull inverter. As stated before, a current-fed parallel-resonant push-pull inverter is suitable to supply the major capacitive behaviour of the OG. Because the inverter will be operated at resonant frequency, the best way to control the output power is by changing the inverter dc input voltage [3]. Since the power supply is intended to be supplied from a 220V ac line, a step down converter would be preferable to decrease voltage stress in the power switches. As a consequence, a buck converter has been selected as upstream converter. Besides, the buck converter incorporates an inductor in the output filter, which can also be used to supply the dc current needed by the current-fed push-pull inverter. In this way, the inductor is used in both stages so that the addition of the buck converter is made at a minimum cost.

Fig. 1: Proposed power stage. In order to achieve output regulation, the circuit is operated in closed loop. Since the output voltage, VO, is very high and not referred to ground, the voltage in the step-up transformer central tap, VTAP, is sensed to close the feedback loop. Note that the average value of VTAP is closely proportional to rms value of Vo. The feedback loop uses resistances Rzd1 and Rzd2 to adapt the voltage level. This voltage signal is also used to provide the inverter operation at resonance. The duty cycle of the buck converter is used to control the mean dc current injected to the push pull inverter through the series inductor Li, and consequently the output voltage and power delivered to the OG.

Analysis of the Power Converter As it is well known, the buck converter generates a square voltage waveform, Vbuck, whose mean value is proportional to the duty cycle, D, and to the dc input voltage, Ve, as follows:

Vbuck = Ve ⋅ D

(1)

Fig. 2 shows the basic operation and waveforms of the buck converter. The control signal of the buck stage is synchronised with one half of the output voltage period [9]. That is why in Fig. 2 one buck period is considered to take half a period, T/2; T being the period of the sinusoidal output voltage. The current-fed inverter shares the filter inductor with the buck converter. The input current to the inverter, Ie, is illustrated in Fig. 2b. It can be approximated by a linear interval during the energizing mode and sinusoidal interval during the off state of the buck converter.

(a)

(b)

Fig. 2: (a) Basic operation, and (b) waveforms of the buck stage. On the other hand, the equivalent circuit and operating waveforms of the push-pull inverter are shown in Fig. 3. The upstream buck stage is substituted by a voltage source. In the push pull inverter each switch is turned on during half the period of the output voltage, T. Thus, neglecting the input current ripple both halves of the transformer primary are fed by a square-wave current. Provided that the circuit is operating near the resonant frequency, each half of the primary side will contribute with half a sinusoidal wave, thereby giving a sinusoidal output voltage waveform, Vo. Under these operation conditions, the rms value of the output voltage Vo, can be obtained using the following expression [3][5]:

Vo _ rms = rt

π ⋅ Vbuck 2 2

= rt

π ⋅ Ve ⋅ D

(2)

2 2

rt being the transformer turn ratio.

(a)

(b)

Fig. 3: Basic operation and waveforms of the current-fed parallel-resonant push-pull inverter with centretapped transformer.

Implementation Buck Filter Inductor The inductor of the buck converter is used to supply the dc current to the current-fed push-pull inverter. The current ripple in this inductor is a critical parameter. It must be designed to be low enough so that the push-pull inverter operation is not disturbed. Since the mean voltage across the inductance must be zero, the mean voltage applied to the transformer tap must be equal to the input voltage times the duty cycle, Ve·D. Thus, the voltage in the inductor is given by the following function: ⎧ ⎛ πD ⎞ ⎪⎪Ve ⎜1 − 2 sin 2π f t ⎟ ⎝ ⎠ VLi (t ) = ⎨ ⎪− V π D sin 2π f t ⎪⎩ e 2

T ; 2 T T D ≤t ≤ ; 2 2 0≤t ≤D

(3)

f being the switching frequency, and T=1/f. Provided that the duty cycle is designed lower than 0.5, VLi will always be positive in the interval [0, DT/2] and negative in the rest of the switching period of the buck stage, as shown in Fig. 4.

Fig. 4: Inductor current and voltage waveforms for D= 0.25. Thus, the peak-to-peak ripple current in the inductor can be calculated as follows: T

⌠ 2 ⎛ π DVe ⎞ sin 2π f t ⎟⎟dt. ΔI L = ⎮ ⎜⎜ ⌡D T ⎝ Li ⎠

(4)

2

Solving equation (6), the following value is obtained: Ve D (5) (1 + cos πD ). 4 Li f It can be noted that this expression depends on the duty cycle and has a maximum value at D=0.416. Therefore, the value of the filter inductance can then be obtained for the worst case using the following expression: Ve (6) Li = . 7.63 ⋅ ΔI L f In the presented prototype, the maximum mean current through the inductor is 1A. The maximum current ripple allowed was designed to be 10% (0.1A), and the switching frequency of the push-pull inverter is 25 kHz (50 kHz for the buck converter). The obtained value for this design is then 17 mH. ΔI L =

Step-up Transformer The step-up transformer of the inverter is designed according to the following guidelines: a) In order to avoid acoustic noise, the magnetizing inductance must be designed so that parallel resonance with the OG capacitance is above 20 kHz. b) The turn ratio must provide a maximum output voltage of 1700 Vrms at maximum output power. The necessary magnetizing inductance to attain a given resonance frequency, fres, is given by 1 (7) ; Lμ = (2 ⋅ π ⋅ f res )2 ⋅ COG COG being the OG’s capacitance. This capacitance has been measured to be 338 pF. As the power in the OG rises, the oscillation frequency will decrease 20% approximately [3][10]. Thus, in order to avoid acoustic frequencies the magnetizing inductance is calculated for a maximum operating frequency of 25 kHz, thus giving a magnetizing inductance of 120 mH.

Regarding the transformer turn ratio, for an output voltage of 1700 Vrms the maximum expected peak voltage in VTAP is 150 V at a duty cycle of 0.3. Therefore, the required turn ratio is equal to16.

Regulation Circuitry Fig. 5 illustrates the basic schematic of the control stage. The output power must be controlled by an analog signal coming from an external Programmable Logic Controller (PLC), which can be either a 010V dc voltage or a 0-10mA dc current. From this analog signal the reference signal, Vcontrol, is then generated.

Fig. 5: Control stage for the power supply. Since, the non-inverting input of the UC3872 built-in error amplifier is fixed to a 1.5 voltage reference, and it is not accessible, an additional adapting stage based on a LM358 operational amplifier is used to implement the control signal VControl. With Vcontrol ranging from 0 to 3.5 V, the output power will vary from 0 to 100 W. In order to achieve this goal, the voltage Vcomp, which generates the duty cycle by comparison with a sawtooth wave, must vary from 0.2 to 0.7 V, so that the duty cycle will change from 0 to 0.5. Based on these requirements the circuit is adjusted as illustrated in Fig. 5. The sensed output voltage VTAP is fed back through the voltage divider Rzd1 and Rzd2. Another voltage divider and low pass filter formed by R5, R6 and C1 is used to generate the dc voltage VSense, proportional to the OG voltage, Vo. The ratio between VSense and Vo can easily be calculated as follows:

VSense 1 2 Rzd 2 R5 . (8) = Vo rt π Rzd 1 + Rzd 2 R5 + R6 The error amplifier is configured as an integrator, adding a pole at zero frequency, thus allowing the system to react against perturbations and to follow the reference. The Vref voltage is used to adjust the voltage range of VControl. As a result, the small signal component of the voltage VComp is given as follows: k=

⎞ 1 ⎛ R2 1 (9) ⎜⎜ VControl ( s ) − VSense ( s ) ⎟⎟ C s ⎝ R1 R3 R4 ⎠ Fig. 6 shows the block diagram of the whole power supply in the Laplace domain. The response of the duty cycle comparator has been included as 1/Vsw, Vsw being the peak-to-peak amplitude of the sawtooth wave used in the comparator, which in this case is 1.2V. VComp ( s ) =

Fig. 6: Block diagram of the power supply in Laplace domain. In this type of system the required response is quite slow. Therefore the dynamic behaviour of the power stages and OG can be neglected, and the equivalent transfer function, G(s), can be reduced to its dc behaviour. This behaviour can be obtained from (2) as follows: G (s) =

Vo ( s ) 1 = ⋅ π rt Ve D( s) 2

(10)

rt being the turn-ratio of the high voltage transformer. Fig. 7 illustrates the Bode diagram and the step response for the complete power supply operating in closed loop. As can be observed, a time response of 10 ms is expected, which more than enough for this type of systems.

(a)

(b)

Fig. 7: (a) Bode diagram and (b) step response of the complete closed-loop power supply.

Implementation Details Fig. 8 shows the schematic diagram of the inverter. A capacitor was placed across the transformer’s tap to damp the oscillations due to the leakage inductance.

Fig. 8: Schematic diagram of the inverter. Ideally the inverter would operate at resonant frequency, so that the MOSFET’s would have a zero voltage switching and they would withstand no reverse voltage. In practice, voltage spikes due to parasitic elements will cause switching losses to increase [3][5]. In addition, an open-circuit operation might occur, leading to reverse voltages on the MOSFET’s that will cause their body diodes to conduct. For these reasons, fast-recovery diodes are connected in series with the MOSFET’s. DBD OG’s are likely to end up failing because of their wear-out. The fail usually consists of sparks between the two electrodes due to isolation breakdown. They result in overheating, decreasing in ozone production and eventually short circuit. Therefore, the power supply must be protected against over current. The proposed approach consists of sensing the current across one of the branches of the inverter for over current protection, and the voltage across the transformer tap for over voltage protection. Two comparators that trigger the protection circuit are used. When these magnitudes grow higher than a reference value, the UC3872 integrated circuit is disabled by means of its enable/disable input [9]. Fig. 9 shows the schematic of the protection circuit. When any of the failures occurs, the RS flip-flop is cleared, shooting down the UC3872 integrated circuit. At regular time intervals, pulses are provided by means of an oscillator connected to the flip-flop set input so that the circuit can be restarted.

Experimental Results A prototype was built upon the previously explained procedure and calculated values. During the tests, the OG was fed with a 5 lpm oxygen flow. The test consisted on varying the control signal VControl so that the power of the OG was regulated. Fig. 10a shows the voltages on the transformer tap (VTAP) and the output voltage of the buck converter for an input power of 50W. The switching frequency is 23 kHz and the duty cycle 0.25. Note that VTAP has some superposed ripple due to the effect of parasitic elements in the transformer. Fig. 10b shows voltage and current in one of the legs of the inverter in the same above-mentioned conditions. It can be seen how voltage and current are in phase, thus achieving operation at resonance. The voltage waveform shows spikes at commutation instants due to the effect of transformer leakage inductance.

Fig. 9: Schematic diagram of the protection circuit.

(a)

(b)

Fig. 10: (a) Top: buck converter output voltage (100V/div), bottom: voltage on transformer tap (50V/div); Horizontal scale: 5µs/div, (b) Top: voltage across one of the inverter branches (100V/div). Bottom: Current across one of the inverter branches (500mA/div); Horizontal scale: 10µs/div. Fig. 11 shows the OG voltage and current waveforms. As expected, the output voltage is a nearly a sinusoidal waveform. Current spikes due to ozone-generating microdischarges can also be identified.

Fig. 11: Top: OG current (100mA/div); bottom: OG voltage (1kV/div); Regarding more quantitative results, Fig. 12 shows the input power of the system and the OG voltage as a function of the buck converter duty cycle. It can be seen how the duty cycle allows regulating the OG voltage quite linearly. The maximum input power was 100W for a peak OG voltage of 2300V.

1600 1500 1400 1300 1200 1100 1000 10.0

12.0

14.0

16.0

18.0

20.0

22.0

24.0

26.0

28.0

30.0

Duty Cycle (%)

(a)

(b)

Fig. 12: (a) Input power, and (b) OG rms voltage As commented previously, as the power delivered to the OG increases its capacitance increases due to a higher electric carrier density in the discharge gap. Thus, Fig. 13 illustrates the variation of the switching frequency as a function of the duty cycle. The minimum switching frequency measured was 22 kHz approximately.

Fig. 13: Switching frequency. Fig. 14(a) shows the ozone production. A maximum of 12 g/h was obtained with the tested OG. In this type of systems, the important parameter to measure the system efficiency is the ratio of ozone production to input power. This parameter depends on both the quality of the power supply and the quality of the OG. Thus, Fig. 14(b) presents the efficiency of the system as a function of the duty cycle. A maximum of 160 gO3/kWh was achieved, which is a very good figure for this type of systems.

(a)

(b)

Fig. 14: (a) Ozone production, and (b) efficiency The closed-loop behaviour of the prototype against input voltage perturbations was also tested. Line voltage was stepped from 180Vrms up to 220Vrms. Fig. 15 shows the dynamic response of the power

supply for this test. The response is illustrated showing the sensed output voltage, Vsense in Fig. 5, which is proportional to the output voltage. As can be seen, the voltage recovers its nominal operation level after a transient of 10 ms. This behaviour practically identical to the response predicted by the theoretical design. The response is adequate for this type of systems and a faster response is not required due to the inherent slow dynamic of the OG. Stopped CH1=500mV DC 1:1

CH2=200mV DC 1:1

2008/03/11 19:35:52 20ms/div (20ms/div) NORM:50kS/s

Fig. 15: Dynamic response of the power supply for a line voltage step from 180 up to 220Vrms. Top: Sensed output voltage, Vsens (200 mV/DIV). Bottom: Line voltage (250V/DIV). Horizontal scale: 20 ms/div.

Conclusions In this paper, a closed-loop current-fed parallel-resonant push-pull inverter has been presented. An upstream buck converter has been proposed as a solution to control and regulate the power delivered to the OG. The inclusion of the buck converter can be made at low cost, since only an extra power switch is required in addition to the inverter stage. Moreover, the control of the whole system can be performed by using commercially available integrated circuits as the UC3872. A design procedure for the complete buck converter for this special application has been proposed, including the optimization of the filter inductor required at the input of the current-fed inverter. The dynamic analysis of the complete converter has also been studied, and a low frequency dynamic model has been proposed. Several experimental results on both electrical issues and ozone generation issues have been presented. As final conclusion, it can be said that the proposed topology appears as one of the best suited solutions for this type of systems.

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[2]

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