Compact Ultra-Wideband (UWB) - IEEE Xplore

IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 59, NO. 6, JUNE 2011

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Compact Ultra-Wideband (UWB) Bandpass Filter With Ultra-Narrow Dual- and Quad-Notched Bands Xun Luo, Student Member, IEEE, Jian-Guo Ma, Senior Member, IEEE, Kiat Seng Yeo, Senior Member, IEEE, and Er-Ping Li, Fellow, IEEE

Abstract—In this paper, compact ultra-wideband (UWB) bandpass filters with ultra-narrow dual- and quad-notched bands are proposed using the broadside-coupled microstrip/coplanar waveguide (CPW) structure. The multiple-modes UWB operation is obtained through the CPW detached-mode resonator (DMR) and broadside-coupled microstrip/CPW transition. To avoid the existing interferences such as the wireless local-area network signals (i.e., 5.2- and 5.8-GHz bands) in the UWB passband simultaneously, dual-notched bands can be finely employed and independently adjusted by the embedded quarter-wavelength ( 4) CPW resonators and the 4 meander slot-line inserted in the DMR, respectively. To further cancel the interferences from the 3.5-GHz worldwide interoperability for microwave access and 6.8-GHz RF identification communication, the 4 meander defected microstrip structure is employed. Based on the structures mentioned above, a series of UWB bandpass filters with dual- and quad-notched bands are then designed and fabricated. With good passband/stopband performances, compact size, and low cost, the proposed filters are attractive for the practical applications. Index Terms—Bandpass filter, broadside-coupled microstrip/ coplanar waveguide (CPW) (BCMC) transition, detached-mode resonator (DMR), dual notched, meander defected microstrip structure (MDMS), quad notched, ultra-wideband (UWB).

I. INTRODUCTION

U

LTRA-WIDEBAND (UWB) radio technology has been getting increasingly popular for high-speed high-data wireless connectivity since the Federal Communications Commission (FCC)’s decision to permit the unlicensed operation band from 3.1 to 10.6 GHz in February 2002 [1]. In the UWB systems, UWB bandpass filters, as one of the essential components, are designed via different methods and structures in recent years. Methods such as the multiple-mode resonator (MMR) [2], [3], complementary split-ring resonator (CSRR) [4], [5], and multilayer aperture-coupled patches [6] can employ a good operating passband with a narrow stopband. To

Manuscript received December 31, 2010; accepted February 11, 2011. Date of publication March 24, 2011; date of current version June 15, 2011. X. Luo is with the School of Electronic Engineering, University of Electronic Science and Technology of China (UESTC), Chengdu 610054, China (e-mail: [email protected]). J.-G. Ma is with the School of Electronic Information Engineering, Tianjin University, Tianjin 300072, China. K. S. Yeo is with the Center for Integrated Circuits and Systems, Nanyang Technological University, Singapore 639798. E.-P. Li was with the Institute of High Performance Computing (IHPC), A*Star, Singapore 138632. He is now with the Department of Information Science and Electronic Engineering, Zhejiang University, Hangzhou 210027, China. Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TMTT.2011.2116800

extend the stopband bandwidth, methods such as the cascaded low-pass/high-pass filters [7]–[9] and electromagnetic (EM) loaded bandgap [10], [11] are introduced to obtain wide stopband and good selectivity. Recently, methods using the multilayer coupled structure, such as the back-to-back microstrip/coplanar waveguide (CPW) [12]–[17], hybrid microstrip/slot structure [18], [19], multibroadside-coupled scheme [20], capacitively coupled quarter-wavelength open T resonator [21], and inductance-loaded Y-shaped resonators [22] are reported to develop UWB bandpass filters with good frequency responses and compact sizes. However, radio signals such as the WiMAX (i.e., 3.5-GHz band), wireless local-area network (WLAN) (i.e., 5.2- and 5.8-GHz bands), and 6.8-GHz RF identification (RFID) communication may interfere with the UWB radio system within the range defined by the FCC. Therefore, UWB bandpass filters with single- and multiple-notched bands are needed to suppress the interferences for the indoor and handheld UWB users. A number of UWB bandpass filters with a single-notched band are investigated and discussed in [23]–[32]. In these filters, meander slot-line [31] the method using an embedded can provide the narrowest 10-dB notched FBW (i.e., 2.06%) in the UWB passband to suppress the WLAN interferences. CPW Meanwhile, another method using the embedded resonator [32] can easily employ a narrow 10 dB notched fractional bandwidth (FBW) less than 2.28% with a competitive attenuation slope (i.e., 67.92 dB/GHz in the lower and 94.34 dB/GHz in the upper transition bands). Recently, to avoid the interferences at different operated frequencies, the UWB bandpass filters [33], [34] with good performances using the multilayer nonuniform periodical structure are introduced to obtain dual- and multiple-notched bands. However, the design of multiple-notch UWB bandpass filters in pursuits of better performance, smaller size, and lower cost based on the two-layer structure remains a great challenge. In this paper, compact UWB bandpass filters with ultra-narrow dual- and quad-notched bands are proposed using the broadside-coupled microstrip/CPW (BCMC) structure. The implemented filters are composed of the CPW detached-mode resonator (DMR) and BCMC transition to achieve the UWB frequency responses. The dual-notched operation to cancel the WLAN signals (i.e., 5.2 and 5.8 GHz) simultaneously in meander the standard UWB passband is achieved by the CPW resonator embedded in the DMR on slot-line and the CPW layer. Meanwhile, the meander defected microstrip structure (MDMS) on the microstrip layer can introduce another two notched bands to avoid the interferences such as the

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Fig. 2. Equivalent circuit of the CPW DMR.

Fig. 1. Configuration of the UWB bandpass filter with dual-notched bands. (a) 3-D view. (b) 2-D view. (c) Transmission-line model.

) of the three-line coupled structure, whereas the inner line in the coupled scheme has the characteristic impedance of and electrical length of , respectively. Note that the physof the ical length (i.e., ) of the identical lines equals to inner line. Meanwhile, to form the BCMC transition, two microstrip open-stubs on the microstrip layer with characteristic and electric length (i.e., and ) are fixed impedance slot-lines with charupon the CPW DMR. In addition, the and two identical CPW resonators acteristic impedance are embedded in the CPW DMR, as shown in Fig. 1. of B. Principle and Mechanism

3.5-GHz band worldwide interoperability for microwave access (WiMAX) and 6.8-GHz RFID communication. Based on the principle and mechanism above, a series of UWB bandpass filters with dual- and quad-notched bands are designed, fabricated, and tested. The proposed filters have the merits of good passband/stopband performances and compact sizes. II. UWB BANDPASS FILTER WITH DUAL-NOTCHED BANDS A. Configuration Fig. 1 shows the configuration of the UWB bandpass filter with dual-notched bands, which is constructed on an RT5880 and a thickness of 0.508 mm. dielectric substrate with As depicted in Fig. 1, the microstrip structure with open stubs on the top of the common substrate is used as the input/output (I/O) feed-lines through the broadside-coupled transition to the CPW structure on the bottom. The top microstrip and the bottom CPW share the same signal ground with a miniaturized size. Fig. 1(c) illustrates the transmission line model of the proposed filter, which is composed of four parts: the CPW DMR, BCMC transitions, a pair of meander slot-lines, and two identically emCPW resonators. The CPW DMR is composed of a bedded three-line coupled structure with a loaded short-stub. The loaded short-stub with the characteristic impedance of and electrical length of is directly connected to the two identical lines (i.e.,

1) DMR: The CPW DMR shown in Fig. 2 has two functions, which are: 1) allocate three split frequencies at the lower end , middle , and higher end to meet the UWB operato determine tion limits and 2) introduce another resonance the upper stopband bandwidth. Based on the calculated investigation and EM simulation, it is found that the coupling between ) of the three-line coupled the first and third lines (i.e., scheme has tiny loading effects on the resonances. Thus, the coupling between the two identical lines of the three-line coupled scheme is assumed to be neglected [35]. Therefore, the model of the CPW DMR in Fig. 2(a) can be converted to the equivalent configuration shown in Fig. 2(b). Since the proposed resonator is a symmetrical structure, the dual-mode [36] analysis depicted in Fig. 2(c) is adopted to characterize the CPW DMR (where the – plane behaves as a magnetic/electric ) wall). Note that the two-line coupled structure (i.e., is not the conventional symmetrical coupled transmission lines (i.e., ). Thus, the even and odd mode is not suitable for the definition of asymmetrical coupled lines, and then the dual modes are defined as the normal mode-1 and mode-2. Here, to simply investigate the characteristic impedances (i.e., and ) of the dual modes, the homogeneous condition is assumed (i.e., the Appendix). For the normal mode-1 excitation, there is a voltage null along the symmetrical plane – of the CPW DMR, which can lead to the equivalent circuit illustrated

LUO et al.: COMPACT UWB BANDPASS FILTER WITH ULTRA-NARROW DUAL- AND QUAD-NOTCHED BANDS

in Fig. 2(c). The characteristic impedance can be expressed as

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for this mode

(1) are the characteristic impedances of the asymwhere and metrical coupled lines for the two modes, which can be derived from the Appendix. Based on the resonance condition of , the resonant frequencies of mode-1 can be obtained from (2) as follows: (2) where (3) This equation states that the resonances of mode-1 occur when the electrical length of the asymmetrical coupled lines is the , where odd times of quarter-wavelength (i.e., ). It is found that the center and first-spurious resonant and ) are dominated and calculated by frequencies (i.e., . mode-1 when Meanwhile, for the normal mode-2 excitation, there is no current flow through the symmetrical plane – . Thus, the CPW DMR is introduced to obtain the equivalent circuit shown in for this mode can Fig. 2(c). The characteristic impedance be obtained as

Fig. 3. Analysis of the mode-2 resonant frequencies under the case (i.e., L = 6:7 mm, L = 2:6 mm, S = 0:2 mm, W = 1:3 mm, and W = 1:2 mm).

Fig. 4. Simulated current density about the two resonances (i.e., f and f ) of mode-2.

(4) TABLE I CALCULATED AND EM-SIMULATED RESULTS OF THE CPW DMR

where (5) Let (6) The curve of (6) can be plotted as shown in Fig. 3, where the substrate has a dielectric constant of 2.2 and a thickness of 0.508 mm (Note that, all designs from Figs. 3–11 are based on this type of substrate). It can be noted that (6) is equal to infinite at 3.71 and 8.65 GHz, that is to say, there are two fundamental mode-2 resonant frequencies (i.e., and ) for the proposed CPW DMR. Fig. 4 depicts the simulated current density about the two resonances of mode-2. It is notable that the current is concentrated on the three lines coupled structure and the short-stub, which can prove that our simplification is reasonably made to (6). Meanwhile, in the case of Fig. 3, the mode-1 resonant frequencies (i.e., and ) can be calculated from (2) and is at 6.3 and 18.9 GHz. It is notable that , which is compatible with the above analysis. Table I depicts the calculated and EM-simulated resonances of the CPW DMR, which shows good agreement. The difference between the results is employed by two effects. The first effect is that the cou-

pling between the two identical lines (i.e., ) of the circuit model is neglected, which could afford a tiny loading effect. This effect can shift the resonances slightly. The second effect is that the homogeneous condition is assumed. However, based on the investigation above, it is found that the resonances (i.e., , , and ) derived from the circuit model can finely meet the design procedure limits, comparing to the EM-simulated results. Fig. 5 shows the normalized resonant frequencies (i.e., and ) versus the length (i.e., ) and , whereas Fig. 6 depicts the normalized resonant frequencies and ) versus the characteristic impedance ratio (i.e., as a function of . These results are all derived from (2) and (6). The data from Figs. 5 and 6 form an important basis for the filter design in our study. The procedure to achieve the desired resonances is as follow. The first step is to determined the center frequency , which can be easily achieved by adjusting the dimensions (i.e., ) of

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Fig. 7. Equivalent-circuit model of the BCMC transition.

(i.e., and ) and center frequency with fixed 3-dB FBW of the filter is investigated. First, based on the homogeneous condition limits, the image impedance of the asymmetrical coupled lines is expressed as Fig. 5. Normalized resonant frequencies of the proposed resonator versus l l ) with various length L under the case of length L (i.e., L Z ;Z , and Z .

= 65

= = = 75

= 55

(7) where and are the characteristic impedances for the two modes of the asymmetrical coupled-lines. Secondly, the matrix of the equivalent-circuit model can be computed as follows from (8):

(8) The image impedance then derived as

of the equivalent-circuit model is

(9)

Fig. 6. Normalized resonant frequencies of the proposed resonator versus characteristic impedance ratio Z =Z with various characteristic impedance l l : mm, and L : mm.) Z L

(

= = =56

=36

the three-line coupled structure. Once is allocated, is determined. The second step is to employ and . Both resonances , , and under various cases can be finely tuned by is determined, and can be of . It is seen that once shifted within a wide range. As such, it is possible to employ a wideband operation to meet the UWB limits by selecting an appropriate I/O coupling structure. 2) BCMC Transition: For the wideband filters, tight couplings are essentially required for the I/O structures [37]. How% have been difficult to achieve ever, filters with due to the single layer coupled structure and fabricated process limits [14]. It is demonstrated that the tight coupling to meet the UWB operation can be easily obtained by the multilayer coupled scheme [22]. Therefore, to achieve strong tight coupling to meet the UWB operation requirements, the asymmetrical BCMC transition shown in Fig. 7 is employed in our study. This asymmetrical coupled scheme can be approximately modeled by the equivalent-circuit model depicted in Fig. 7(b) [38]. In this study, the relation between the characteristic impedances

As demonstrated in [39], the image impedance of the of the equivalent-circuit model is approximately equal to asymmetrical coupled-line section for (i.e., ), which corresponds to the center frequency of the bandpass filter. Therefore, from (7) and (9) (i.e., ), the following value can be obtained as: (10) Thus, the external quality factor pled lines is given as

of the asymmetrical cou-

(11) and 3-dB bandwidth Besides, the loaded quality factor (i.e., ) of a parallel coupled-structure is related by (12) It is well known that for the lossless case [22], and thus the following expression, can be derived from (11) and (12) as: (13)


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Fig. 9. Effects of the BCMC transition on the passband enhancement. (L = 6:65 mm, L = 2:4 mm, L = 6:8 mm, L = 1:5 mm, S = 0:2 mm, t = 0:8 mm, W = 1:3 mm, W = 1:2 mm, and W = 1:1 mm.)

Q

Fig. 8. (a) Simulated for the asymmetrical BCMC transition versus (b) Simulated for the asymmetrical BCMC transition versus .

Q

g

W

.

According to the above analysis, when and are specified, and can be determined. As such, the dimensions of the BCMC transition can be obtained. Fig. 8(a) depicts the simulated with the overlapped width (i.e., ) for difand ) of the coupled lines using the ferent sizes (i.e., EM-simulator IE3D [40]. It can be seen that the wider the or the wider and , the smaller the values. Besides, the effects of the coupled location (i.e., ) of the broadside-coupled structure on are discussed in Fig. 8(b). Once the dimensions of the BCMC transition are determined, the strong enough broadside coupling can be finely achieved to employ the wideband passband enhancement. Therefore, the insertion loss around the split frequencies is reduced and becomes smooth to meet the FCC limit, as shown in Fig. 9. In addition, it can be and ) are allocated seen that two transmission zeros (i.e., and occurred at the lower and upper sides of the passband, respectively. The mechanism of these two transmission zeros are discussed as follow. The lower transmission zero is employed by the tapped I/O feed-lines [41], which can be easily obtained by adjusting the location of the tapped line (i.e., ). Besides, the upper transmission zero is dominated by the physical length of the over-coupled end stage (i.e., ) [42], [43]. Fig. 10 shows the effects of these structures on the transmission zeros. It is seen that both transmission zeros can be finely tuned. As such, the passband selectivity of the UWB operation can be enhanced. 3) Dual-Notched Bands Operation: The dual-notched bands operation can be employed by the meander slot-lines and

t

Z

Fig. 10. (a) Effects of the tapped length on the lower transmission zero . on the upper transmission (b) Effects of the over coupled scheme length zero .

Z

L

CPW resonators simultaneously and independently, which meander are embedded in the CPW DMR. First, a pair of slot-lines is introduced to provide the lower notched band with the center frequency of , whereas the physical length of the slot-line [44] can be calculated as (14) where (15) (16)

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Fig. 12. Layout of the fabricated UWB bandpass filter with dual-notched bands. (g : mm, g : mm, g : mm, L : mm, L : mm, L : mm, L : mm, L : mm, : mm, L : mm, L : mm, L mm, L : mm, S : mm, t : mm, W : mm, W : S : mm, W : mm, W : mm, W : mm, mm, W and W : mm.)

= 02 = 08 = 24 = 62 = 34 = 6 05 = 13 = 4 15 = 63 = 23 = 10 = 10 8 =02 =02 =08 = 1 65 =16 =16 =12 = 1 25 =02 = 02

(i.e., ) the proposed filter with tuning electrical length (i.e., ) of the meander slot-line, and characteristics respectively. It is clearly observed that with increasing length , the first-notched band is shifted to lower frequency and , the bandwidth of the notched band with decreasing width slightly decreases. Meanwhile, the effects of the embedded (i.e., ) CPW resonator with adjusting electrical length and (i.e., under the case of mm) on the notched band are depicted in Fig. 11(c) and (d). It is found that the second-notched band is moved to higher frequency , whereas the notched bandwidth with decreasing length (i.e., decreases slightly with increasing ratio of increases). Therefore, dual-notched bands operation with narrow bandwidth and desired center frequencies can be finely adjusted and obtained. C. Filter Design and Experimental Results

Fig. 11. (a) Curve to relate the center frequency f of the first-notched band to meander slot-line length L . (b) Curve to relate the bandwidth of the firstnotched band to meander slot-line width W . (c) Curve to relate the center frequency f of notched band to embedded CPW resonator length L . (d) Curve to relate the bandwidth of notched band to embedded CPW resonator ratio of strip W to slot S under the case of S : mm.

=02

and and is the substrate dielectric constant and thickness of the dielectric substrate, respectively. Meanwhile, the length related to the second notched resonance provided by embedded CPW resonator can be simply expressed as the (17) where is the effective dielectric constant. In addition, the bandwidths of both notched bands can be properly controlled by tuning the characteristic impedances (i.e., and ) of the meander slot-lines and embedded CPW resonators. Fig. 11(a) and (b) shows the simulated frequency response of

Based on the investigation above, the procedure of the UWB bandpass filter with dual-notched bands is as follows. The first step is to determine the first three resonant modes occurring , center , and higher end around the lower end of the UWB band. Once is allocated, the first spurious can be obtained, which could dominate the stopband bandwidth of the proposed filter. It is demonstrated [i.e., Figs. 3–6 and (2) and (6)] that the CPW DMR can position these resonances to meet the UWB operation. The second step is to obtain the strong tight coupling for the passband enhancement around the first three resonances to meet the UWB operation limits. As discussed in the previous sections, once the center frequency is determined, of the three-line coupled scheme can be obtained. The dimensions of the broadside-coupled microstrip structure can then be derived from (13) with the desired FBW. The third step is to introduce two transmission zeros to enhance the passband selectivity. These transmission zeros can be finely adjusted by the tapped-line and over-coupled scheme. The fourth step is to occur the dual-notch responses in the UWB meander slot-lines and embedded operation using the CPW resonators, which can avoid the interferences from the WLAN signals (i.e., 5.2 and 5.8 GHz). As such, the dimensions of the UWB bandpass filter with dual-notched bands can be determined, as shown in Fig. 12. The -parameters and groupdelay response measurements are performed using an Agilent 5230A network analyzer over the frequency range from 10 MHz


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Fig. 13. Measured and simulated results of the fabricated UWB bandpass filter with dual-notched bands.

to 16 GHz. Fig. 13 demonstrates the simulated and measured results of the proposed filter, where an excellent agreement is obtained. The proposed filter exhibits a good UWB bandpass performance. The measured results show the emergence of the first-notched band at 5.23 GHz with 21.9-dB insertion loss and 10-dB notched FBW of 1.91% (i.e., 5.18–5.28 GHz). The measured second-notched band is located at 5.81 GHz with a rejection level of 23.97 dB and a 10-dB bandwidth of 2.07% (i.e., 5.75–5.87 GHz). The measured minimum insertion loss is found to be 0.69, 1.49, and 0.63 dB in the lower, middle, and upper passbands, respectively. The measured return loss is better than 10.3 dB among the passbands. Two transmission zeros are found at 0.81 and 10.5 GHz, respectively. The upper stopband is observed from 10.3 to 15.2 GHz. Meanwhile, the implemented filter exhibits a flat group-delay response below 0.59 ns and group-delay variation less than 0.36 ns over the whole passband. In addition, the proposed filter has the merit of a very compact size of 12.6 mm 16.2 mm (i.e., 0.359 by 0.462 , where is the guided wavelength of microstrip structure at the center frequency of 6.52 GHz), including the 50- microstrip feed-lines. III. UWB BANDPASS FILTER WITH QUAD-NOTCHED BANDS A. Schematic and Principle To cancel the interferences such as the WiMAX signals (i.e., 3.5-GHz band), WLAN signals (i.e., 5.2 and 5.8 GHz) and 6.8-GHz-band RFID communication, which operate in the UWB passband, the UWB bandpass filter with quad-notched bands is further developed. Fig. 14 depicts the configuration of the proposed UWB bandpass filter with quad-notched bands. Compared with the proposed filter shown in Fig. 1, the meander defected microstrip structures (MDMSs) are introduced. It can be seen that the MDMS is composed of two meander slot-lines embedded on the microstrip feed-lines. This structure has the similar characteristics and frequency responses as the spur-line scheme [45], [46]. The notched center frequencies (i.e., and ) can be finely and separately achieved while the electrical length ( and ) of the meander slot-lines is chosen about at and , respectively. Thus, the physical

Fig. 14. Configuration of the UWB bandpass filter with dual-notched bands. (a) 3-D view. (b) 2-D view. (c) Transmission-line model.

length of the meander slot-lines in the MDMS can be simply derived as (18) where is the effective dielectric constant. Besides, the bandwidth of the notched bands can be finely tuned by the characterand ) of the meander slot-lines in istic impedances (i.e., the MDMS. The effects of the MDMS on the center frequency and bandwidth are depicted in Fig. 15. It is seen that with in(i.e., ) the notched bands are creasing the length shifted to lower frequency and with decreasing the width , the bandwidth of the notched bands slightly decreases. Meanwhile, it is notable that the narrowest bandwidth can be obtained when there are two corners in the MDMS, as shown in Fig. 15(c). Therefore, the specific notched bands to meet the requirements can be properly achieved. B. Filter Design and Experimental Results The design procedures of the UWB bandpass filter with quad-notched bands are as follows. The first step is to obtain the UWB operation, which meets the FCC limit. Following the mechanism in Section II, the UWB bandpass filter with good passband and stopband performances can be finely achieved. The second step is to meet the requirement of the notched

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Fig. 16. Layout of the fabricated UWB bandpass filter with quad-notched bands. (g : mm, g : mm, g : mm, L : mm, L : mm, L : mm, L : mm, L : mm, : mm, L : mm, L : mm, L : mm, L : mm, L : mm, S : mm, S : mm, L : mm, S : mm, t : mm, W : mm, S : mm, W : mm, W : mm, W mm, W : mm, and W : mm.) W

= 02 = 3 35 = 13 = 97 = 01 =1 =02

Fig. 15. (a) Effects of the MDMS length L on the center frequency of the notched band. (b) Effects of the MDMS width S on the center frequency of the notched band. (c) Effects of the MDMS corners on the center frequency of the notched band.

bands, which can reject interferences such as the WiMAX signals (i.e., 3.5 GHz), WLAN signals (i.e., 5.2 and 5.8 GHz), and 6.8-GHz band RFID communication. The desired center frequencies and bandwidth of the WLAN notched bands can meander be finely achieved by restructuring the embedded slot-lines and CPW resonators on the CPW layer. Besides, the notched bands to avoid the WiMAX signals and RFID communication interferences can be finely provided by the MDMS on the microstrip layer. Thus, the UWB bandpass filter with quad-notched bands can be easily employed. Fig. 16 shows the layout of the UWB bandpass filters with quad-notched bands, which has a compact size of 12.1 mm 16.2 mm (0.345 by 0.462 , where is the microstrip guided wavelength at 6.52 GHz), including the 50- microstrip feed-lines. The measured results are carried out using the Agilent 5230A network analyzer over the frequency range from 10 MHz to 16 GHz. Fig. 17 shows the simulated and experimental results of the proposed filter, where excellent agreement is obtained. It is seen that the quad-notched bands are achieved in the UWB operation. The first notched band is located at 3.51 GHz with insertion loss of 20.8 dB and a 10-dB notched FBW of 1.14% (i.e., 3.49–3.53 GHz). The second notched band is allocated at

= 08 = 18 3 = 4 35 = 11 1 = 02 = 16 =02

= 22 = 71 = 60 = 02 = 06 = 12

= 58 = 6 25 = 21 = 01 = 1 75 = 1 35

Fig. 17. Measured and simulated results of the fabricated UWB bandpass filter with quad-notched bands.

5.23 GHz with a rejection level of 21.1 dB and a 10-dB FBW of 1.53% (i.e., 5.19–5.27 GHz). The third notched band is measured at 5.81 GHz with insertion loss of 27.5 dB and a 10-dB FBW of 2.75% (i.e., 5.73–5.89 GHz). Meanwhile, the fourth notched band is observed at 6.81 GHz with a rejection level of 24.6 dB and a 10-dB FBW of 1.17% (i.e., 6.77–6.85 GHz). The measured insertion losses of five sub passbands are 0.76 dB at 3.3 GHz, 0.89 dB at 4.4 GHz, 1.21 dB at 5.5 GHz, 0.81 dB at 6.4 GHz, and 0.65 dB at 8.5 GHz, respectively. The typical return loss is about 13.3 dB in the whole passband. Two transmission zeros are found at 0.89 and 10.67 GHz, respectively. The upper stopband is observed from 10.3 to 15.1 GHz. In addition, the implemented filter exhibits a flat group-delay response below 0.46 ns and group-delay variation less than 0.21 ns over the whole passband. IV. CONCLUSION In this paper, compact UWB bandpass filters with ultra narrow dual- and quad-notched bands are proposed using the dual-layer coupled structure. The UWB operation can be easily achieved by the BCMC DMR. Meanwhile, the embedded meander slot-lines and CPW resonators on the CPW layer and the MDMS on the microstrip layer are introduced to cancel the existing interferences (i.e., 3.5-GHz WiMAX signals,


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(30)

5.2/5.8-GHz WLAN signals, and 6.8-GHz band RFID communication), respectively. With good frequency performances and compact sizes, the proposed UWB bandpass filters with dualand quad-notched bands are attractive to the UWB systems for the purpose of blocking unwanted radio signals. APPENDIX The impedance parameters of the asymmetrical coupled lines for the homogeneous condition can be derived as follows [36]:

(19) (20) (21)

(22)

(23)

(24) (25) where (i.e., ) are the self-admittance per unit length , and is the of line in the presence of line mutual admittance per unit length, respectively. Examination of and in terms of line constant reveals that the even- and odd-mode impedances of the two lines as defined by for line 1 and , for line 2 as follows [47], respectively: (26) (27) (28) Meanwhile,

can be calculated as [48] (29)

where and (i.e., ) are the self-inductance and capacitance per unit length of line in the presence of line , and and are the mutual inductance and capacitance per unit length, respectively, shown in (30) at the top of this page.

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Xun Luo (S’08) was born in Sichuan, China. He received the B.E. degree (with honors) in electronic engineering from the University of Electronic Science and Technology of China (UESTC), Chengdu, China, in 2005, and is currently working toward the Ph.D. degree in electrical engineering at UESTC. From 2002 to 2005, he was with Chengdu Sine Science and Technology Ltd., where he was the Deputy Team-Leader of the 0.2–20-GHz UWB front-end project. From 2005 to 2006, he was with the 802nd Research Institute of Shanghai Academy of Spaceflight Technology (SAST), where he involved in the measurement of microwave receivers. In 2007, he investigated the radio front-end devices of the multimode and multicard mobile systems with Philips Research East-Asia, Shanghai. He has filed and holds numerous China patents in mobile systems and antenna design. His research interests include RF/microwave/millimeter-wave transceiver design, CMOS/BiCMOS RF integrated-circuit (RFIC), and monolithic-microwave integrated circuit (MMIC) applications, and low-temperature co-fired ceramics (LTCC) and printed circuit board (PCB) processes. Mr. Luo is a technical reviewer for several prestigious international journals, including the IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES and IEEE MICROWAVE AND COMPONENTS LETTERS. He was the recipient of the 2008 Huawei Scholarship, the 2009 Intel Fellowship, the 2009 Rohde & Schwarz Scholarship, and the 2010 Mediatek Scholarship.

Jian-Guo Ma (M’96–SM’97) received the B.Sc. and M.Sc. degrees from Lanzhou University, Lanzhou, China, in 1982 and 1988, respectively, and the Doctoral degree in engineering from Duisburg University, Duisburg, Germany in 1996. From April 1996 to September 1997, he was with the Technical University of Nova Scotia (TUNS), Halifax, NS, Canada, as a Postdoctoral Fellow. From October 1997 to November 2005, he was with Nanyang Technological University (NTU), Singapore, as a faculty member, where he was also the founding Director of the Center for Integrated Circuits and Systems, NTU. From December 2005 to October 2009, he was with the University of Electronic Science and Technology of China (UESTC), Chengdu, China. Since November 2008, he has concurrently been the Technical Director for the Tianjin Integrated Circuit (IC) Design Center, and since October 2009, he has been the Dean for the School of Electronic Information Engineering, Tianjin University. He has authored or coauthored about 245 technical papers and two books. He holds six U.S. patents and 15 filed/granted China patents. His research interests include RFICs and RF integrated systems for wireless, RF device characterization modeling, MMIC, RF/microwave Ccircuits and systems, electromagnetic interference (EMI) in wireless, RFID, and wireless sensing networks. Dr. Ma was the associate editor of IEEE MICROWAVE AND COMPONENTS LETTERS (2004–2005). He is one of the founding members of the IEEE Chengdu Section and is the external liaison chair and Technical Activity chair for the IEEE Chengdu Section. He founded the IEEE Electron Device Society (EDS) Chengdu Chapter and is the Chapter chair. He was the recipient of the prestigious 2008 Changjiang (Yangtze) Chair Professorship Award of the Ministry of Education of China. He was a 2008 Distinguished Young Investigator of the National Natural Science Foundation of China.


Kiat Seng Yeo (M’00–SM’09) received the B.Eng. degree (with Honors) and Ph.D. degree from Nanyang Technological University, Singapore, in 1993 and 1996, respectively, both in electrical engineering. In 1996, he joined the School of Electrical and Electronic Engineering, Nanyang Technological University, where he is Head of the Division of Circuits and Systems. He is currently a Professor with the School of Electrical and Electronic Engineering and Founding Director of VIRTUS, a new research center of excellence jointly set up by Nanyang Technological University and the Singapore Economic Development Board. He is a widely known authority on low-power IC design and a recognized expert in CMOS technology and RF IC design. As a result of his innovative pioneering work in the field of IC design, he has successfully attracted over S$30 million of external research funding from various funding agencies and industry over the last five years. He has authored over 300 refereed papers in journals and conferences. He authored six books and three book chapters. He holds 25 patents, including two patents for the world’s smallest integrated transformer and several patents for 60-GHz applications. Dr. Yeo serves on the Editorial Board of the IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES. He is a Board member of the Singapore Semiconductor Industry Association (SSIA). He provides consultation to multinational corporations. He was general chair, co-general chair, and technical chair of several international conferences. He has given keynotes and invited presentations at various scientific meetings, workshops and seminars. He was the recipient of the Public Administration Medal (Bronze) on National Day 2009 by the President of the Republic of Singapore. He was also the recipeint of the Nanyang Alumni Award in 2009.

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Er-Ping Li (S’91–M’92–SM’01–F’08) received the Ph.D. degree in electrical engineering from Sheffield Hallam University, Sheffield, U.K., in 1992. From 1989 to 1992, he was a Research Associate/ Fellow with the School of Electronic and Information Technology, Sheffield Hallam University. From 1993 to 1999, he was a Senior Research Fellow, Principal Research Engineer, and the Technical Director of the Singapore Research Institute and Industry. In 2000, he joined the Singapore National Research Institute of High Performance Computing, as a Principal Scientist and Director of the Electronic and Photonics Department. Since 2010, he has been a Professor with Zhejiang University, Hangzhou, China. He has authored or coauthored over 200 papers published in referred international journals and conferences. He holds and has filed a number of U.S. patents. His research interests include fast and efficient computational electromagnetics, microscale/nanoscale integrated circuits and electronic packaging, and plasmonic technology. Dr. Li is a Fellow of the U.S. Electromagnetics Academy. He was the IEEE Electromagnetic Compatibility Distinguished Lecturer from 2007 to 2008. He was an associate editor for the IEEE MICROWAVE AND WIRELESS COMPONENTS LETTERS (2006–2008). He is an associate editor for the IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY. He was a guest editor for the IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY and the IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES. He has been a technical chair and session chair for many international conferences. He was the president for the International Zurich Symposium on EMC, Singapore (2006 and 2008), the general chair for the Asia–Pacific EMC Symposium (2008 and 2010), the Technical Program Committee chair for the Asia–Pacific Microwave Conference (2009), and the chairman of the IEEE EMC Singapore Chapter (2005-2006). He has been invited to give numerous invited talks and keynote speeches at various international conferences and forums. He was the recipient of the Changjiang (Yangtze) Chair Professorship Award of the Ministry of Education in China and the IEEE Technical Achievement Award (EMC) and Singapore Institution of Engineers Singapore Prestigious Engineering Award.