Design of a Shape-Constrained Dual-Band Polygonal ... - IEEE Xplore

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IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 57, NO. 3, MAY 2008

Design of a Shape-Constrained Dual-Band Polygonal Monopole for Car Roof Mounting Matteo Cerretelli, Vasco Tesi, and Guido Biffi Gentili

Abstract—This paper describes the design of a new compact dual-band roof antenna (which is capable of operating in both U.S. and European mobile telephone bands) for automotive applications. The flipper shape of the antenna and its dimensional constrains are chosen according to prevailing automotive market trends. The chosen basic geometry of the radiating element is a planar printed monopole structured as a vertical fan with two narrow strips folding toward the ground plane and etched on a low-cost, medium-permittivity thin substrate. The geometrical variations of the basic radiating structure are introduced afterward to constrain the antenna shape under the assigned profile and to allow easy integration of a small Global Positioning System patch antenna without affecting radiation and bandwidth performances. The integration of a matching network directly over the etched monopole trace allows a good input matching over the lower and upper operative bands to be obtained, slightly affecting the radiation efficiency of the whole antenna. Index Terms—Impedance matching, mobile antennas, multifrequency antennas, road vehicles.

I. I NTRODUCTION

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HE DRAMATIC growth in mobile wireless communications is now attracting much attention on small-sized and hidden multiband antennas. The ever-increasing number of personal communications services and telematics available in modern cars calls for multiband–multiservice integrated antennas that are structured to be fitted in the vehicle with minimum impact on aesthetics and aerodynamics. The characteristic and radiation properties of the classical monopole antenna mounted on the roof are well suited for obtaining azimuthal gain uniformity and protection of passengers from unwanted electromagnetic (EM) pollution. However, the automotive industry has been pressing for increasingly cheaper solutions without sacrificing the radiating performances of the antenna system. Antennas printed on low-cost substrates (FR4 or equivalent substrate) are the best candidates to conjugate compactness, easy fabrication, inexpensive production, and radiation performances. Multiband and wideband monopoles have been a recent topic of interest; among these are the fractal Sierpinski gasket [1] and the circular disk monopole [2]. Multiband multiple-ring monopole antennas evolving from the Manuscript received November 11, 2005; revised December 19, 2006, September 19, 2007, and September 25, 2007. This work was supported in part by ASK Industries S.p.A. The review of this paper was coordinated by Prof. S. Chaudhuri. M. Cerretelli is with ASK Industries S.p.A., 42100 Reggio Emilia, Italy (e-mail: [email protected]). V. Tesi and G. B. Gentili are with the Dipartimento di Elettronica e Telecomunicazioni, Università degli Studi di Firenze, 50139 Florence, Italy (e-mail: [email protected]; [email protected]). Digital Object Identifier 10.1109/TVT.2007.912153

Fig. 1. Geometry of the basic antenna configuration. The dashed line represents the even symmetry axis.

principles of the previous antenna structures have more recently been proposed [3]. The most stringent requirement concerns the lower operation frequency that substantially determines the height of a selfresonant monopole and, thus, the overall antenna height above the vehicle roof. The aim of this paper is to design an antenna with an overall maximum height of 50 mm, i.e., to be compliant with the requirements of the automotive market. The new antenna should be capable of operating in the American and European mobile phone bands, i.e., from the Advanced Mobile Phone Service (starting at 824 MHz) up to the Global System for Mobile Communications [(GSM) ending at 960 MHz], and from the Digital Cellular System [(DCS) starting at 1.71 GHz] up to the Universal Mobile Telecommunications System [(UMTS) ending at 2.17 GHz], with acceptable radiation efficiency. The proposed basic geometry is a polygonal symmetrical monopole printed on a medium-permittivity substrate with two narrow strips folding to the ground plane. It is expected that the presence of such strips might offer more freedom to optimize the overall antenna characteristics. Starting from this basic radiating element, several variations of the initial geometry have been numerically and experimentally analyzed to allow the best performances with maximum compactness. The experimental results of a prototype built according to the optimized geometry are finally introduced and compared with numerical predictions. II. B ASIC P OLYGONAL M ONOPOLE AND I TS R ADIATION P ROPERTIES The basic geometry of the proposed antenna, shown in Fig. 1, is a folded polygonal monopole (FPM) printed on a relatively thin (1.2 mm) FR4 substrate [4]. A probe feed is introduced to excite the monopole by directly connecting its lower vertex to the inner conductor of the input coaxial cable. A couple

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CERRETELLI et al.: SHAPE-CONSTRAINED DUAL-BAND POLYGONAL MONOPOLE FOR CAR ROOF MOUNTING

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lower band, the gain pattern is almost omnidirectional, as shown in Fig. 3(a). At higher frequencies, the array effect of the residual strip currents becomes perceptible because the distance between them approaches half-wave length. This phenomenon, as evidenced in Fig. 3(b) with a reduction of the gain toward 90◦ and 270◦ with respect to the gain toward 0◦ and 180◦ , is negligible at frequencies below 2 GHz and can be tolerated for UMTS band operation. Fig. 2.

Surface current amplitude at (a) 900 MHz and (b) 1800 MHz.

III. D ESIGN S TEPS of vertical strips connecting the upper vertices to the ground plane allows the radiating radio-frequency current at the lower frequency band to be folded toward ground. The working principle is similar to those described in [5], although, in this case, a printed and more compact version is used. It is worth noting that the radiating structure comes from the superposition of three radiating elements, i.e., a fan monopole constituting the central part of the antenna and two folded monopoles represented by the lateral vertical strips. The fan monopole is directly derived from the classical butterfly antenna and presents the same broadband characteristics. The aperture angle at the feeding point is chosen to obtain an input impedance close to 50 Ω at higher frequencies, whereas, at half of the fan height, the classical “V” shape profile is altered to reduce the horizontal dimension without significantly affecting the input impedance. The lateral folding strips reduce the frequency of the first natural resonance of the fan monopole and improve its radiating performances without increasing the monopole width. The resulting symmetrical structure presents two distinct resonances near the desired frequency bands and can then be assumed to be the starting geometry for the design of the flippershaped antenna. The initial dimensions of 52 mm × 52 mm has been chosen as an acceptable compromise between the desired resonant frequencies (see Section III) and the maximum acceptable dimensions for the automotive market (height of 50 mm, length as short as possible). At the lower frequency band, the vertical strips carry a strong symmetry in phase electrical currents, as shown in Fig. 2(a), and because of their small distance compared to the wavelength in a vacuum (about 0.15 λo), the resulting radiation pattern is quite omnidirectional in azimuth. With increasing frequency [Fig. 2(b)], the main contribution to radiation is due to the currents flowing on the fan vertex and borders, while the current density on the strips is rather reduced. The numerical simulations have been performed by a CST EM 3-D simulator [6]. The gain pattern required for automotive applications should present a radiation mainly concentrated toward low elevation angles (XZ plane with respect to Fig. 2) and very omnidirectional in this plane, with a mean gain of over 360◦ in the horizontal rotation that is as close as possible to the gain of the corresponding λ/4 reference monopole, with a maximum available degradation of 1–2 dB. The gain patterns of the proposed basic antenna mounted on an infinite ground plane are shown in Fig. 3(a) and (b) for three elevation angles (0◦ , 30◦ , and 60◦ ) at different operating frequencies in the GSM and UMTS bands. In the

Despite the fair radiation characteristics of the basic FPM antenna, the bandwidths in the lower and higher operational bands do not comply with the specifications of the automotive market, which requires an input voltage standing wave ratio (VSWR) of below 2 (conventionally accepted as a good compromise between performances and dimensional constrain) in the 824–960- and 1710–2170-MHz bands. The input matching and the azimuthal pattern uniformity of the FPM antenna can be improved by breaking the symmetry of the planar polygonal monopole. By removing one of the two vertical strips and by eliminating part of the corresponding upper corner, as depicted in Fig. 4(a), the shape of the monopole looks more like that of a flipper, and it is possible to maintain a wide aperture angle at the bottom vertex of the fan, improving the matching of the antenna, as depicted in Fig. 4(b), where the input return loss of the modified geometry is compared with that of the basic FPM antenna. The improvement of the input matching is clearly evidenced, i.e., associated with a slight increase of the first resonant frequency. Furthermore, the gain pattern of the modified antenna essentially remains omnidirectional in azimuth, with the frequency varying from 800 to 2100 MHz. It is worth noting that the new geometry easily allows the placement of a GPS ceramic patch antenna, as depicted in Fig. 4(a). The presence of the GPS patch antenna does not significantly modify the behavior of the telephone antenna, as shown by some simulations, whereas the GPS radiation diagram is modified by the closeness of the proposed antenna. Despite this, the presence of the GPS antenna will not be taken into account in further simulations because mechanical and aesthetical constrains allow neither an increase in the distance between the two antennas nor the modification of the relative position. Consequently, the performance of the GPS antenna has to be accepted once the geometry of the complete system is fixed. A prototype of the asymmetric FPM [shown in Fig. 5(a)] was fabricated and tested. Fig. 5(b) compares the measured and simulated return losses of the proposed FPM antenna on a finite-sized ground plane (a square plate with side as length as a wavelength at 800 MHz, which is the lower frequency of interest) and its corresponding numerical model with an infinite and finite ground plane, with the same dimensions as the real one. Comparing the resonant frequencies of the measured prototype with those of the numerical model, a very small difference of between 1% for the higher band and 3% for the lower band results (calculated as a drift respect to the center frequency of each band), which allows validation of both the

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Fig. 3. Gain pattern of the basic FPM antenna at (a) 900 MHz and (b) 2 GHz.

Fig. 6. Layout constrains for (dark area) the new antenna compared with (grey area) the previous dimensions.

Fig. 4. (a) Shape of the modified FPM antenna and (b) comparison of its input return loss in decibels (continuous line) with that of the basic FPM antenna (dotted line). Horizontal line shows VSWR = 2.

Fig. 5. (a) Realized prototype and (b) measured return loss of the prototype compared with numerical results. (Continuous line) Prototype. (Dotted line) Infinite ground plane model. (Dashed-dotted line) Finite ground plane model.

EM simulator results and the accuracy of the measurement made in an anechoic chamber. The final step of the design was to further reduce the fin width to fully comply the size constrains and to decrease the shadowing effects on the GPS patch antenna, as shown in Fig. 6. This very compact version of the antenna maintains the radiative properties of the original monopole but does not allow the required lower and upper operating bandwidths to be obtained. The specifications were definitively satisfied by embedding a matching network in the same vertical printed circuit board that accommodates the monopole and by optimizing the values of its lumped and distributed elements. The optimization procedure was carried through the combined use of a full-wave simulator (CST Microwave Studio [6]) and a microwave-circuit computer-aided design (AWR Microwave Office [7]). The matching network was realized by embedding some lumped and distributed reactive elements into the monopole body, as depicted in Fig. 7(a), showing the layout on the antenna. Fig. 7(b) shows the circuital structure of the network. After the feeding point, the capacitor C2 is used to decouple the signal, and the direct current is used to test for the presence of the antenna (by means of 10-kΩ resistor that is not shown in the scheme). The parallel inductive stub Lp is used to match the antenna input impedance, whereas the series line Ls helps to maintain a small height of the antenna by inductively loading it at its base. The parallel resonators L1–C1 and L2–Cp at the ends of the folding strip are used to selectively connect the strip itself to the rest of the antenna to


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Fig. 8. (a) Numerical model in CST MicroWave Studio and (b) photograph of the realized prototype close to a 2 C coin.

Fig. 7. (a) Embedded matching network layout showing element positions and values on the CST Microwave Studio model and (b) the network circuital structure.

obtain the desired input matching. Many preliminary full-wave simulations were done to assess a very definitive geometry layout and the proper placement of the matching elements. The values of the distributed inductor Ls and capacitor Cp were preliminarily determined and fixed. However, the final step of optimizing the values of the remaining five lumped and distributed (Lp) components represents the very difficult task of having to use the full-wave simulator due to the excessive numerical overload. To increase the efficiency of the optimization process, we decided to move the optimization task from EM to circuit level using the AWR Microwave Office [7]. All the elements to be optimized were substituted into the fullwave model by the corresponding discrete ports, and then, only one multiport full-wave analysis was carried out. The inductive stub Lp was truncated at the section indicated in Fig. 7(a), whereas in the circuital model, a series microstrip was inserted at the same virtual section, allowing the tuning of its length. The multiport scattering matrix resulting from the full-wave simulation was exported in Touchstone format and directly transferred to the circuit simulator. The port representing the antenna input was connected to the feeding coaxial line, whereas the remaining ports were connected to the equivalent circuit representing the tunable part of the distributed matching network. In such a manner, a very fast optimization was made possible, fully exploiting the flexibility and numerical efficiency of the AWR Microwave Office optimizer. Among the available optimization algorithms, the “genetic with Gaussian mutation” was chosen. The “maximum iteration” parameter

was set to 10 000, whereas the “population size” was 100, and the “standard deviation %” was 30. The optimized component values were then reinserted into the full-wave simulator to check the results and verify the properness of the approach because the proposed optimization procedure disregards some EM coupling phenomena. Fig. 8(a) shows the antenna architecture implemented in Microwave Studio, which comprises a simplified but representative model of the plastic cover. The simulated gain pattern of the antenna over an infinite ground plane at the end of the optimization procedure, shown in Fig. 9, is omnidirectional at 900 MHz, whereas at 2 GHz, it shows a slight reduction (2 dB) of the radiated power in the frontal direction (90◦ ) for low elevation angles, which is a good compromise with the significant size reduction obtained with respect to the basic FPM antenna. The prototype of the final version of the antenna, which is depicted in Fig. 8(b), was fabricated to test its compliance to the specifications, also in the presence of both the GPS patch antenna and the plastic cover. Fig. 10(a) shows the comparison between measured and simulated antenna return losses; the curves agree well, and no tuning was needed on the prototype to comply with the specifications (all the bands of interest are below the showed limit line at −9.54 dB, which corresponds to VSWR = 2). The small differences between numerical and measured results can be attributed to the mechanical imperfections of the prototype and the additive losses introduced by the cover, which was fabricated using a fast prototyping milling machine and an unfit plastic material. The antenna was mounted on the roof of a car, and the horizontal gain function was measured in an open site for an elevation angle of about 4◦ above the horizon. The measured samples with 2◦ step were taken in the full 360◦ rotation, and the angular average for each frequency band was calculated. For each operating band, the angular average gain of the FPM antenna was normalized to that of a corresponding λ/4

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Fig. 9. Simulated gain pattern of the final FPM antenna at (a) 900 MHz and (b) 2 GHz.

improved by choosing a suitable geometry of the planar radiating monopole and by introducing a simple matching network embedded into the radiating element itself. It has been shown that the final optimization process can be performed at circuit level rather than at EM (full-wave) level, thus allowing an optimum design without numerical overload. It is worth noting that the suggested two-phase optimization procedure is applicable, strictly speaking, only if the radiating element is not electromagnetically coupled with the matching network. In practice, this was obtained by using very small lumped and distributed elements to maximally reduce the unwanted EM couplings. R EFERENCES

Fig. 10. (a) Measured return loss (continuous) compared with the simulated one (dashed) and (b) measured horizontal gain (averaged on 360◦ ) of the antenna prototype (mounted on the roof of a car) relative to a couple of λ/4 monopoles.

reference monopole mounted in the same position of the antenna under test. The measured results in Fig. 10(b) show that in the whole GSM band, the proposed antenna presents an average gain of only 0.9 dB lower than the corresponding reference monopole, and the difference is reduced to 0.6 dB in the DCS band. For the UMTS band, the new antenna presents a gain that is 0.35 dB greater than the reference monopole. The measures demonstrate a fairly good radiation efficiency, which will be reasonably increased in the final version, where the cover will be realized with low-loss plastics. IV. C ONCLUSION This paper has presented a new compact dual-band antenna for rooftop automotive use and demonstrated that the performances of the proposed antenna can be significantly

[1] C. Puente, J. Romeu, R. Pous, X. Garcia, and F. Benitez, “Fractal multiband antenna based on Sierpinski gasket,” Electron. Lett., vol. 32, no. 1, pp. 1–2, 1996. [2] S. Hando, M. Ito, H. Seki, and Y. Jinbo, “A disk monopole antenna with 1:8 impedance bandwidth and omnidirectional pattern,” in Proc. ISAP, Sapporo, Japan, 1992, pp. 1145–1148. [3] C. T. P. Song, P. S. Hall, and H. Ghafouri-Shiraz, “Multiband multiple ring monopole antennas,” IEEE Trans. Antennas Propag., vol. 51, no. 4, pp. 722–729, Apr. 2003. [4] European Patent Application EP1471599 and Chinese Patent Application CN1551410. [5] K. L. Lau, P. Li, and K. M. Luk, “A monopolar patch antenna with very wide impedance bandwidth,” IEEE Trans. Antennas Propag., vol. 53, no. 2, pp. 655–661, Feb. 2005. [6] MW Studio 5 by CST Computer Simulation Technology, Darmstadt, Germany. [7] Microwave Ofﬁce 6. Applied Wave Research, Inc.

Matteo Cerretelli was born in Florence, Italy, on May 31, 1973. He received the degree in electronic engineering from the University of Florence in 1999. In 2000, he was with the Department of Electronic and Telecommunications, University of Florence. He is currently with ASK Industries S.p.A., Reggio Emilia, Italy. His main research interests are in the analysis and design of automotive antennas, microwave sensors, and biological applications of microwaves.


Vasco Tesi was born in Pistoia, Italy, on September 26, 1959. He received the Laurea degree in electronic engineering from the Università degli Studi di Firenze, Florence, Italy, in 1989. He is currently a Research Fellow with the Dipartimento di Elettronica e Telecomunicazioni, Università degli Studi di Firenze, where he is working on the development of microwave sensors for industrial process monitoring. His research interests are in biomedical applications, radio frequency and microwave circuit design, and the design of radio devices and systems.

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Guido Biffi Gentili was born in Lucca, Italy, on August 9, 1943. He received the Ph.D. degree in electronic engineering from the University of Pisa, Pisa, Italy, in 1970. In 1976, he joined the School of Engineering, University of Florence, Florence, Italy, where he is currently a Full Professor of electromagnetic theory and techniques and the Chair of the Dipartimento di Elettronica e Telecomunicazioni. He has authored over 100 refereed journals and conference proceedings, reports, and book chapters. He has been a scientific consultant to many Italian companies, working in the fields of defense and consumer and medical electronics. He has authored 15 national and international patents on ultrasonic and microwave sensors, vehicular compact antennas, and RFID technologies. He is currently responsible for many research projects being done by public and private companies in the fields of microwave engineering. His recent research activities have focused on the modeling and design of active and passive multilayer planar antennas and near-field sensors, wireless communication systems, millimeter-wave imaging techniques, industrial applications of high-frequency high-power EM fields, and microwave and RF hyperthermia.