Design of Multiple-Input Power Converter for Hybrid ... - IEEE Xplore

IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 20, NO. 5, SEPTEMBER 2005

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Design of Multiple-Input Power Converter for Hybrid Vehicles Luca Solero, Member, IEEE, Alessandro Lidozzi, Student Member, IEEE, and Josè Antenor Pomilio, Senior Member, IEEE

Abstract—This paper deals with designing and sizing of a multiple-input power electronic converter (MIPEC) to be used in an electric vehicle propulsion system that includes a fuel cell (FC) generator and a combined storage unit. The combined storage unit is composed by an ultracapacitors tank (UC) and a battery unit (BU). MIPEC is responsible for power-flow management on-board the vehicle for each mode of operation. Specifications for MIPEC designing come out from many considerations concerning traction drive and reference driving cycle, on-board power source and storage unit characteristics. However, to date sizing and configuration of both storage units and on-board generators are directly related to traction drive and driving profile (i.e., vehicle performances and characteristics) and no relation with power electronic interface is considered during preliminary design. Then, power electronic interface is selected in order to fit traction drive requirements with power source and storage unit characteristics; as a consequence converter mode of operation lacks of optimization, as well dynamic behavior and efficiency cannot be maximized. In this paper, MIPEC design and power source and storage unit selection are achieved at the same project stage according to traction drive requirements. Experimental results on 60-kW power electronic interface are presented. Index Terms—Control design, dc–dc converter, fuel cell, ultracapacitors.

I. INTRODUCTION

P

RESENT research concerning electric vehicles (EV) and hybrid-electric vehicles (HEV) concentrate in the search for a compact, lightweight, and efficient energy storage system that is both affordable and has acceptable cycle life. The traction system, composed by electric motor, inverter, and associated control circuitry is not the limiting factor to obtain high performance and to permit large-scale production of such vehicles. Attention is now increasingly focused on fuel cell (FC) and hybrid technologies as a way of producing breakthrough vehicles with alternative power plants. A number of auto makers see fuel cell powered vehicles as the ultimate route to achieving sustainable long-term alternative propulsion systems. A number of drive-train architectures have recently been proposed to combine two or more on-board generation units and storage units

Manuscript received February 17, 2004; revised February 11, 2005. Recommended by Associate Editor X. Xu. L. Solero and A. Lidozzi are with the Department of Mechanical and Industrial Engineering, University of Rome, Rome, Italy (e-mail: [email protected]). J. A. Pomilio is with the School of Electrical and Computer Engineering, State University of Campinas, Campinas, Brazil (e-mail: [email protected]. unicamp.br). Digital Object Identifier 10.1109/TPEL.2005.854020

and to overcome constraints related to fuel consumption, pollution, vehicles’ long distance capability. Interfacing of traction drive requirements with characteristics and modes of operation of on-board generation units and storage units calls for suitable power electronic converter configuration and control. In this paper, a three-inputs/one-output converter is proposed for a propulsion system where the generation unit is a 18 kW FC and the combined storage unit is formed by lead-acid batteries and ultracapacitors (UCs); however, same converter configuration is appropriate also for different either generation units or storage units. In terms of power sources, the proton exchange membrane FCs are being increasingly accepted as the most appropriate supply for EVs [1], [2] because they offer clean and efficient energy without penalizing performance or driving range. A battery storage unit (BU) can be combined with the FC stack to achieve the maximum efficiency for the FC system. The BU delivers the difference between the energy required by the traction drive and the energy supplied by the FC system. In such a system the BU has to deal with power peaks being on demand during either acceleration or braking phases. Such transients result in a hard constraint for the battery unit, which increases the losses and temperature, and reduces its lifetime. Thereby, it is desirable to minimize these power peaks by introducing an additional auxiliary power device: the ultracapacitors [3], which present high power density, obtain regeneration energy at high efficiency during decelerations and supply the stored energy during accelerations. In spite of reaching thousands of Farads, the UCs support very low voltages (1–2.5 V). A stack of series-connected UCs can produce an equivalent capacitor of tens of Farads that is able to hold up tens of Volts. The UC stack must supply the power required in excess of the FC-BU system rated power, provided that the ultracapacitors’ state of charge (SOC) is greater than a minimum threshold. Whenever the power required to operate the vehicle is lower than the FC-BU rated power, the ultracapacitors can be charged with the power in excess. Whenever regenerative braking operations occur, energy is put into the UC tank provided this device is not fully charged yet. The investigated propulsion system arrangement is shown in Fig. 1, where the FC is the main generation unit and BU and UCs form the combined storage unit. Under light load conditions, due to the poor efficiency of the fuel cell, the battery is used to supply the power to the load. The UC tank is used to satisfy acceleration and regenerative braking requirements accomplishing system load transients and improving on-board BU cyclic life. Additionally, it is responsible to control the dc-link

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Fig. 1.


Proposed hybrid drive-train.

voltage, while the other sources are current controlled in order to limit the current variation ratio and to prevent excessive peaks. The goal of this paper is to develop designing and sizing for the dc–dc converters in order to achieve the best compromise for FC generator and combined storage unit sizing, and system dynamic behavior; thus it will be analyzed the influence of the system components in choosing the best feedback variable for each converter and to designing the satisfactory regulators. II. MIPEC TOPOLOGY As mentioned, a multiple-input power electronic converter (MIPEC) is proposed to interface traction drive requirements with characteristics of on-board generation units and storage units. Both FC and UC typically present a lower terminal voltage than the dc voltage necessary to feed the traction inverter. Also for the BU would be of practical interest to use a lower voltage, in order to minimize the series resistance. In such cases it is necessary to use step-up converters for connecting the sources with the common dc bus. Additionally, for the BU and for the UCs it is necessary to have step-down operation in order to recharge them and to accomplish regenerative braking, what means that these converters must be bidirectional in current. A convenient topology is shown in Fig. 2. Each dc–dc converter can be built using a branch of a three-phase dc–ac converter, what means that there are power modules and drives already available in the market. Considering that the common dc-link voltage is the highest, the bottom transistor, together with the top diode, configures a boost converter, while the bottom diode and the top transistor realize the buck converter. For the FC converter the buck action must not occur because this apparatus does not take charge from the dc-link. A filter capacitor is connected at each source terminals in order to minimize the circulation of high-frequency components through the supplies. This filtering is as effective due to the presence of the sources series resistance. The converters can be voltage or current controlled, depending on the source role in the overall system, and their limitations. For example, it is important to limit the current variation in the FC, as well as in the BU. As any capacitor, the UC can be controlled in voltage mode, using a maximum current protection. The reference signals for the control loops

Fig. 2. Proposed MIPEC topology.

are derived from many parameters: the instantaneous load current, the dc-link voltage, the BU and UC state of charges, the FC output power, etc. In the following the expressions for reference signals, to be used in MIPEC control, are provided

const

(1)

where and are, respectively, the dc-link and fuel cell and are the current values measured current, of charging and discharging for ultracapacitor tank and battery unit whenever storage units’ SOC is either lower or higher of the , and are duty cycles of, reordinary admitted range, spectively, BU and FC converters and is the dc-link voltage. First two expressions give reference currents for fuel cell and battery unit converters, reference signals’ variation is controlled on the basis of generator and storage unit characteristics. The ultracapacitor converter is regulated to keep dc-link voltage either constant or at the most suitable value for traction drive mode of operation. III. DYNAMIC MODELING Dynamic modeling is necessary to evince the relationships between system transient behavior and either on-board power source or combined storage unit or traction drive parameters.

SOLERO et al.: DESIGN OF MULTIPLE-INPUT POWER CONVERTER

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C for the first topologic state and the second one. The system behavior can be obtained averaging each matrix by the duty-cycle, , in which it is valid (2) C

C (3)

Fig. 3. DC–DC converter for dynamic modeling.

Small signal modeling, considering the average value of the state variables over one switching period, is a well-known method to analyze time-varying nonlinear systems, like a switched-mode power supply [4], [5]. The resulting model is valid in a frequency range sufficiently below the switching frequency. The state equations or the equivalent transfer function can be used to design the regulators in order to achieve a desired system performance. There are many methods to achieve dynamic modeling of power converters and in this paper the one described in [6] is used. However, some modifications to the well known state variables averaging method have been included in order to take into account possible dependence of output variables from inputs’ vector. The proposed modifications allow to include parasitics of both converter power inductors and capacitors and to take in consideration inner resistance of both vehicle on-board power sources and storage units. Fig. 3 shows the single dc-dc converter considered, including mentioned parasitics of power components. If the converter works as step-up, the average value , and are positive. In the step-down mode of the currents (necessary to recharge BU and UC), the average values are negative. As the dynamic behavior as boost converter imposes more severe restrictions for the control loop design, this case is analyzed at the beginning and, afterwards, the buck operation is verified. As the power switches operate in complementary way, the converter always operates in continuous conduction mode (CCM). Notice that, in steady state, the duty-cycle depends only on the voltages and (neglecting the parasitic resistances). The average current is adjusted during the transients and does not depend on the voltages. A. State Variables Averaging Method The state variables, usually the inductors current and the capacitors voltage, are represented in the vector . The sources are represented in the vector . For the next analysis the sources are supposed of fixed value. For each topologic situation, the differential equations should be obtained and put in the format . These equations are valid during one topologic combination, for example, while the transistor is on. During the diode conduction, the equations will be . As the circuit operates in CCM, there are only these two cases. The same procedure is used to obtain the equations that describe the output variable: C , for

It is possible to split the state variables, the output and the control variable (duty-cycle) in their average value plus a perturbation

(4) Substituting (4) into (2) and (3), and neglecting the product of two perturbations, it is possible to obtain the desired transfer function and the output average value C

C

C (5) (6)

C where

C

C

C

B. Boost-Converter Let us consider the boost converter, in the CCM, having a capacitive input filter and including parasitics of power inductors and capacitors. The voltage source presents a series resistance . The load is represented by a current source that, for the dynamic analysis, is an additional input. Fig. 3 shows the circuit and Fig. 4 indicates both equivalent topologies. Taking the voltage as the output variable, the transfer function to the duty-cycle, that is the control variable, is calculated using the equations shown at the bottom of the next page. Thus, the resulting transfer function is C C

C

(7)

, and C C [7]. where, in case of ideal converter, This expression is used to analyze UC converter dynamic behavior; as it will be more clearly detailed in the next paragraph, the behavior of the resulting transfer function is different from ideal converter transfer function mainly for the presence of an additional zero; which is caused by output capacitor resistance and capacitance C and it is usually positioned at quite high frequency.

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capacitor avoids the additional low-pass filter and regulator design has less constraints. Taking the current as the output variable, no differences are found in A and B matrices since they depend on circuit modeling, whereas C, G, and E are the following: C

C (a)

thus, the resulting function to the duty cycle is different from ideal converter situation only for and it can be expressed as C (8) When the current E are the following:

(b)

C

C

Fig. 4. (a) Low-side switch conduction (either step-up switch or step-down diode). (b) High-side switch conduction (either step-up diode or step-down switch).

As FC and BU converters are controlled in current mode, it is necessary to define in which point the current should be controlled. The inductor current and the source current are the two options which have been investigated. When the inductor current is the controlled variable an additional low-pass filter in the feedback path is required for low inductance value, thus making quite difficult to design a regulator for having wide compensation band with secure phase margin. In case of source current as controlled variable, the natural filtering introduced by the input

C

C

(9) FC and BU converters’ dynamic behavior can be investigated by using either (8) or (9). However, control of the source current allows at least a reduced order filter in the feedback path; thus,

C C

C

C

C

C C

C

the resulting function to the duty cycle is different from the ideal converter case for the presence of the G matrix and because

C

C

is the output variable matrices C, G, and

C


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the resulting higher phase margin of the converter transfer function tolerates high gain regulators and dynamic of the converter is improved.

TABLE I GENERATION AND STORAGE UNITS CONFIGURATION

IV. MIPEC DESIGN Power sizing of both generation and storage units has been achieved on the basis of standard driving cycle and desired performances for a small size HEVs class. The chosen generation unit is a 18-kW FC since it is able to deliver the average power required by the most common combined urban-highway driving cycles. Efficiency of FC generator dramatically decreases when required power is less than 10–15% of the FC maximum deliverable power; therefore, the FC generator should be switched off and the BU is in charge of supplying the required power. Total energy required for the BU, considering that it is also responsible of on-board electric loads, is almost 7000 kJ; besides the BU itself should be able to deliver 10 kW for at least 10 min in order to assure 130 km/h cruising speed of the vehicle for the mentioned time. A 30 kW–260 kJ UC tank is needed as it is responsible of great part of vehicle accelerations and regenerative brakings, in fact current variations are limited for both FC generator and BU to reduce stress and to assure them a sufficient life-time. The commercial traction drive used in the proposed propulsion system is formed of a VSI-inverter and an induction motor, and it is rated 216 V–140 A at dc-link. Selection of voltage and current rated values for both generation and storage units must comply with traction drive and MIPEC topology specifications, in fact the number of elements that must be connected in series to form either the FC generator or the BU or the UC tank has minimum and maximum values related to both dc-link voltage and acceptable duty cycle values for MIPEC switches. The following expressions can be used to find out the most suitable number of elements to be connected in series (10) (11) where and are the lowest and highest acceptable element voltage, is the equivalent inner resistance for each element, and are the minimum and maximum accepted values for and are, respectively, each unit under investigation, the lowest and highest value for switch duty cycle in boost mode and of operation, whereas the same meaning is related to in buck mode of operation. Investigation on commercial products and the iterative applying of (10) and (11) for both UC tank and BU, and only (10) for FC generator, led to define generation and storage units’ configuration as shown in Table I. Final FC generator configuration is rated 18 kW–160 A, 112 V (inner voltage drop is included) at rated power—and elements number of 200 was chosen among the results satis203) in order to limit at 0.5 fying (10) (i.e., 135 in steady state condition and improve the switch utilization factor. Similar considerations on improving switch utilization 13 is the factor led to choose 12 elements for solution to (10) and (11)—that is formed of Genesis batteries

TABLE II MIPEC POWER COMPONENTS

rated 12 V, 13 Ah. Cost saving and energy specification for UC tank affected the choice of UC modules number; in fact 4 on the basis of the the best solution for (10) and (11) is switch utilization factor. However, on the basis of commercial products available in the market it would result in over-sizing the UC unit energy, thus three modules of Maxwell UCs each rated 42 V, 145 F have been considered. Traction drive dc-link voltage and current values of generation and storage units represent specifications for switching components’ selection. Voltage ripple in dc-link is the parameter used for MIPEC output capacitance sizing, however it must be ensured that selected capacitor tolerates RMS value of the output ripple current [8]. Input capacitor and inductor are responsible of input current ripple reduction: choice of 15 kHz as switching frequency led to components’ selection as shown in Table II. Parameters’ values shown in Tables I and II are used in transfer functions (7)–(9) for MIPEC dynamic investigation. It is found that output capacitors’ equivalent series resistance (ESR) introduces a high frequency additional zero in the output voltage transfer function with respect to ideal converter case as it can be noticed in Fig. 5; being the frequency position of C , the additional zero inversely related to the product

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Fig. 5.

UC converter Bode diagram: output voltage V

Fig. 6.

UC converter: RHP zero frequency versus load current.

to duty cycle.

Fig. 7.

UC converter Bode diagram: inductor i current to duty cycle.

Fig. 8. UC converter Bode diagram: power source current i to duty cycle.

significant values of the output capacitance (decrease of ESR is not linear with capacitance increase) affect regulators design since the additional zero frequency is lowered. Influence of the additional zero is significant in design of regulators for UC converter, which is supposed to react with very high dynamic to dc-link variations. Poles and zeroes values of the output voltage transfer function depend on converter operation point (average duty-cycle and average load current). One of the zeros is at the right half-plane (RHP) and its frequency decreases as the output current increases, as shown in Fig. 6. Bode diagrams of transfer functions for inductor current and input power unit current are, respectively, shown in Figs. 7 and 8. Modeling of non ideal components doesn’t affect at all the resulting transfer function is the output variable, whereas an additional zero is when present for the case. Also in this case the additional zero is C. at high frequency and it is related to the product Extensive investigation on sensitivity of non ideal components in the MIPEC model resulted that converter input

capacitance and inductance form a resonant path which affects regulators design when input power units have not negligible equivalent inner resistance (i.e., large number of series connected elements). In order to reduce the resonance, the number of series connected elements should be restricted; in particular for FC generator and BU, such a requirement is in conflict with appropriate sizing of input power units, then maximum tolerable resonance amplitude should be taken into account at definition of the series elements’ number. V. SIMULATION AND EXPERIMENTAL RESULTS Current loop with PI type regulator is chosen for both FC and BU power stage regulation in order to directly control each source current; measured currents are filtered by means of Butterworth second order continuous-time active filter to cut off switching ripple when inductor current is the output variable. UC power stage is devoted to dc link voltage control, thus a configuration with outer voltage loop plus inner current loop is proposed for the investigated application; the


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TABLE III MIPEC REGULATORS PARAMETERS

Fig. 10. Simulation results: UC converter response (v current step variation.

Fig. 9. Simulation results: UC converter response (v step variation.

and i ) at load current

current loop has come out to be indispensable to control current either fed or soaked by UC tank whenever any dc-link voltage unbalance occurs. A well-designed fast control loop can greatly improve the whole system dynamic performance during load transients, as well the propulsion system makes the best use of UCs own dynamic characteristics and high power density. To this purpose a PI type regulator is chosen for the current loop, whereas a two-zeros/three-poles regulator has been designed for the voltage loop [7]. Both current and voltage regulators have been tuned by means of Bode diagram in order to achieve a satisfactory behavior for the whole system regulator converter filter . Poles and zeros placement has produced the following expressions for respectively current regulators and voltage regulator transfer functions:

(12) where the parameters for each regulator are shown in Table III, and are, respectively, the current and the voltage error, and and are the outputs of, respectively, the current and the voltage regulator. Control loops’ design has been investigated by means of Matlab-Simulink models in which quantization of measures and control discrete transfer functions have been taken in consideration as well as both true calculus mode adopted on DSP and control loop delays have been included. Dynamic response and stability for each converter included in MIPEC configuration have been tested at different reference signals and load variations, achieved results show good dynamic performance in every simulated operating condition. In Figs. 9 and 10, UC converter response is shown for load current step

and

i

) at load

variation of 30 A, respectively, when and are used as regulated variable in current loop. Both simulations show a smooth regulation of the output voltage; however, the mode of operation of a Butterworth second order continuous-time active filter is considered when inductor current is controlled, whereas no filtering of the controlled current is required to control the power source current. A fixed-point 16-b DSP from Analog Devices has been used in order to implement the MIPEC whole control system. MIPEC switching frequency is chosen to be 15 kHz according to hardware components specifications; besides, a fixed length of 133 s is chosen as maximum period required for the whole control algorithm to be completely executed. As a consequence, the maximum achievable sampling frequency of 7.5 kHz has been chosen to implement regulators’ transfer functions in discrete form for both BU and FC current loops; whereas the frequency of 1.875 kHz has been used for the UC double loop. In fact, a four times reduced frequency improves stiffness of the UC control system by reducing the effects of the voltage loop RHP-zero. DSP standard fixed point configuration was adopted for the whole algorithm except for UC regulator implemented by using the emulated floating point mode of operation. Control algorithm includes time-variation limiting of currents should be supplied by either FC generator or BU to achieve a safe dynamic mode of operation for both the power source and the main energy storage. UC power stage currents have no time-variation limitation in order to achieve the most effective output voltage regulation. DSP is also responsible for system protection actions (i.e., overcurrents, overvoltages and overtemperature). Experimental measurement revealed the DSP takes 110 s to complete the control algorithm execution, this value well fits the previously chosen maximum available time. Figs. 11 and 12 show, respectively, the block diagram of the whole system and the block diagram of the UC power stage control in which are schematically depicted the regulators, filters, AD converters and PWM generators. Block diagrams of both FC and BU power stage control are similar to the one shown in Fig. 12 where the voltage regulator should not be included. For experimental testing activity purpose, FC generator has been simulated by means of a 20 kW regulated dc power

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Fig. 11.


MIPEC control block diagram.

Fig. 13. Experimental results: UC converter response (v current step variation.

Fig. 12.

and

i

) at load

Block diagram of the UC power stage control.

source, UC tank was accomplished with three BMOD0115A09 Maxwell modules (42 V, 145 F) in series connections, series connection of 12 modules from Genesis (12 V, 13 Ah) form the battery unit. Dynamic response for each power stage converter has been tested; transients of UC power stage at step variation of voltage reference have been investigated in detail at no-load operation, which is the worst case for output voltage control as MIPEC output capacitors provide a very low damping effect. In Fig. 13, the UC converter experimental response is shown for load current step variation of about 25 A when is used as regulated variable in current loop. The comparison with time response and output voltage regulation of Fig. 9 validates the proposed theoretical approach for designing the regulators for both current loop and voltage loop; however, non idealities in cables, connections, and components of the experimental set-up slightly change current and voltage transients, therefore, the experimental testing shows a more damped behavior of the system than the achieved simulations.

Fig. 14. Experimental results: MIPEC load transient operation at 0–50 A load current step variation.

Load testing has been carried out by operating all MIPEC power stages at the same time. Figs. 14 and 15 show MIPEC dynamic performance at load operation corresponding to resistive load step variation respectively from no-load to 4.33 (i.e., 50 A at 216 V) and from 4.33 to no-load; current variation versus time limitation is implemented for both FC generator and BU storage, UC converter acts in order to compensate the output voltage variation. In case of quite low SOC for storage units, as transient is completed, BU and UC would require to be charged (at constant current) from FC generator. Complete propulsion system has been loaded with several different driving cycles and tested at ENEA lab facilities. Fig. 16 shows the current waveforms for each input power source and dc-link when almost 50 Nm torque step is applied to the traction


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driving cycle, whose accelerations have been 40% increased in order to achieve a more realistic testing. It can be noticed that UCs run during short and severe both accelerations and brakings, whereas gentle speed variations are accomplished by FC generator.

VI. CONCLUSION

Fig. 15. Experimental results: MIPEC load transient operation at 50-0 A load current step variation.

The topology for a three-inputs/one-output power converter (MIPEC) devoted to HEVs applications has been presented. On-board generation and storage units’ sizing as well MIPEC design and prototypal realization have been carried on according to specifications from traction drive and vehicle performances. Dynamic modeling of the proposed converter has been achieved to evince dependence of system transient behavior from parameters of on-board generator, storage units and traction drive. Simulations confirmed performances of designed regulators and control strategy. A 60-kW MIPEC prototype has been used for the hybrid propulsion system where a 18-kW fuel cell is the main generation unit and batteries and ultracapacitors form the combined storage unit. Experimental testing of the whole system has been accomplished at a suitable HEV test-bed where applied traction drive torque transients and several driving cycles proved MIPEC good dynamic behavior.

ACKNOWLEDGMENT The authors wish to thank A. Puccetti for supporting the experimental activities at ENEA laboratory facilities.

Fig. 16. Experimental results: propulsion system transient operation at traction motor torque step.

Fig. 17. Experimental results: propulsion system testing at modified urban ECE-15 driving cycle.

motor; whereas Fig. 17 depicts same current waveforms when the complete propulsion system is loaded with urban ECE-15

REFERENCES [1] V. Raman, “The hydrogen fuel option for fuel cell vehicle fleets,” Fuel Cell Power Transport., pp. SAE SP–1425, 1999. [2] K. Dircks, “Recent advances in fuel cells for transportation applications,” Fuel Cell Power Transport., pp. SAE SP–1425, 1999. [3] A. Rufer and P. Barrade, “Key developments for supercapacitive energy storage: Power electronic converters, systems, and control,” in Proc. 2nd Boostcap Meeting. Fribourg, Switzerland, 2005. [4] A. Di Napoli, F. Crescimbini, F. G. Capponi, and L. Solero, “Control strategy for multiple input dc–dc power converters devoted to hybrid vehicle propulsion system,” in Proc. IEEE ISIE’02, L’Aquila, Italy, Jul. 2002, pp. 1036–1041. [5] R. D. Middlebrook and S. Cuk, “A general unified approach to modeling switching converter power stage,” in Proc. IEEE PESC, 1976, pp. 18–34. [6] P. T. Krein, J. Bentsman, R. M. Bass, and B. L. Lesieutre, “On the use of averaging for the analysis of power electronic systems,” IEEE Trans. Power Electron., vol. 5, no. 2, pp. 182–190, Apr. 1990. [7] J. A. Pomilio, L. Solero, F. Crescimbini, and A. Di Napoli, “Dynamic modeling and regulators design for multiple input power converters for the propulsion system of electric vehicles,” in Proc. COBEP’03, Fortaleza, Brazil, Sep. 2003, pp. 362–367. [8] A. Di Napoli, F. Crescimbini, S. Rodo, and L. Solero, “Multiple input DC-DC converter for fuel-cell powered hybrid vehicles,” in Proc. IEEE PESC’02, Cairns, Australia, Jun. 2002.

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Luca Solero (M’98) received the M.S. degree in electrical engineering from the University of Rome, Rome, Italy, in 1994. Since 1996, he has been with the Department of Mechanical and Industrial Engineering, University of “Roma Tre,” where he is currently an Assistant Professor. During 2002, he was a Visiting Scholar at the Center for Power Electronics Systems (CPES), Virginia Polytechnic Institute and University, Blacksburg. He has coauthored more than 60 published technical papers and has been involved in several government and industry-sponsored projects in the fields of power electronics and electrical drives. His research interests include power converter topologies, permanent magnet motor drive and control systems design for unconventional applications such as electric and hybrid vehicle, and renewable energy systems. Mr. Solero is a member of the IEEE Industry Applications, IEEE Power Electronics, and IEEE Industrial Electronics Societies.

Alessandro Lidozzi (S’04) received the M.S. degree in electronic engineering from the University of Rome, Rome, Italy, in 2003, where he is currently pursuing the Ph.D. degree. His research interests are mainly focused in multiconverter based applications, dc–dc power converters modeling and control, and nonlinear control of permanent magnet motor drives. Mr. Lidozzi received the Student Award and a Travel Grant from the International Symposium on Industrial Electronics (ISIE) in 2004. He is member of the IEEE Industrial Electronics Society.

José Antenor Pomilio (M’92–SM’02) was born in Jundiaí, Brazil, in 1960. He received the B.S., M.S., and Ph.D. degrees in electrical engineering from the University of Campinas, Brazil, in 1983, 1986, and 1991, respectively. From 1988 to 1991, he was head of the Power Electronics Group, Brazilian Synchrotron Laboratory. Currently, he is a Professor at the School of Electrical and Computer Engineering, University of Campinas, where he has been since 1984. In 1993 and 2003, he was Visiting Professor at the University of Padova, Padova, Italy, and at the Third University of Rome, Rome, Italy, respectively. His main interests are switching-mode power supplies, power factor correction, electrical drives, and active power filters. Dr. Pomilio is an Associate Editor of the IEEE TRANSACTIONS ON POWER ELECTRONICS.