Four-band frequency selective surface with double ... - IEEE Xplore

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 42, NO. 12, DECEMBER 1994

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Four-Band Frequency Selective Surface with Double-Square-Loop Patch Elements Te-Kao Wu

Abstract-Design and experimental verifications are presented for a four-band frequency selective surface (FSS) with perfectly conducting double-square-loop (DSL) patch elements. A single screen DSL element FSS is demonstrated for 1) a four-band FSS which reflects the X- and Ka-band signals while passing the S- and Ku-band signals, and 2) a lowpass (or Ka-add-on) FSS that reflects the Ka-band signal while passing the S-, X-, and Ku-band signals. Cascading the above low-pass FSS with a previously published single screen tri-band FSS, a double screen FSS is also successfully demonstrated for the Cassini four-band application. The good agreement obtained between the measured and the computed results verified both the single and double screen four-frequency FSS approaehes for the Cassini Project.

I. INTRODUCTION Frequency selective surfaces (FSS) have often been considered for reflector antenna applications [ 11-[8]. Recently, the NASA Cassini Project [5] required the use of multiple microwave frequencies at S-, X-, Ku-, and Ka-band for science investigations and data communication links. A single HGA with an FSS subreflector, as illustrated in Fig. 1, was proposed. This arrangement allows a Cassegrain configuration at X (7.2 and 8.4 GHz) and Ka (32 and 34.5 GHz) bands and a prime focus configuration at S (2.3 GHz) and Ku (13.8 GHz) bands [5]-[8]. Circular polarizations are required for all frequency bands except the Ku-band. The FSS performance should also remain unchanged as the incident angle is steered from normal to 45". In the past, the cross-dipole patch element FSS was used for the subreflector design in the reflector antennas of Voyager [ 13 for reflecting the X-band waves and passing the S-band waves, and the Tracking and Data Relay Satellite System (TDRSS) for diplexing the S- and Ku-band waves [2]. The characteristics of the cross-dipole element FSS change drastically as the incident angle is steered from normal to 40". Thus a large band separation is required to minimize the RF losses for these dual band applications. This is evidenced by the reflection and transmission hand ratio ( f r / f t ) being 7:l for a single screen FSS [2] or 4.1 for a double screen FSS [l] with cross-dipole patch elements. Much closer band spacings and a stable bandwidth insensitive to the incident angle variation are available from dipoles sandwiched between dielectrics about X/2 thick [9]. However, if a lighter weight structure or the circular polarization is required then thin dielectrics supported by low-mass material must be used, and other elements have to be sought. FSS's with double square loop (DSL) and double ring patch elements have been designed for frequency band ratios (fr/ft) from 1.5 to 2 [3]-[8], [lo], [ll].Their resonant frequencies are fairly stable with respect to changes in the incident angle and polarizations. In addition, the grid geometry is symmetrical in the z and y directions. This implies that it is also good for circular polarizations. Hence, the DSL and double ring elements are considered for the Cassini FSS design to achieve 1) the multiplexing of four frequency bands, 2) smaller frequency-band separations ( f t / f r less than 1.7), and 3) less sensitivity to the incident angle variation and polarizations. Since the Manuscript received December 15, 1993; revised July 5, 1994.

The author is with the Jet Propulsion Laboratory, California Institute of Technology, Pasadena, CA 91 109 USA. IEEE, Log Number 9407154.

Fig. 1.

Proposed Cassini high gain antenna with a four-frequency FSS.

double ring element FSS was discussed in [6]-[8], this paper will concentrate on the DSL element FSS. In another parallel effort using the double ring patch element [8], it was found that only the tri-band FSS application can be designed with the single screen approach. This implies that the DSL element is superior to the double ring element for multiband (24-band) FSS applications. To meet the Cassini antenna subsystem's RF requirements, two design approaches as shown in Fig. 2, are proposed. In addition, the FSS screen was bonded on to a Kevlar honeycomb panel to meet the mechanical and thermal environmental requirements. The first approach, implementing the two-screen design, uses two FSS grids. The front FSS grid is called the Ka-add-on FSS. It reflects Ka-band waves but passes S-, X-, and Ku-band waves. The back FSS grid is called the three-frequency (or tri-band) FSS. It reflects X-band waves but passes S- and Ku-band waves. The resultant FSS reflects both X- and Ka-band waves but passes both S- and Ku-band waves. The second approach, implementing the single screen design, uses only a single FSS grid to reflect the X- and Ka-band waves and to pass the S- and Ku-band waves. The analysis of a DSL FSS has been well documented in [ 121-[ 171. In this paper all the DSL FSS's were designed using the accurate integral equation formulation (IEF) [13]-[17]. In the following sections, the design and performance of multiband FSS's with DSL elements will be demonstrated.

11. SINGLESCREENFOUR-FREQUENCY FSS The design and performance of a tri-band FSS with DSL elements can be found in [4]. In that paper, the DSL FSS was designed for one reflection band (or only the first resonance is stable enough for a large variation of incident angle application). However, with a careful design, the DSL FSS should provide two resonances (or two reflection bands), i.e., one at a lower frequency (caused by the larger loop) and the other at a higher frequency (caused by the smaller loop). Therefore, one may be able to design a single screen DSL element FSS for the Cassini's four-frequency FSS. In other words, only one DSL element FSS grid is needed for reflecting the X- and Ka-band while passing the S- and Ku-band waves. To avoid grating lobes, the four-frequency integrated FSS was etched on a 0.01" thick Duroid 6010.5 substrate. The substrate has a dielectric constant of 11 and the loss tangent is 0.0028. The geometry and configuration of the four-frequency FSS with a Kevlar honeycomb are given in Fig. 3. Fig. 3 also shows the computed and measured transmission data of this DSL element FSS for the incident angle steered from normal to 45". The resonant

0018-926X/94$04.000 1994 IEEE


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Geometryv O r Arrays (Double S quare LOOP)

SIDE VIEW

TOP VIEW

Fig. 2. Cassini four-frequency DSL FSS design approaches. TABLE I

COMPUTED LOSS (dB)

OF THE

SINGLE SCREEN FOUR-FREQUENCY FSS

Frequency 0,= Oo GHz

2.3 7.2 8.4 13.8 32 34

I

B l

Fig. 3. Computed and measured transmission characteristics of a single screen four-frequency DSL FSS with the Kevlar honeycomb. frequencies are very stable with respect to the incident angle variation and are near the design frequencies, i.e., 8.45 GHz and 33 GHz. Only representative measured data at normal incidence are shown here. Good agreement between measured and predicted results are observed for incident angles at 0, 30, and 45", and for both TE and TM polarizations. This verified the single screen four-frequency FSS design approach. Table I summarizes the computed RF loss performance of this DSL element FSS. Note that the losses at 2.3 and 13.8 GHz are the transmission losses while the losses at other frequencies are reflection losses. The single screen FSS has the advantages of lower mass, smaller volume, and easier fabrication than the double screen approach,

0.95 0.45 0.08 0.37 0.09

0.14

30'

TE 1.2 0.37 0.07 0.56 0.17 0.2

45O TM

TE

TM

0.73

1.6 0.27 0.06 0.9 0.16 0.13

0.5 0.9 0.16 0.2 0.69

0.61 0.11

0.29 0.13 0.21

0.43

since neither accurate alignment nor a dielectric spacer with uniform thickness and dielectric properties is required. However, to insure that this FSS operates at all the frequency bands (i.e., from 2 to 35 GHz), high dielectric constant (er 2 11) substrate is required for eliminating the grating lobe at 32 and 34 GHz. Currently, none of the high dielectric constant materials (e.g., the Duroid 6010.5 laminate) has been qualified for space applications. Therefore, the following double screen DSL FSS is considered for the Cassini four-frequency FSS. 111. DOUBLESCREENFOUR-FREQUENCY FSS

As mentioned in Section I, one can add a low-pass (or Ka-addon) FSS in front of the tri-band FSS to form the Cassini double screen four-frequency FSS. In operation, the Ka- and X-band waves are reflected by the front (top) and back (bottom) grids, respectively. Both S- and Ku-band waves will pass through this dual-screen FSS with minimum RF insertion loss. The three-frequency FSS has been described in [4]. Hence in this section the design and performance of a single screen Ka-add-on FSS will be discussed first, then the

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Fig. 4. Comparison of the computed and measured transmission characteristics of a single screen low-pass DSL FSS on a thin Kapton at normal incidence. Frequency

cascading of the Ka-add-on and three-frequency FSS's (or the double screen four-frequency FSS) will be described next. Fig. 4 gives the geometry of a single screen Ka-add-on DSL FSS etched on a 1 mil thick and 20" by 20" sized Kapton substrate. Fig. 4 shows representatively the good agreement between the measured and the calculated transmission results at normal incidence. It also shows that this Ka-add-on FSS is designed to reflect the Ka-band wave and to transmit the lower frequency waves at S-, X-, and Kuband. Thus it is also called a low-pass FSS. Furthermore, another Ka-add-on FSS may be designed with the presence of a Kevlar honeycomb of Fig. 2(b) with the element dimensions given in Fig. 5. The predicted and measured transmission performance is plotted in Fig. 5 for the incident angle steered from normal to 45" and for both TE and TM polarizations. Fig. 5 also illustrates the comparison between the computed and measured data for this FSS's transmission performance. Again the good agreement between the measured and calculated results validates this single screen low-pass DSL FSS approach. The cascading of two nonsimilar FSS screens is very difficult to analyze exactly. However, using the following systematic procedures, a single-mode cascading approach may readily be employed to get a first order assessment of this double-screen FSS's performance. Consider the double-screen FSS as shown in Fig. 2(b). First, one divides the FSS at the mid-plane of the Kevlar honeycomb. Upwards from this middle plane is considered the first FSS screen section, and below this plane is the second FSS screen section. Each FSS section can be accurately modeled by the single screen FSS analysis described earlier. Since the dividing plane is electrically far from the FSS grid and the FSS element spacing is less than a half wavelength, one can assume that only the 0th order mode is significant. Thus one may next cascade the two sections by converting the scattering matrix [SI from each section to a transmission matrix [t] and multiplying the resulting [t] matrices. The conversion from [SI matrix to [t] matrix is the same as that described in [18], [19]. The final [t] matrix product is then converted back to a scattering matrix, which yields the transmission and reflection coefficients for the double screen FSS. To fabricate a double screen four-frequency FSS, the above mentioned thin screen Ka-add-on FSS (Fig. 4) and the thin screen three-frequency FSS (Fig. 1 of [4]) were assembled together with a low density foam spacer. The foam spacer is a 0.75" thick Rohacell 51-IG foam. Fig. 6 presents representatively the good

(CHn)

Fig. 5. Computed and measured transmission performance of a single screen low-pass DSL FSS with the Kevlar honeycomb.

0

-10

3

H

-20

3 L-

-30

NORMAL INCIDENCE -X- COMPUTED

-+0

EXPERIMENTAL I

I

I

10

20

30

*

40

F'RMUENCY (GHr)

Fig. 6. Comparison of computed and measured transmission performance of a double screen four-frequency DSL FSS with a low density foam spacer.

agreement obtained between the predicted and measured transmission performances of this double screen FSS at normal incidence. This verified the accuracy of this efficient method and the double screen four-frequency FSS design approach. The computed loss performance at the four bands (i.e., S/X/KU/KA bands) is summarized in Table 11. Note that the losses at 2.3 and 13.8 GHz are the transmission losses while the losses at 7.2,8.4,32, and 34 GHz are reflection losses. Since the closed cell foam material is not acceptable in space applications, a low-mass open cell Kevlar honeycomb sandwich is considered next for the double screen four-frequency FSS design approach. The above mentioned Ka-add-on DSL FSS with a Kevlar honeycomb (Fig. 5 ) was then assembled with the three-frequency DSL

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 42,

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FREQUENCY (GHz)

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30 31 32

Fig. 7. Comparison of computed and measured transmission performance of a double screen four- frequency DSL FSS with a Kevlar honeycomb. TABLE II COMPUTED Loss (dB)OF THE DOUBLE SCREEN FOUR-FREQUENCY FSS WITH FOAMSPACER Frequency 0;= 0’ GHz 2.3 7.2 8.4 13.8 32 34

0.42 0.24 0.04 0.46 0.33 0.02

4oo

30’

TE 0.48 0.28 0.01 0.23 0.2 0.04

TM

TE

TM

0.33 0.44 0.01 0.17 0.3 0.03

0.55 0.33 0.02 0.23 0.06 0.17

0.28 0.73 0.03 0.15 0.27 0.18

TABLE III COMPUTED Loss (dB) OF THE DOUBLESCREEN FOUR-FRFQUENCY FSS m HONEYCOMB Frequency 0;= Oo GHz

TE

TM

TE

TM

2.3 7.2 8.4 13.8 32 34

0.5 0.73 0.17 1.2 0.19 0.28

0.33 1.1 0.19 0.73 0.22 0.33

0.68 0.85 0.22 2.1 0.21 0.2

0.23 1.95 0.29 0.53 0.48 0.3

0.4 1 0.65 0.14 1.1 0.53 0.21

30’

45O

FSS (Fig. 6 of [4]) to become a double screen four-frequency FSS as shown in Fig. 2(b). A representative comparison between the measured and computed transmission performances of this double screen FSS is shown in Fig. 7 at normal incidence. Note that there is no S band measured data, since the measurement requires much larger size than the present 2 0 by 20“ at this particular frequency. Nevertheless, the accuracy of the DSL FSS design software has been checked thoroughly by measurements at higher frequencies. In addition, the accuracy of the DSL FSS design software has been verified at S-band by a waveguide simulator measurement [20]. Thus the computed results at S-band are considered to be accurate. The computed RF loss performance for this double screen FSS is summarized in Table 111. The losses are higher than the ones (Table 11) of the double screen DSL FSS with the low density foam. This is due to the relatively higher loss tangent of the Kevlar/Epoxy skin materials. IV. CONCLUSION Design and measurement results are presented for four-band FSS’s with DSL patch elements. Good agreement between the measured and computed results verifies the various design approaches described in this paper. For applications in the NASA Cassini Project, both the single and double screen DSL element FSS have been successfully demonstrated for the four-band FSS. In another parallel effort using

the double ring patch element [8], it was found that only the hi-bund application can be designed with the single screen approach. This implies that the DSL element is superior to the double ring element for multiband (3.4-band) FSS applications. Because the Duroid 6010.5 substrate has not yet been qualified for space applications, the double screen DSL FSS with a Kevlar honeycomb was also developed for the Cassini four-frequency FSS. ACKNOWLEDGMENT The work described in this paper was carried out by the Jet Propulsion Laboratoq, California Institute of Technology, under contract with the National Aeronautics and Space Administration. The author wishes to thank Dr. J. Vacchione for providing the double screen FSS results of Fig. 16, Dr. K. Woo for managerial support, Mr. G. Hicky for fabricating the honeycomb sandwiches, and Mr. C. Chavez for performing the FSS measurements.

REFERENCES [I] G. H. Schennum, “Frequency-selective surfaces for multiple frequency antennas,” Microwave J., vol. 16, pp. 55-57, May 1973. [2] V. D. Agrawal and W. A. Imbriale, “Design of a dichroic Cassegrah subreflector,” IEEE Trans. Antennas Propugat., vol. AP-27, no. 4, pp. 466-473, July 1979. [3] K. Ueno et al., “Characteristicsof FSS for a multi-band communication satellite,” in 1991 Inf. IEEEAP-S Symp. Dig., Ont., Canada, June 1991, pp. 735-738. [4] T. K. Wu, “Single screen triband FSS with double-square-loop elements,” Microwave and Opt. Tech. Lett., vol. 5, no. 2, pp. 56-59, Feb. 1992. [5] -, “Double-square-loop FSS for multiplexing four (S/X/Ku/Ka) bands,” in I991 Int. IEEE AP-S Symp. Dig.,Ont., Canada, June 1991, pp. 1885-1888. [6] J. Huang, T. K. Wu, and S. W. Lee, ‘Tri-band FSS with circular ring elements,” 1991 Int. IEEE Trans. Antenneas Propagar., vol. 42, no. 2, pp. 166-175, Feb. 1994. [7] T. K. Wu et al., “Multi-ring element FSS for multi-band applications,” in 1992 Inf. IEEEAP-SSymp. Dig.,Chicago, IL,July 1992, pp. 1775-1778; also accepted for publication in IEEE Trans. Anfennas Propagar.. [8] T. K . Wu, M. Zimmerman, and S. W. Lee, “Evaluation of frequency selective reflector antenna systems,” Microwave and Opt. Technol. Left, vol. 6, no. 3, pp. 175-179, Mar. 1993. [9] P. Callaghan, E. Parker, and R. Langley, “Influence of supporting dielectric layers on the transmission properties of frequency selective surfaces,’’IEE Proc.-H, vol. 138, no. 5, pp. 448-454, Oct. 1991. [IO] R. J. Langley and E. A. Parker, “Double-square frequency-selective surfaces and their equivalent circuit,” Elecfron.Lett., vol. 19, no. 17, p. 675, Aug. 18, 1983. [Ill C. K. Lee and R. Langley, “Equivalent circuit models for frequency selective surfaces at oblique angle of incidence,” IEE Proc., vol. 132, pt. H, no. 6, pp. 395-398, Oct. 1985. [I21 I. Anderson, “On the theory of self-resonant grids,’’ Bell Syst. Tech. J., vol. 54, pp. 1725-1731, Dec. 1975.

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T. K. Wu, “Quasi-optical grids with rectangular patcWaperture elements,” Int. J. Infrared and MMW, vol. 14, no. 5, pp. 1017-1033, May,

1993. R. Mittra, C. Chan, and T. Cwik, “Techniques for analyzing frequency selective surfaces-A review,” IEEE Proc., vol. 76, no. 23, pp. 1593-1615, Dec. 1988. B. J. Rubin and H. L. Bertoni, “Reflection from a periodically perforated plane using a subsectional current approximation,” IEEE Trans. Antennas Propagat., vol. AP-31, no. 6, pp. 829-836, Nov. 1983. S . W. Lee, “Scattering by dielectric-loaded screen,” IEEE Trans.Antennas Propagat., vol. AP-19, no. 5, p. 656, Sept. 1971. C. C. Chen, “Diffraction of electromagnetic waves by a conducting screen perforated with circular holes,” IEEE Trans. Microwave Theory Tech., vol. MlT-19, pp. 475431, May 1971. J. Vacchione and T. K. Wu, “Analysis of a dual, non-similar, screen FSS using simple cascading procedures,” in 1992 Int. IEEE AP-S Symp. Dig., Chicago, IL,July 1992, pp. 1779-1782. S . Ramo et al., Fields and Waves in Communication Electronics. New York Wiley, 1965, p. 605. T. K. Wu, “Cassini frequency selective surface development,” J. Electromagnetic Waves and Appl., vol. 8, no. 12, Dec. 1994.

+=nn

+=0

YP +=nn

Fig. 1. Geometry for the three-dimensional scattering at an impedance wedge.

Current Evaluation on the Faces of an Impedance Wedge Illuminated by an Electromagnetic Pulse G. Pelosi, G. Manara, A. Freni, and J. M. L. Bernard Abstract-The scattering of an electromagnetic time-dependent plane wave by the edge of an impedance wedge is analyzed. Suitable expressions are presented for the surface currents which are induced on the two faces of the wedge. Numerical results are shown for different electrical and geometrical configurations and compared with data available in the literature for the case of a perfectly conducting wedge.

I.

INTRODUCTION

Time-domain field solutions are of great importance for a large variety of practical applications. They can be very useful in evaluating the effects of the exposition of a complex structure to impulsive electromagnetic excitations, as for example in the cases of the lightning flash or the nuclear pulse. In particular, several timedomain analytical solutions have been derived for the canonical problem of the electromagnetic scattering from perfectly [ 11-[6] and nonperfectly conducting wedges [7]-[ 131. Recently, approximate (early-time) expressions for the time-domain response of a perfectly conducting wedge [6] and an impedance wedge [13] have been presented, that are derived from the diffraction coefficients of the Uniform Geometrical Theory of Diffraction (UTD) [ 141. However, it is important to observe that the frequency-domain solution for the total field in a wedge-shaped region can be in general expressed in terms of a Sommerfeld integral. It follows that, when the boundary conditions on the two faces of the wedge do not depend on frequency, the response of the wedge to an incident impulsive Manuscript received March 7, 1994; revised August 8, 1994. G. Pelosi and A. Freni are with the Microwave Laboratory, Department of Electrical Engineering, University of Florence, 1-50134 Florence, Italy. G. Manara is with the Department of Information Engineering, University of Pisa, 1-56126 Pisa, Italy. J. M. L. Bernard is with the Le Centre Thomson #Applications Radars (LCTAR),F-78143 Velizy-Villacoublay Cedex, France. IEEE Log Number 9407153.

plane wave can be determined in a closed form by applying the basic procedure proposed by Cagniard and de Hoop [ 151, [161, successively modified in [17] by introducing the complex angle representation (see also [4, Sects. 6.3c, 6.5h, and 1.6bl). More general expressions for the fields excited by an incident plane wave with arbitrary timedependence can be recovered from this rigorous impulsive response by evaluating a simple convolution integral. They allow efficient calculations of fields and currents. Suitable and rigorous time-domain expressions for the field are. presented and discussed in Section 11, that are valid at any time and in particular also beyond the early-time period of the response of the wedge. Due to their importance for engineering applications, explicit formulas for the transient surface currents distributions which are induced on the two faces of the wedge are provided. Finally, some numerical results are shown in Section III.

II. FORMULATION The geometry for the three-dimensional scattering by an impedance wedge is depicted in Fig. 1. Two different uniform isotropic impedance boundary conditions are imposed on the two faces of the wedge. A time-dependent plane wave impinges on the edge from a direction which is determined by the two angles p and 4’. The angle p is a measure of the incidence direction skewness with respect to the edge of the wedge and p = 7r/2 corresponds to normal incidence. The observation point is at P and the exterior angle of the wedge is n7r. The geometric and electric properties of the wedge are supposed to be independent of z . The surface impedances of the q5 = 0 and q5 = n7r faces are complex constants which are not supposed to depend on frequency; they will be denoted by 20 and Z,, respectively. To simplify the discussion, their values are chosen so that the two faces of the wedge can not support surface wave propagation. The time domain response of the wedge can be obtained from the correspondent time-harmonic expression, by directly applying an inverse Laplace transform. In particular, by denoting with ua (t) and u ( t ) the z-components of the incident and total magnetic (electric)

0018-926)(/94$04.00 0 1994 IEEE