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Half-Bridge Integrated ZVS Full-Bridge Converter With Reduced Conduction Loss for Electric Vehicle Battery Chargers Il-Oun Lee, Member, IEEE, and Gun-Woo Moon, Member, IEEE

Abstract—A half-bridge integrated zero-voltage-switching (ZVS) full-bridge converter with reduced conduction loss for battery on-board chargers in electric vehicles (EVs) or plug-in hybrid electric vehicles (PHEVs) is proposed in this paper. The proposed converter features a reduction in primary-conduction loss and a lower secondary-voltage stress. In addition, the proposed converter has the most favorable characteristics as battery chargers as follows: a full ZVS capability and a significantly reduced output filter size due to the improved output waveform. In this paper, the circuit configuration, operation principle, and relevant analysis results of the proposed converter are described, followed by the experimental results on a prototype converter realized with a scale-downed 2-kW battery charger for EVs or PHEVs. The experimental results validate the theoretical analysis and show the effectiveness of the proposed converter as battery on-board chargers for EVs or PHEVs. Index Terms—Battery charger, electric vehicle (EV), full-bridge (FB) converter, hybrid electric vehicle (HEV), zero-voltage switching (ZVS).

I. I NTRODUCTION

W

ITH accelerated global warming, decreasing natural resources, increasing fuel price, and economical issues, vehicles with electric propulsion, such as hybrid electric vehicles (HEVs), plug-in HEVs (PHEVs), battery electric vehicles (BEVs or EVs), and fuel cell electric vehicles, are gradually growing. These vehicles need commonly rechargeable batteries as the energy source of electric traction system [1]. Among them, PHEVs or EVs require a higher capacity and larger sized battery pack compared with other vehicles because the battery is a main energy source in PHEVs or EVs [2], [3]. The high-energy-density battery pack in PHEVs or EVs is typically recharged from the ac utility grid via an ac–dc converter named as battery charger. For low harmonic contents on the ac

Manuscript received April 8, 2013; revised July 5, 2013; accepted August 18, 2013. Date of publication September 18, 2013; date of current version February 7, 2014. This work was supported by the National Research Foundation of Korea under Grant 2012-0000981 funded by the Ministry of Education, Science and Technology of the Korean government. I.-O. Lee is with the Power Advanced Development Group, Samsung Electro-Mechanics Company, Ltd., Suwon 443-743, Korea. G.-W. Moon is with the Department of Electrical Engineering, Korea Advanced Institute of Science and Technology, Daejeon 305-701, Korea (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TIE.2013.2282608

utility grid and high efficiency, most of battery chargers have generally the basic form of an ac–dc converter with a power factor corrector (PFC), followed by an isolated dc–dc converter [4]–[9]. There are key requirements in the development of EV battery chargers. First, it is imperative to reduce their size and weight in order to facilitate packaging and to highlight the utilization factor of energy. Namely, the design for higher power density and lower weight is required. Furthermore, the conversion efficiency should be maximized during whole output conditions or battery recharging process to maximize the fuel saving and emission reduction. In order to achieve these requirements, it is necessary to adopt higher switching frequencies and softswitching technologies since a higher switching frequency is the key to reducing the size and weight of passive components used in high-power applications, and soft-switching technologies significantly lower the generated switching losses. In addition, in the PFC stage, a bridgeless design should be carried out because excessive conduction loss is generated due to the forward voltage drop for each of the bridge diodes, particularly at a lower line input voltage, which decreases the overall efficiency and greatly increases the size and weight of heat sink [10], [11]. In order to obtain higher conversion efficiency, particularly at high power levels, interleaving or parallel approaches can be considered because they significantly reduce the generated conduction loss [12]–[14]. In addition, in the case where the output voltage requirement of the battery charger is high, the rectifier diodes in the dc–dc converter could experience a serious voltage oscillation and spike. Then, lossy snubber circuitry and higher voltage-rated diodes must be required, which cause the increase in power loss, size, and weight. Thus, in designing the rectifier stage in the dc–dc converter, the design that can avoid the aforementioned problem should be also taken into account [15]. In addition, in order to maintain high efficiency under lowpower conditions, it is necessary to minimize the amount of circulating energy in the dc–dc converter. In this paper, a zero-voltage-switching (ZVS) full-bridge (FB) converter with reduced conduction loss for battery onboard chargers in PHEVs or EVs is proposed. The proposed converter consists of an FB converter integrated with a symmetric half-bridge (HB) converter in parallel. This architecture makes it possible for the proposed converter to feature a reduction in primary-conduction loss, which comes from the reduction of circulating current and the much smaller turns ratio of FB transformer. The proposed converter also has a

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Fig. 2. Rectifier output voltage waveform of the converter in [20].

Fig. 1. ZVS FB dc–dc converter with wide ZVS range and low-secondaryvoltage stress for EV battery chargers.

lower secondary-voltage stress because of the hybrid structure, which can result in a reduction of secondary-conduction loss. In addition, in the proposed converter, a full ZVS capability without adding any heavy extra inductors and a significantly reduced output filter size due to the improved output waveform are obtained. Thus, the converter’s weight or size can be greatly reduced, which is necessarily required in EV battery charger applications. The circuit configuration, operation principle, and relevant analysis results of the proposed converter are described in this paper. The experimental results on a prototype converter realized with a scale-downed 2-kW battery charger for EVs or PHEVs validate the theoretical analysis and show the effectiveness of the proposed converter as EV battery chargers. II. R EVIEW OF DC–DC C ONVERTERS FOR BATTERY C HARGERS The most preferred dc–dc topology for battery on-board chargers in EVs or PHEVs is the ZVS FB converter operated at a constant frequency under pulsewidth control strategy [16]–[18]. The converter features minimal voltage and current stresses in the power devices, a natural ZVS capability, a low current ripple in the output current, and a simple structure. Since the battery voltage is very high and varies over a wide range during battery recharging process, however, the converter suffers from excessive conduction loss due to the large circulating current and large RMS current stress in the primary side and the very high voltage stress in the secondary side. In addition, ZVS operation is not achieved over whole output conditions. For these reasons, it is widely known that it is difficult to maximize the conversion efficiency as battery chargers. In addition, the converter needs a very large output filter inductor for sufficiently reducing the current ripple of battery recharging current. To increase the conversion efficiency, the ZVS FB converter in Fig. 1 can be considered for battery chargers. Since the

Fig. 3. Battery recharging strategy consisting of constant-current and constant-voltage modes.

resonant inductor LR added in the lagging-leg switches (Q3 and Q4 ) extends ZVS range and the additional diodes of Da1 and Da2 significantly reduce the voltage stress of the rectifier stage, the conduction loss in the rectifier stage and the switching loss are greatly reduced. However, there still exist other drawbacks such as the large circulating current and large RMS current stress in the primary side and the requirement of a large output filter inductor. In order to further improve the performance of EV battery chargers, new ZVS FB converters are presented in [6] and [7] and ZVS FB converters integrated with HB converter have been recently proposed in [19]–[21]. The converters commonly feature a wide ZVS range, a reduced circulating current, a reduced voltage stress in rectifier stage, and a significantly reduced output filter inductor. These make the converters very suitable for battery chargers. However, the converters in [7], [19], and [21] require many power diodes in the rectifier stage, which could increase the conduction loss of the rectifier and may need additional heat sinks for radiating the heat generated from the diodes. Moreover, the converter in [6] needs many additional passive components such as an inductor and a capacitor, which has no powering role. The converters in [7], [20], and [21] use insulated-gate bipolar transistors as lagging-leg switches, which precludes the use of high switching frequency to realize smaller magnetic components and capacitors. Its switching frequency was very low, such as 41 or 47 kHz. In addition, the transformer’s utilization of the converter in [20] is very low. Fig. 2 shows the rectifier output voltage waveform of the converter in [20], where n1 and n2 mean the turns ratios of FB and HB transformers, respectively. As shown in the figure, the sections of FB and HB powering are separated over the switching period. Hence, if the converter is worked with the battery recharging strategy in Fig. 3 [22], [23], the HB transformer will transfer most of the battery recharging power

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Fig. 4. Proposed dc–dc converter.

when the battery voltage is low, i.e., 250 V. In addition, the section of FB powering will gradually increase as the battery voltage increases. In summary, maximum HB powering occurs at the beginning of recharging (A) and can be estimated as 250 V × IBattery . Similarly, maximum FB powering occurs at the end of constant-current charging (B) and is calculated as 450 V × IBattery . However, the effective power rating required in the converter with the battery recharging profile in Fig. 3 is 450 V × IBattery . Then the transformer’s utilization T.U. can be calculated as follows: T.U. =

=

Peﬀective_power PActually_designed_power 450 V · IBattery = 0.64. 250 V · IBattery + 450 V · IBattery

(1)

As shown in (1), the T.U. is low, which means that the transformers are not effectively used as EV battery chargers. III. P ROPOSED C ONVERTER FOR EV BATTERY C HARGERS A. Circuit Configuration Fig. 4 shows the circuit configuration of the proposed ZVS FB converter. A ZVS FB converter is composed of active switches of Q1-4 , a transformer T1 , a resonant capacitor Cr , an FB rectifier consisting of power diodes D1-4 , and an output LC filter, where Q1 and Q2 are leading-leg switches and Q3 and Q4 are lagging-leg switches. An HB converter, which consists of the lagging-leg switches, a transformer T2 , a blocking capacitor CB , and an FB rectifier of D3 , D4 , D5 , and D6 , is in parallel with the FB converter. Llk1 and Llk2 are the leakage inductances of two transformers. In the proposed converter, the dot placement of two transformers is very important in obtaining the aforementioned advantages. As shown in Fig. 4, the primary dots are in the same direction, but the secondary dots should be arranged in the opposite direction. B. Operation Principle Fig. 5 shows key operating waveforms of the proposed converter. In the figure, TS is the switching period, Tdead is the dead time between the same leg switches’ driving signals, and TΦ is the phase-shifted time between the two legs. The proposed converter is controlled by adjusting TΦ to regulate

Fig. 5.

Key operating waveforms of the proposed converter.

the output current or voltage. In the proposed converter, each switching period is divided into two half cycles, i.e., t0 –t6 and t6 –t12 , and the operational principles of two half cycles

LEE AND MOON: HB ZVS FB CONVERTER WITH REDUCED CONDUCTION LOSS FOR EV BATTERY CHARGERS

Fig. 6.

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Operating circuits during the first half cycle: (a) Mode 1, (b) Mode 2, (c) Mode 3, (d) Mode 4, (e) Mode 5, and (f) Mode 6.

are symmetric. Thus, only the first half cycle is described in this section. This half cycle can be subdivided into six modes, whose operating circuits are shown in Fig. 6. For simplicity of analysis, several assumptions are made. 1) Output inductor LO is large enough to be considered as a constant-current source during a switching period. 2) Two transformers T1 and T2 have the turns ratios of n1 and n2 , respectively. 3) T1 and T2 have the leakage inductances of Llk1 and Llk2 , respectively. 4) Blocking capacitor CB is large enough to be considered as a constant-voltage source of VIN /2. 5) The primary active switches are all MOSFETs with the parasitic diodes of Db1 , Db2 , Db3 , and Db4 . 6) The output capacitance values of all the MOSFETs have the same capacitance of COSS . Mode 1 [t0 –t1 ]: Mode 1 begins when Q1 and Q3 are in ONstate and only D1 and D5 are conducting in the rectifier stage. During this mode, the primary voltage Vp1 of transformer T1 is the positive input voltage and the primary voltage Vp2 of transformer T2 is the negative half of the input voltage. The primary currents of ip1 and ip2 are given as ip1 (t) = n1 IO

and

ip2 (t) = −n2 IO .

(2)

Resonant capacitor voltage VCr is calculated from ip1 as VCr (t) =

n1 IO n1 DIO TS (t − t0 )−0.5ΔVR , where ΔVR ≈ . Cr Cr (3)

Then, the secondary voltages of transformers and the input voltage of the LC output filter are obtained as follows: VS1 (t) = n1 (VIN − VCr (t)) = n1 VIN + 0.5n1 ΔVR − VS2 (t) = 0.5n2 VIN

n21 IO (t − t0 ) Cr

Vrec (t) = VS1 (t) + VS2 (t).

(4)

From (4), it is noted that in this mode, both FB and HB converters transfer the power required in the output port. Mode 2 [t1 –t2 ]: Mode 2 begins when Q1 is turned OFF at t1 . In the rectifier stage, the output inductor current IO still flows through D1 and D5 . Hence, the voltage across COSS1 is linearly charged, and the voltage across COSS2 is linearly discharged by the energy stored in the output inductor LO as follows: VQ1 (t) =

n1 IO (t − t1 ) and VQ2 (t) = VIN − VQ1 (t). (5) 2COSS

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Vp1 is the same with VQ2 , and VCr continuously increases with the same slope as in mode 1. VS1 ramped down to zero, and Vrec falls to 0.5n2 VIN with the slope of 1 dVS1 (t) d Vrec (t) 1 2 = = −n1 IO + . (6) dt dt 2COSS Cr Mode 3 [t2 –t3 ]: Mode 3 begins when the voltage across COSS2 reaches zero at t2 . Then, the parasitic diode Db2 of Q2 begins to conduct, and Q2 is turned ON with ZVS. Vp1 and VS1 are continuously maintained at zero in this mode, and Llk1 starts to resonate with Cr . By the resonance, ip1 is reset as in Fig. 5 and the commutation between D1 and D4 is progressed. This means that the excessive circulating current existing in the conventional ZVS FB converter is removed, which results in a reduction in primary-conduction loss. VCr increases very slightly to 0.5ΔVR . Mode 4 [t3 –t4 ]: Mode 4 begins when the commutation between D1 and D4 is completed at t3 , and only D4 and D5 conduct in the rectifier. During this mode, ip1 remains zero and the FB converter stops the power transfer. That is, in this mode, only the HB converter transfers the power required in the output port. Mode 5 [t4 –t5 ]: Mode 5 begins when Q3 is turned OFF at t4 . Then, the resonance of COSS3 , COSS4 , Llk1 , and Llk2 occurs in the primary power path. The voltage across COSS3 or COSS4 is charged or discharged by the resonance, respectively. Vp1 is decreased from zero to −VIN , and Vrec (t) is the same with VS1 , which decreases with a resonant waveform. The commutation between D2 and D5 is also progressed. The state equations in this mode are given as follows: n2 d VQ3 (t) = (1 + )ip1 (t) + n2 IO 2COSS dt n1 2 n1 Llk1 + n22 Llk2 dip1 (t) n1 dt VQ4 (t) = VIN − VQ3 (t) Vp1 (t) = −VQ3 (t), Vp2 (t) = VQ3 (t) − 0.5VIN VS1 (t) = −n1 VQ3 (t) − 0.5n1 ΔVR dip1 (t) , VS1 (t) + VS2 (t) = 0 dt n2 ip1 (t) ip2 (t) = −n2 IO − ip1 (t), iD2 (t) = − , n1 n1 − n1 Llk1

ip2 (t) . n2

(7)

The initial conditions for solving the stage equations are as follows: ip1 (t4 ) = 0

and

VQ3 (t4 ) = 0.

During this mode, Vp1 and Vp2 become the negative input voltage and the positive half of the input voltage, respectively. In the rectifier stage, the commutation between D4 and D6 is progressed and VS2 remains zero. In this mode, ip1 is −n1 IO and VCr decreases from 0.5ΔVR with the slope of n1 IO /Cr . The other relevant currents are expressed as ip2 (t) =

0.5VIN (t − t5 ) Llk2

iD6 (t) =

iD4 (t) = IO − iD6 (t).

ip2 (t) n2 (9)

Modes 7–12 [t6 –t12 ]: The operations from mode 7 to mode 12 are the same as previous modes except for the direction of powering path. C. Voltage Conversion Ratio M

= −(n1 + n2 )VQ3 (t) + 0.5(n2 VIN − n1 ΔVR )

iD5 (t) = −

Fig. 7. Simplified waveforms of rectifier output: (a) proposed converter and (b) ZVS FB converter in Fig. 1.

Since the duration periods of modes 2, 5, and 6 are practically very narrow in the proposed converter and hence they can be ignored, the simplified rectifier output voltage Vrec (t) can be shown as in Fig. 7(a). Fig. 7(b) shows the rectifier output voltage waveform of the ZVS FB converter in Fig. 1. Comparing the two figures, we can know that the freewheeling time, which Vrec (t) is maintained at zero, does not exist in the proposed converter. This means that the proposed converter continuously transfers the power required in the output port without any circulating current in the primary side. In the proposed converter, the voltage of resonant capacitor VCr is practically small. Thus, averaging the voltage waveform of Vrec (t) without considering the effect of Cr in Fig. 7(a) gives the voltage conversion ratio of the proposed converter as follows: M (D) =

(8)

Mode 6 [t5 –t6 ]: Mode 6 begins when the voltage across COSS4 reaches zero at t5 . Then, the parasitic diode Db4 of Q4 begins to conduct, and Q4 is turned ON with ZVS. The commutation between D2 and D5 is also completed at t5 .

VO = 2n1 D + 0.5n2 . VIN

(10)

Similarly, the voltage conversion ratio of the ZVS FB converter in Fig. 1 is given as M (D) =

VO = 2nD. VIN

(11)

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Fig. 9. ZVS energy of lagging-leg switches. Fig. 8.

Normalized voltage conversion ratio versus duty ratio.

For comparison, the two conversion ratios can be normalized based on the turns ratio of the FB transformer and are obtained respectively as follows: Mnorm_proposed (D) = Mnorm_conventional (D) =

VO = 2D + 0.5α n1 VIN VO = 2D nVIN

(12)

where 0 < α(= n2 /n1 ) ≤ 1. The two normalized M (D)’s are shown in Fig. 8. From the figure, it is noted that the gain of the proposed converter is much higher than that of the conventional converter. Due to this, the turns ratio of the FB transformer in the proposed converter can be designed with a smaller value than that in the conventional converter with same duty ratio so that it is possible that both primary-conduction loss and secondary-voltage stress are significantly reduced. For simply analyzing the voltage conversion ratio, we ignored the effect of resonant capacitor Cr . However, with considering the resonant capacitor, the turns ratio of T1 can be slightly changed. As explained in the previous section, the voltage of resonant capacitor appears on Vrec (t), as shown in Fig. 7(a), and the voltage conversion ratio is slightly increased in practice. Consequently, the turns ratio of T1 will be designed with smaller value than the value designed without considering the effect of Cr . D. ZVS Range and Duty-Cycle Loss As described in Section III-B, the ZVS operation of leadingleg switches is easily achieved over whole output conditions due to the sufficient energy stored in the output inductor. However, since the ZVS operation of lagging-leg switches is carried out by the resonance of Llk1 , Llk2 , and 2Coss , it could not be well achieved in the case that the energy stored in the two leakage inductors is small. Adding large resonant inductors in series with transformers (or increasing the two leakage inductances) can extend the ZVS range, but it is widely known that it reduces effective duty cycle (or increases dutycycle loss). To compensate for this, the turns ratio of the transformer should be designed to be larger, thereby increasing both primary-conduction loss and secondary-voltage stress.

In order to extend the ZVS range of lagging-leg switches, the proposed converter uses only magnetizing current of T2 , i.e., imag_T 2 (t), as described in Fig. 9. As shown in Fig. 9, the current ripple of ΔIR obtained from small magnetizing inductance of T2 increases the ZVS energy. Thus, it is possible that ZVS operation of lagging-leg switches is well achieved during the battery recharging process in Fig. 3. Since the magnetizing inductor of T2 is in parallel with powering path, the effect of duty-cycle loss is minimized. In addition, although the small magnetizing inductance of T2 is used, RMS current stress in the primary side never increases over that of the conventional ZVS FB converter, according to the principle in [24]–[27]. The RMS current stress will be described in the next section in detail. E. Primary-Conduction Loss There exists circulating current in the ZVS FB converter in Fig. 1, which flows through a transformer and switches although the primary voltage of the transformer remains zero. This current greatly increases primary-conduction loss in constant-input and variable-output applications such as EV battery chargers and RF power generator. On the other hand, in the proposed converter, the primary current of the FB transformer is reset during mode 3 and is maintained at zero, whereas the primary voltage in the FB side remains zero. That is, it can be said that there is no circulating current on the FB side of the proposed converter during battery recharging process. As mentioned in Section III-B, the turns ratio of FB transformer is designed with smaller value than the ZVS FB converter in Fig. 1 due to higher gain. Smaller turns ratio significantly reduces the magnitude of battery recharging current reflected to the primary of the FB side. Thus, the current stresses of leading-leg switches (Q1 and Q2 ) and FB transformer (T1 ) are greatly reduced. In the proposed converter, magnetizing inductance of the HB transformer is designed to be small for extending the ZVS energy of lagging-leg switches. However, the RMS current stress of leading-leg switches and the FB transformer never increases because the magnetizing current of the HB transformer never flows through Q1 , Q2 , and T1 . This is in contrast to the fact that, in conventional ZVS FB converters, using small magnetizing inductance for wider ZVS range increases the current stress of

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Fig. 10. Equivalent circuit of rectifier stage in the case that: (a) both FB and HB converters transfer the power; and (b) only HB converter transfers the power.

all switches and transformers [26], [27]. In addition, in spite of using small magnetizing inductance, the RMS current stresses of lagging-leg switches (Q3 and Q4 ) and HB transformer (T2 ) become smaller than those of the ZVS FB converter in Fig. 1 because the average value of magnetizing current of T2 is zero within a half-switching period, and its contribution to RMS current stress is negligible at medium-to-high power conditions [24]–[27]. In summary, due to no circulating current, smaller turns ratio, and the same principle in [24]–[27], the RMS current stresses of all switches and transformers are smaller than those of the conventional ZVS FB converter so that it is possible that primary-conduction loss is significantly reduced. This will be verified with the experimental results in the final section.

Fig. 11.

Normalized voltage stress of rectifier diodes.

Fig. 12.

Output filter inductance required for IPP = 500 mA.

F. Secondary-Voltage Stress Fig. 10 shows equivalent circuits of rectifier stage during two powering modes in order to get the voltage stress of rectifier diodes in the proposed converter. From the equivalent circuits, the voltage stress of diodes can be approximately obtained considering voltage spike caused by parasitic circuit elements as follows: 2+α VO_ max = k1 VO_ max Vstress(D1,2,5,6) = 0.9 + 0.5α 1 Vstress(D3,4) = VO_ max = k2 VO_ max (13) 0.9 + 0.5α

TABLE I C OMPONENTS L IST

where 0 < α(= n2 /n1 ) < 1. To get the voltage stress of diodes in a conventional ZVS FB converter, let us first assume that the maximum value of effective duty ratio is 45%. In turn, by considering the voltage spike caused by parasitic circuit elements and the battery recharging profile in Fig. 3, the voltage stress can be obtained as Vstress(D1,2,3,4) = 2.2VO_ max .

(14)

Equations (13) and (14) can be normalized by VO , and Fig. 11 shows the normalized voltage stresses. From the figure, it is noted that the proposed converter requires four high-

voltage-rated diodes and two low-voltage-rated diodes, whereas conventional ZVS FB converter needs only high-voltage-rated diodes. Note that low-voltage diodes have advantages such as much lower ON-state voltage and better recovery characteristics compared with high-voltage diodes.

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Fig. 13.

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Operating waveforms (ip1 (t), ip2 (t), and Vp1 (t)) of the proposed converter: (a) when VO = 250 V, (b) when VO = 350 V, and (c) when VO = 450 V.

Fig. 14. Operating waveforms (Vp1 (t) and Vrec (t)) of the proposed converter: (a) when VO = 250 V, (b) when VO = 350 V, and (c) when VO = 450 V.

The conventional ZVS FB converter uses the four highvoltage diodes during entire modes of the battery recharging process in Fig. 3. On the other hand, in the proposed converter, the two low-voltage diodes of D3 and D4 , which feature good performance, are mainly used to transfer the power required in the output port while the section of HB powering is increasing over the battery recharging process, particularly during constant-current mode or constant-voltage mode with lightload conditions. Due to this operation, the conduction loss of rectifier stage can be greatly reduced and the conversion efficiency can be further improved. G. Filter Requirement The output filter inductor’s values can be derived from the rectifier output voltages in Fig. 6 using (10), (11), and (15) as (16) and (17) V = LO

IPP ΔT

(15)

where IPP is the peak-to-peak value of current flowing through LO , and V is the voltage applied to LO during the time of ΔT . For the proposed converter LO_proposed =

VO TS (1 + 0.5α − Mnorm (D)) IPP 0.5α × 1− . Mnorm (D)

(16)

Fig. 15. Output filter inductor: (a) for the conventional ZVS FB converter (1.3 mH, CM400060) and (b) for the proposed converter (700 μH, CM343060).

For the conventional ZVS FB converter LO_Conventional =

VO TS (1 − Mnorm (D)) . 2IPP

(17)

For a quantitative analysis, these two parameters are used. • IPP = 500 mA. • Switching frequency fS = 100 kHz. Since in EV battery chargers, the current ripple of LO is worst when the battery voltage is lowest, VO is given as 250 V in Fig. 3.

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Fig. 16. ZVS waveforms of lagging-leg switches: (a) when VO = 250 V, (b) when VO = 350 V, and (c) when VO = 450 V.

Fig. 17. ZVS waveforms of leading-leg switches: (a) when VO = 250 V, (b) when VO = 350 V, and (c) when VO = 450 V.

Fig. 12 shows the required filter inductance in function of normalized voltage conversion ratio. With Figs. 8 and 12, it is known that the output inductor’s value of the proposed converter is much smaller than that of the conventional converter. The use of the output filter inductor with smaller inductance is very beneficial in terms of size and weight as EV battery chargers. H. Transformer Utilization T.U. In [20], the sections of FB and HB powering are separated over the switching period, which lower transformer utilization for EV battery chargers. On the other hand, in the proposed converter, HB part always helps powering of the FB part over the entire switching period, as shown in Fig. 7(a). This improves transformer utilization over the battery recharging profile in Fig. 3. For example, maximum HB powering occurs at the beginning of recharging (A) in Fig. 3 and can be simply estimated as 250 V × IBattery . As the battery voltage increases, FB powering will gradually increase, but it always receives the help of HB powering with 250 V × IBattery . Thus, maximum FB powering occurs at the end of constant-current charging (B) in Fig. 3 and can be calculated as 200 V × IBattery . Then, the transformer’s utilization, T.U. can be calculated as follows: T.U. = =

Peﬀective_power PActually_designed_power 450 V · IBattery = 1.0. (18) 250 V · IBattery + 200 V · IBattery

Consequently, it is known that the proposed converter has better T.U. than the topology in [20] consisting of FB and HB converters. IV. E XPERIMENTAL R ESULTS In order to verify the performance of the proposed converter as EV battery chargers, it is realized with the battery charger specification given below and the battery recharging profile in Fig. 3: • input voltage VIN = 400 V; • maximum output power PO(max) = 2 kW; • switching frequency fS = 100 kHz. Considering the fact that the power-handling capacity of the HB converter is generally smaller than that of the FB converter, the HB transformer was designed to be able to process only about 33% of the total power by adjusting the turns ratio. A prototype converter was built using the components listed in Table I, and the magnetizing inductance of the HB transformer T2 was set as 100 μH for extending the ZVS range of laggingleg switches. Figs. 13 and 14 show key waveforms of the proposed converter during constant-current mode in Fig. 3. As shown in the figures, all measured waveforms are well following the theoretical waveforms described in Fig. 5 and the primary current of HB inverter ip2 (t) exactly corresponds to Fig. 9. We can also see that the circulating current existing in the conventional ZVS FB converter is removed in the proposed converter from ip1 (t) in Fig. 13. In addition, due to the staircase output voltage

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Fig. 19. Efficiency over the battery charging profile in Fig. 3: (a) efficiency during constant-current mode and (b) efficiency during constant-voltage mode.

Fig. 19 shows efficiency values measured during battery recharging process. As shown in the figures, the efficiency of the proposed converter is advanced over that of the conventional ZVS FB converter during constant-current mode. This is the result of reduced primary-conduction loss coming from no circulating current, lower turns ratio, and the same principle in [24]–[27]. The efficiency during constant-voltage mode is also over the conventional converter, which results from wider ZVS range. Fig. 18. RMS current stress measured during constant-current charging mode: (a) lagging-leg switches, (b) leading-leg switches, and (c) transformers.

V. C ONCLUSION waveform of the rectifier in Fig. 14, the output filter inductor’ size becomes smaller than that of the conventional ZVS FB converter, as shown in Fig. 15. Figs. 16 and 17 show ZVS waveforms of switches during constant-current charging mode. From Figs. 16 and 17, it is clear that all switches in the proposed converter are always turned ON with ZVS during the battery recharging process. Fig. 18 shows RMS current stress measured during constantcurrent charging mode in Fig. 3. It is verified that, as explained in Section III-E, the proposed converter has much smaller RMS current stress in the primary side in spite of wider ZVS range compared with the conventional ZVS FB converter.

In order to extend the driving range in EV mode, the power rating of battery chargers in EVs or PHEVs will continue to increase. As a result, research on high-power high-efficiency topologies is necessarily required. In this paper, a new softswitching dc–dc converter for EV battery on-board chargers has been proposed. From the analysis results, it is worth noting that the proposed converter has following advantages: 1) reduction of primary- and secondary-conduction losses; 2) no circulating current, less duty-cycle loss, and full ZVS capability; 3) smaller-sized output filter inductor.

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Due to these advantages, it is possible for the proposed converter to have smaller power loss and higher efficiency compared with conventional ZVS FB converters as battery chargers. From the experiment results of the proposed converter realized with a scale-downed 2-kW battery charger, all aforementioned advantages were verified and its effectiveness and feasibility were confirmed. Therefore, it can be said that the proposed converter is suitable for the dc–dc converter in high-power battery chargers for EVs or PHEVs. R EFERENCES [1] I. Aharon and A. Kuperman, “Topological overview of powertrains for battery-powered vehicles with range extenders,” IEEE Trans. Power Electron., vol. 26, no. 3, pp. 868–876, Mar. 2011. [2] J. Cao and A. Emadi, “A new battery/ultracapacitor hybrid energy storage system for electric, hybrid, and plug-in hybrid electric vehicles,” IEEE Trans. Power Electron., vol. 27, no. 1, pp. 122–132, Jan. 2012. [3] M. B. Camara, H. Gualous, F. Gustin, A. Berthon, and B. Dakyo, “DC/DC converter design for supercapacitor and battery power management in hybrid vehicle applications-polynomial control strategy,” IEEE Trans. Ind. Electron., vol. 57, no. 2, pp. 587–597, Feb. 2010. [4] M. Pahlevaninezhad, P. Das, J. Drobnik, G. Moschopoulos, P. Jain, and A. Bakhshai, “A nonlinear optimal control approach based on the controlLyapunov function for an AC/DC converter used in electric vehicles,” IEEE Trans. Ind. Inform., vol. 8, no. 3, pp. 596–612, Aug. 2012. [5] M. Yilmaz and P. T. Krein, “Review of charging power levels and infrastructure for plug-in electric and hybrid vehicle,” in Proc. IEVC, 2012, pp. 1–8. [6] M. Pahlevaninezhad, P. Das, J. Drobnik, P. Jain, and A. Bakhshai, “A novel ZVZCS full-bridge DC/DC converter used for electric vehicle,” IEEE Trans. Power Electron., vol. 27, no. 6, pp. 2752–2769, Jun. 2012. [7] B. Gu, J. Lai, N. Kees, and C. Zheng, “Hybrid-switching full-bridge DC–DC converter with minimal voltage stress of bridge rectifier, reduced circulating losses and filter requirement for electric vehicle battery chargers,” IEEE Trans. Power Electron., vol. 28, no. 3, pp. 1132–11441, Mar. 2013. [8] Y. Cho and J. S. Lai, “Digital plug-in repetitive controller for single-phase bridgeless PFC converters,” IEEE Trans. Power Electron., vol. 28, no. 1, pp. 165–175, Jan. 2013. [9] M. Pahlevaninezhad, P. Das, J. Drobnik, P. Jain, and A. Bakhshai, “A new control approach based on the differential flatness theory for an AC/DC converter used in electric vehicles,” IEEE Trans. Power Electron., vol. 27, no. 4, pp. 2085–2103, Apr. 2012. [10] F. Musavi, W. Eberle, and W. G. Dunford, “A high-performance singlephase bridgeless interleaved PFC converter for plug-in hybrid electric vehicle battery chargers,” IEEE Trans. Ind. Appl., vol. 47, no. 4, pp. 1833– 1843, Jul. 2011. [11] F. Musavi, M. Edington, W. Eberle, and W. G. Dunford, “Evaluation and efficiency comparison of front end AC–DC plug-in hybrid charger topologies,” IEEE Trans. Smart Grid., vol. 3, no. 1, pp. 413–421, Mar. 2012. [12] D. Zhang, F. Wang, R. Burgos, and D. Boroyevich, “Total flux minimization control for integrated inter-phase inductors in paralleled, interleaved three-phase two-level voltage-source converters with discontinuous space-vector modulation,” IEEE Trans. Power Electron., vol. 27, no. 4, pp. 1679–1688, Apr. 2012. [13] M. Pahlevaninezhad, P. Das, J. Drobnik, P. Jain, and A. Bakhshai, “A ZVS interleaved boost AC/DC converter used in plug-in electric vehicles,” IEEE Trans. Power Electron., vol. 27, no. 8, pp. 3513–3529, Aug. 2012. [14] J. A. A. Qahouq, L. Huang, and D. Huard, “Efficiency-based auto-tuning of current sensing and sharing loops in multiphase converters,” IEEE Trans. Power Electron., vol. 23, no. 2, pp. 1009–1013, Mar. 2008. [15] J. Zhang, F. Zhang, X. Xie, D. Jiao, and Z. Qian, “A novel ZVS DC/DC converter for high power applications,” IEEE Trans. Power Electron., vol. 19, no. 2, pp. 420–429, Mar. 2004. [16] T. Kim, S. Lee, and W. Choi, “Design and control of the phase shift full bridge converter for the on-board battery charger of electric forklifts,” J. Power Electron., vol. 12, no. 1, pp. 113–119, Jan. 2012.

[17] B. P. McGrath, D. G. Holmes, P. J. McGoldrick, and A. D. McIver, “Design of a soft-switched 6-kW battery charger for traction applications,” IEEE Trans. Power Electron., vol. 22, no. 4, pp. 1136–1144, Jul. 2007. [18] D. Gautam, F. Musavi, M. Edington, W. Eberle, and W. G. Dunford, “An automotive on-board 3.3 kW battery charger for PHEV application,” IEEE Trans. Veh. Technol., vol. 61, no. 8, pp. 3466–3474, Oct. 2012. [19] C. Liu, B. Gu, J. Lai, M. Wang, Y. Ji, G. Cai, Z. Zhao, C. Chen, C. Zheng, and P. Sue, “High-efficiency hybrid full-bridge-half-bridge converter with shared ZVS lagging leg and dual outputs in series,” IEEE Trans. Power Electron., vol. 28, no. 2, pp. 849–861, Feb. 2013. [20] W. Yu, J. Lai, W. Lai, and H. Wan, “Hybrid resonant and PWM converter with high efficiency and full soft-switching range,” IEEE Trans. Power Electron., vol. 27, no. 12, pp. 4925–4933, Dec. 2012. [21] B. Gu, C. Lin, B. Chen, J. Dominic, and J. Lai, “Zero-voltage-switching PWM resonant full-bridge converter with minimized circulating losses and minimal voltage stresses of bridge rectifiers for electric vehicle battery chargers,” IEEE Trans. Power Electron., vol. 28, no. 10, pp. 4657–4667, Oct. 2013. [22] E. Inoa and J. Wang, “PHEV charging strategies for maximized energy saving,” IEEE Trans. Veh. Technol., vol. 60, no. 7, pp. 2978–2986, Sep. 2011. [23] B.-Y. Chen and Y.-S. Lai, “New digital-controlled technique for battery charger with constant current and voltage control without current feedback,” IEEE Trans. Ind. Electron., vol. 59, no. 3, pp. 1545–1553, Mar. 2012. [24] M. Borage, S. Tiwari, S. Bhardwaj, and S. Kotaiah, “A full-bridge DC–DC converter with zero-voltage-switching over the entire conversion range,” IEEE Trans. Power Electron., vol. 23, no. 4, pp. 1743–1750, Jul. 2008. [25] M. Borage, S. Tiwari, and S. Kotaiah, “A passive auxiliary circuit achieves zero-voltage-switching in full-bridge converter over entire conversion range,” IEEE Trans. Power Electron., vol. 3, no. 4, pp. 141–143, Dec. 2005. [26] I. Lee and G. W. Moon, “Soft-switching DC/DC converter with a full ZVS range and reduced output filter for high-voltage applications,” IEEE Trans. Power Electron., vol. 28, no. 1, pp. 112–122, Jan. 2013. [27] I. Lee, S. Y. Cho, and G. W. Moon, “Improved phase-shift PWM converters for larger-sized PDP slim sustain power module,” IEEE Trans. Power Electron., vol. 28, no. 2, pp. 945–958, Feb. 2013.

Il-Oun Lee (S’10–M’13) received the B.S. degree in electrical and electronic engineering from Kyungpook National University, Daegu, Korea, in 2000; the M.S. degree in electrical engineering from Seoul National University, Seoul, Korea, in 2002; and the Ph.D. degree from Korea Advanced Institute of Science and Technology, Daejeon, Korea, in 2013. He was a Research Engineer in the Plasma Display Panel Development Group, Samsung SDI, Yongin, Korea, for five years beginning in 2003. Since 2008, he has been a Senior Engineer with the Power Advanced Development Group, Samsung Electro-Mechanics Company, Ltd., Suwon, Korea. His current research interests include dc–dc converters, powerfactor-correction ac–dc converters, LED driver, battery charger for electric vehicle, digital display power systems, and digital control approach of dc–dc converters.

Gun-Woo Moon (S’92–M’00) received the M.S. and Ph.D. degrees in electrical engineering from Korea Advanced Institute of Science and Technology (KAIST), Daejeon, Korea, in 1992 and 1996, respectively. He is currently a Professor with the Department of Electrical Engineering, KAIST. His research interests include modeling, design, and control of power converters; soft-switching power converters; resonant inverters; distributed power systems; power factor correction; electric drive systems; driver circuits of plasma display panels; and flexible ac transmission systems.

IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 61, NO. 8, AUGUST 2014

Half-Bridge Integrated ZVS Full-Bridge Converter With Reduced Conduction Loss for Electric Vehicle Battery Chargers Il-Oun Lee, Member, IEEE, and Gun-Woo Moon, Member, IEEE

Abstract—A half-bridge integrated zero-voltage-switching (ZVS) full-bridge converter with reduced conduction loss for battery on-board chargers in electric vehicles (EVs) or plug-in hybrid electric vehicles (PHEVs) is proposed in this paper. The proposed converter features a reduction in primary-conduction loss and a lower secondary-voltage stress. In addition, the proposed converter has the most favorable characteristics as battery chargers as follows: a full ZVS capability and a significantly reduced output filter size due to the improved output waveform. In this paper, the circuit configuration, operation principle, and relevant analysis results of the proposed converter are described, followed by the experimental results on a prototype converter realized with a scale-downed 2-kW battery charger for EVs or PHEVs. The experimental results validate the theoretical analysis and show the effectiveness of the proposed converter as battery on-board chargers for EVs or PHEVs. Index Terms—Battery charger, electric vehicle (EV), full-bridge (FB) converter, hybrid electric vehicle (HEV), zero-voltage switching (ZVS).

I. I NTRODUCTION

W

ITH accelerated global warming, decreasing natural resources, increasing fuel price, and economical issues, vehicles with electric propulsion, such as hybrid electric vehicles (HEVs), plug-in HEVs (PHEVs), battery electric vehicles (BEVs or EVs), and fuel cell electric vehicles, are gradually growing. These vehicles need commonly rechargeable batteries as the energy source of electric traction system [1]. Among them, PHEVs or EVs require a higher capacity and larger sized battery pack compared with other vehicles because the battery is a main energy source in PHEVs or EVs [2], [3]. The high-energy-density battery pack in PHEVs or EVs is typically recharged from the ac utility grid via an ac–dc converter named as battery charger. For low harmonic contents on the ac

Manuscript received April 8, 2013; revised July 5, 2013; accepted August 18, 2013. Date of publication September 18, 2013; date of current version February 7, 2014. This work was supported by the National Research Foundation of Korea under Grant 2012-0000981 funded by the Ministry of Education, Science and Technology of the Korean government. I.-O. Lee is with the Power Advanced Development Group, Samsung Electro-Mechanics Company, Ltd., Suwon 443-743, Korea. G.-W. Moon is with the Department of Electrical Engineering, Korea Advanced Institute of Science and Technology, Daejeon 305-701, Korea (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TIE.2013.2282608

utility grid and high efficiency, most of battery chargers have generally the basic form of an ac–dc converter with a power factor corrector (PFC), followed by an isolated dc–dc converter [4]–[9]. There are key requirements in the development of EV battery chargers. First, it is imperative to reduce their size and weight in order to facilitate packaging and to highlight the utilization factor of energy. Namely, the design for higher power density and lower weight is required. Furthermore, the conversion efficiency should be maximized during whole output conditions or battery recharging process to maximize the fuel saving and emission reduction. In order to achieve these requirements, it is necessary to adopt higher switching frequencies and softswitching technologies since a higher switching frequency is the key to reducing the size and weight of passive components used in high-power applications, and soft-switching technologies significantly lower the generated switching losses. In addition, in the PFC stage, a bridgeless design should be carried out because excessive conduction loss is generated due to the forward voltage drop for each of the bridge diodes, particularly at a lower line input voltage, which decreases the overall efficiency and greatly increases the size and weight of heat sink [10], [11]. In order to obtain higher conversion efficiency, particularly at high power levels, interleaving or parallel approaches can be considered because they significantly reduce the generated conduction loss [12]–[14]. In addition, in the case where the output voltage requirement of the battery charger is high, the rectifier diodes in the dc–dc converter could experience a serious voltage oscillation and spike. Then, lossy snubber circuitry and higher voltage-rated diodes must be required, which cause the increase in power loss, size, and weight. Thus, in designing the rectifier stage in the dc–dc converter, the design that can avoid the aforementioned problem should be also taken into account [15]. In addition, in order to maintain high efficiency under lowpower conditions, it is necessary to minimize the amount of circulating energy in the dc–dc converter. In this paper, a zero-voltage-switching (ZVS) full-bridge (FB) converter with reduced conduction loss for battery onboard chargers in PHEVs or EVs is proposed. The proposed converter consists of an FB converter integrated with a symmetric half-bridge (HB) converter in parallel. This architecture makes it possible for the proposed converter to feature a reduction in primary-conduction loss, which comes from the reduction of circulating current and the much smaller turns ratio of FB transformer. The proposed converter also has a

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Fig. 2. Rectifier output voltage waveform of the converter in [20].

Fig. 1. ZVS FB dc–dc converter with wide ZVS range and low-secondaryvoltage stress for EV battery chargers.

lower secondary-voltage stress because of the hybrid structure, which can result in a reduction of secondary-conduction loss. In addition, in the proposed converter, a full ZVS capability without adding any heavy extra inductors and a significantly reduced output filter size due to the improved output waveform are obtained. Thus, the converter’s weight or size can be greatly reduced, which is necessarily required in EV battery charger applications. The circuit configuration, operation principle, and relevant analysis results of the proposed converter are described in this paper. The experimental results on a prototype converter realized with a scale-downed 2-kW battery charger for EVs or PHEVs validate the theoretical analysis and show the effectiveness of the proposed converter as EV battery chargers. II. R EVIEW OF DC–DC C ONVERTERS FOR BATTERY C HARGERS The most preferred dc–dc topology for battery on-board chargers in EVs or PHEVs is the ZVS FB converter operated at a constant frequency under pulsewidth control strategy [16]–[18]. The converter features minimal voltage and current stresses in the power devices, a natural ZVS capability, a low current ripple in the output current, and a simple structure. Since the battery voltage is very high and varies over a wide range during battery recharging process, however, the converter suffers from excessive conduction loss due to the large circulating current and large RMS current stress in the primary side and the very high voltage stress in the secondary side. In addition, ZVS operation is not achieved over whole output conditions. For these reasons, it is widely known that it is difficult to maximize the conversion efficiency as battery chargers. In addition, the converter needs a very large output filter inductor for sufficiently reducing the current ripple of battery recharging current. To increase the conversion efficiency, the ZVS FB converter in Fig. 1 can be considered for battery chargers. Since the

Fig. 3. Battery recharging strategy consisting of constant-current and constant-voltage modes.

resonant inductor LR added in the lagging-leg switches (Q3 and Q4 ) extends ZVS range and the additional diodes of Da1 and Da2 significantly reduce the voltage stress of the rectifier stage, the conduction loss in the rectifier stage and the switching loss are greatly reduced. However, there still exist other drawbacks such as the large circulating current and large RMS current stress in the primary side and the requirement of a large output filter inductor. In order to further improve the performance of EV battery chargers, new ZVS FB converters are presented in [6] and [7] and ZVS FB converters integrated with HB converter have been recently proposed in [19]–[21]. The converters commonly feature a wide ZVS range, a reduced circulating current, a reduced voltage stress in rectifier stage, and a significantly reduced output filter inductor. These make the converters very suitable for battery chargers. However, the converters in [7], [19], and [21] require many power diodes in the rectifier stage, which could increase the conduction loss of the rectifier and may need additional heat sinks for radiating the heat generated from the diodes. Moreover, the converter in [6] needs many additional passive components such as an inductor and a capacitor, which has no powering role. The converters in [7], [20], and [21] use insulated-gate bipolar transistors as lagging-leg switches, which precludes the use of high switching frequency to realize smaller magnetic components and capacitors. Its switching frequency was very low, such as 41 or 47 kHz. In addition, the transformer’s utilization of the converter in [20] is very low. Fig. 2 shows the rectifier output voltage waveform of the converter in [20], where n1 and n2 mean the turns ratios of FB and HB transformers, respectively. As shown in the figure, the sections of FB and HB powering are separated over the switching period. Hence, if the converter is worked with the battery recharging strategy in Fig. 3 [22], [23], the HB transformer will transfer most of the battery recharging power

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Fig. 4. Proposed dc–dc converter.

when the battery voltage is low, i.e., 250 V. In addition, the section of FB powering will gradually increase as the battery voltage increases. In summary, maximum HB powering occurs at the beginning of recharging (A) and can be estimated as 250 V × IBattery . Similarly, maximum FB powering occurs at the end of constant-current charging (B) and is calculated as 450 V × IBattery . However, the effective power rating required in the converter with the battery recharging profile in Fig. 3 is 450 V × IBattery . Then the transformer’s utilization T.U. can be calculated as follows: T.U. =

=

Peﬀective_power PActually_designed_power 450 V · IBattery = 0.64. 250 V · IBattery + 450 V · IBattery

(1)

As shown in (1), the T.U. is low, which means that the transformers are not effectively used as EV battery chargers. III. P ROPOSED C ONVERTER FOR EV BATTERY C HARGERS A. Circuit Configuration Fig. 4 shows the circuit configuration of the proposed ZVS FB converter. A ZVS FB converter is composed of active switches of Q1-4 , a transformer T1 , a resonant capacitor Cr , an FB rectifier consisting of power diodes D1-4 , and an output LC filter, where Q1 and Q2 are leading-leg switches and Q3 and Q4 are lagging-leg switches. An HB converter, which consists of the lagging-leg switches, a transformer T2 , a blocking capacitor CB , and an FB rectifier of D3 , D4 , D5 , and D6 , is in parallel with the FB converter. Llk1 and Llk2 are the leakage inductances of two transformers. In the proposed converter, the dot placement of two transformers is very important in obtaining the aforementioned advantages. As shown in Fig. 4, the primary dots are in the same direction, but the secondary dots should be arranged in the opposite direction. B. Operation Principle Fig. 5 shows key operating waveforms of the proposed converter. In the figure, TS is the switching period, Tdead is the dead time between the same leg switches’ driving signals, and TΦ is the phase-shifted time between the two legs. The proposed converter is controlled by adjusting TΦ to regulate

Fig. 5.

Key operating waveforms of the proposed converter.

the output current or voltage. In the proposed converter, each switching period is divided into two half cycles, i.e., t0 –t6 and t6 –t12 , and the operational principles of two half cycles

LEE AND MOON: HB ZVS FB CONVERTER WITH REDUCED CONDUCTION LOSS FOR EV BATTERY CHARGERS

Fig. 6.

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Operating circuits during the first half cycle: (a) Mode 1, (b) Mode 2, (c) Mode 3, (d) Mode 4, (e) Mode 5, and (f) Mode 6.

are symmetric. Thus, only the first half cycle is described in this section. This half cycle can be subdivided into six modes, whose operating circuits are shown in Fig. 6. For simplicity of analysis, several assumptions are made. 1) Output inductor LO is large enough to be considered as a constant-current source during a switching period. 2) Two transformers T1 and T2 have the turns ratios of n1 and n2 , respectively. 3) T1 and T2 have the leakage inductances of Llk1 and Llk2 , respectively. 4) Blocking capacitor CB is large enough to be considered as a constant-voltage source of VIN /2. 5) The primary active switches are all MOSFETs with the parasitic diodes of Db1 , Db2 , Db3 , and Db4 . 6) The output capacitance values of all the MOSFETs have the same capacitance of COSS . Mode 1 [t0 –t1 ]: Mode 1 begins when Q1 and Q3 are in ONstate and only D1 and D5 are conducting in the rectifier stage. During this mode, the primary voltage Vp1 of transformer T1 is the positive input voltage and the primary voltage Vp2 of transformer T2 is the negative half of the input voltage. The primary currents of ip1 and ip2 are given as ip1 (t) = n1 IO

and

ip2 (t) = −n2 IO .

(2)

Resonant capacitor voltage VCr is calculated from ip1 as VCr (t) =

n1 IO n1 DIO TS (t − t0 )−0.5ΔVR , where ΔVR ≈ . Cr Cr (3)

Then, the secondary voltages of transformers and the input voltage of the LC output filter are obtained as follows: VS1 (t) = n1 (VIN − VCr (t)) = n1 VIN + 0.5n1 ΔVR − VS2 (t) = 0.5n2 VIN

n21 IO (t − t0 ) Cr

Vrec (t) = VS1 (t) + VS2 (t).

(4)

From (4), it is noted that in this mode, both FB and HB converters transfer the power required in the output port. Mode 2 [t1 –t2 ]: Mode 2 begins when Q1 is turned OFF at t1 . In the rectifier stage, the output inductor current IO still flows through D1 and D5 . Hence, the voltage across COSS1 is linearly charged, and the voltage across COSS2 is linearly discharged by the energy stored in the output inductor LO as follows: VQ1 (t) =

n1 IO (t − t1 ) and VQ2 (t) = VIN − VQ1 (t). (5) 2COSS

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Vp1 is the same with VQ2 , and VCr continuously increases with the same slope as in mode 1. VS1 ramped down to zero, and Vrec falls to 0.5n2 VIN with the slope of 1 dVS1 (t) d Vrec (t) 1 2 = = −n1 IO + . (6) dt dt 2COSS Cr Mode 3 [t2 –t3 ]: Mode 3 begins when the voltage across COSS2 reaches zero at t2 . Then, the parasitic diode Db2 of Q2 begins to conduct, and Q2 is turned ON with ZVS. Vp1 and VS1 are continuously maintained at zero in this mode, and Llk1 starts to resonate with Cr . By the resonance, ip1 is reset as in Fig. 5 and the commutation between D1 and D4 is progressed. This means that the excessive circulating current existing in the conventional ZVS FB converter is removed, which results in a reduction in primary-conduction loss. VCr increases very slightly to 0.5ΔVR . Mode 4 [t3 –t4 ]: Mode 4 begins when the commutation between D1 and D4 is completed at t3 , and only D4 and D5 conduct in the rectifier. During this mode, ip1 remains zero and the FB converter stops the power transfer. That is, in this mode, only the HB converter transfers the power required in the output port. Mode 5 [t4 –t5 ]: Mode 5 begins when Q3 is turned OFF at t4 . Then, the resonance of COSS3 , COSS4 , Llk1 , and Llk2 occurs in the primary power path. The voltage across COSS3 or COSS4 is charged or discharged by the resonance, respectively. Vp1 is decreased from zero to −VIN , and Vrec (t) is the same with VS1 , which decreases with a resonant waveform. The commutation between D2 and D5 is also progressed. The state equations in this mode are given as follows: n2 d VQ3 (t) = (1 + )ip1 (t) + n2 IO 2COSS dt n1 2 n1 Llk1 + n22 Llk2 dip1 (t) n1 dt VQ4 (t) = VIN − VQ3 (t) Vp1 (t) = −VQ3 (t), Vp2 (t) = VQ3 (t) − 0.5VIN VS1 (t) = −n1 VQ3 (t) − 0.5n1 ΔVR dip1 (t) , VS1 (t) + VS2 (t) = 0 dt n2 ip1 (t) ip2 (t) = −n2 IO − ip1 (t), iD2 (t) = − , n1 n1 − n1 Llk1

ip2 (t) . n2

(7)

The initial conditions for solving the stage equations are as follows: ip1 (t4 ) = 0

and

VQ3 (t4 ) = 0.

During this mode, Vp1 and Vp2 become the negative input voltage and the positive half of the input voltage, respectively. In the rectifier stage, the commutation between D4 and D6 is progressed and VS2 remains zero. In this mode, ip1 is −n1 IO and VCr decreases from 0.5ΔVR with the slope of n1 IO /Cr . The other relevant currents are expressed as ip2 (t) =

0.5VIN (t − t5 ) Llk2

iD6 (t) =

iD4 (t) = IO − iD6 (t).

ip2 (t) n2 (9)

Modes 7–12 [t6 –t12 ]: The operations from mode 7 to mode 12 are the same as previous modes except for the direction of powering path. C. Voltage Conversion Ratio M

= −(n1 + n2 )VQ3 (t) + 0.5(n2 VIN − n1 ΔVR )

iD5 (t) = −

Fig. 7. Simplified waveforms of rectifier output: (a) proposed converter and (b) ZVS FB converter in Fig. 1.

Since the duration periods of modes 2, 5, and 6 are practically very narrow in the proposed converter and hence they can be ignored, the simplified rectifier output voltage Vrec (t) can be shown as in Fig. 7(a). Fig. 7(b) shows the rectifier output voltage waveform of the ZVS FB converter in Fig. 1. Comparing the two figures, we can know that the freewheeling time, which Vrec (t) is maintained at zero, does not exist in the proposed converter. This means that the proposed converter continuously transfers the power required in the output port without any circulating current in the primary side. In the proposed converter, the voltage of resonant capacitor VCr is practically small. Thus, averaging the voltage waveform of Vrec (t) without considering the effect of Cr in Fig. 7(a) gives the voltage conversion ratio of the proposed converter as follows: M (D) =

(8)

Mode 6 [t5 –t6 ]: Mode 6 begins when the voltage across COSS4 reaches zero at t5 . Then, the parasitic diode Db4 of Q4 begins to conduct, and Q4 is turned ON with ZVS. The commutation between D2 and D5 is also completed at t5 .

VO = 2n1 D + 0.5n2 . VIN

(10)

Similarly, the voltage conversion ratio of the ZVS FB converter in Fig. 1 is given as M (D) =

VO = 2nD. VIN

(11)

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Fig. 9. ZVS energy of lagging-leg switches. Fig. 8.

Normalized voltage conversion ratio versus duty ratio.

For comparison, the two conversion ratios can be normalized based on the turns ratio of the FB transformer and are obtained respectively as follows: Mnorm_proposed (D) = Mnorm_conventional (D) =

VO = 2D + 0.5α n1 VIN VO = 2D nVIN

(12)

where 0 < α(= n2 /n1 ) ≤ 1. The two normalized M (D)’s are shown in Fig. 8. From the figure, it is noted that the gain of the proposed converter is much higher than that of the conventional converter. Due to this, the turns ratio of the FB transformer in the proposed converter can be designed with a smaller value than that in the conventional converter with same duty ratio so that it is possible that both primary-conduction loss and secondary-voltage stress are significantly reduced. For simply analyzing the voltage conversion ratio, we ignored the effect of resonant capacitor Cr . However, with considering the resonant capacitor, the turns ratio of T1 can be slightly changed. As explained in the previous section, the voltage of resonant capacitor appears on Vrec (t), as shown in Fig. 7(a), and the voltage conversion ratio is slightly increased in practice. Consequently, the turns ratio of T1 will be designed with smaller value than the value designed without considering the effect of Cr . D. ZVS Range and Duty-Cycle Loss As described in Section III-B, the ZVS operation of leadingleg switches is easily achieved over whole output conditions due to the sufficient energy stored in the output inductor. However, since the ZVS operation of lagging-leg switches is carried out by the resonance of Llk1 , Llk2 , and 2Coss , it could not be well achieved in the case that the energy stored in the two leakage inductors is small. Adding large resonant inductors in series with transformers (or increasing the two leakage inductances) can extend the ZVS range, but it is widely known that it reduces effective duty cycle (or increases dutycycle loss). To compensate for this, the turns ratio of the transformer should be designed to be larger, thereby increasing both primary-conduction loss and secondary-voltage stress.

In order to extend the ZVS range of lagging-leg switches, the proposed converter uses only magnetizing current of T2 , i.e., imag_T 2 (t), as described in Fig. 9. As shown in Fig. 9, the current ripple of ΔIR obtained from small magnetizing inductance of T2 increases the ZVS energy. Thus, it is possible that ZVS operation of lagging-leg switches is well achieved during the battery recharging process in Fig. 3. Since the magnetizing inductor of T2 is in parallel with powering path, the effect of duty-cycle loss is minimized. In addition, although the small magnetizing inductance of T2 is used, RMS current stress in the primary side never increases over that of the conventional ZVS FB converter, according to the principle in [24]–[27]. The RMS current stress will be described in the next section in detail. E. Primary-Conduction Loss There exists circulating current in the ZVS FB converter in Fig. 1, which flows through a transformer and switches although the primary voltage of the transformer remains zero. This current greatly increases primary-conduction loss in constant-input and variable-output applications such as EV battery chargers and RF power generator. On the other hand, in the proposed converter, the primary current of the FB transformer is reset during mode 3 and is maintained at zero, whereas the primary voltage in the FB side remains zero. That is, it can be said that there is no circulating current on the FB side of the proposed converter during battery recharging process. As mentioned in Section III-B, the turns ratio of FB transformer is designed with smaller value than the ZVS FB converter in Fig. 1 due to higher gain. Smaller turns ratio significantly reduces the magnitude of battery recharging current reflected to the primary of the FB side. Thus, the current stresses of leading-leg switches (Q1 and Q2 ) and FB transformer (T1 ) are greatly reduced. In the proposed converter, magnetizing inductance of the HB transformer is designed to be small for extending the ZVS energy of lagging-leg switches. However, the RMS current stress of leading-leg switches and the FB transformer never increases because the magnetizing current of the HB transformer never flows through Q1 , Q2 , and T1 . This is in contrast to the fact that, in conventional ZVS FB converters, using small magnetizing inductance for wider ZVS range increases the current stress of

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Fig. 10. Equivalent circuit of rectifier stage in the case that: (a) both FB and HB converters transfer the power; and (b) only HB converter transfers the power.

all switches and transformers [26], [27]. In addition, in spite of using small magnetizing inductance, the RMS current stresses of lagging-leg switches (Q3 and Q4 ) and HB transformer (T2 ) become smaller than those of the ZVS FB converter in Fig. 1 because the average value of magnetizing current of T2 is zero within a half-switching period, and its contribution to RMS current stress is negligible at medium-to-high power conditions [24]–[27]. In summary, due to no circulating current, smaller turns ratio, and the same principle in [24]–[27], the RMS current stresses of all switches and transformers are smaller than those of the conventional ZVS FB converter so that it is possible that primary-conduction loss is significantly reduced. This will be verified with the experimental results in the final section.

Fig. 11.

Normalized voltage stress of rectifier diodes.

Fig. 12.

Output filter inductance required for IPP = 500 mA.

F. Secondary-Voltage Stress Fig. 10 shows equivalent circuits of rectifier stage during two powering modes in order to get the voltage stress of rectifier diodes in the proposed converter. From the equivalent circuits, the voltage stress of diodes can be approximately obtained considering voltage spike caused by parasitic circuit elements as follows: 2+α VO_ max = k1 VO_ max Vstress(D1,2,5,6) = 0.9 + 0.5α 1 Vstress(D3,4) = VO_ max = k2 VO_ max (13) 0.9 + 0.5α

TABLE I C OMPONENTS L IST

where 0 < α(= n2 /n1 ) < 1. To get the voltage stress of diodes in a conventional ZVS FB converter, let us first assume that the maximum value of effective duty ratio is 45%. In turn, by considering the voltage spike caused by parasitic circuit elements and the battery recharging profile in Fig. 3, the voltage stress can be obtained as Vstress(D1,2,3,4) = 2.2VO_ max .

(14)

Equations (13) and (14) can be normalized by VO , and Fig. 11 shows the normalized voltage stresses. From the figure, it is noted that the proposed converter requires four high-

voltage-rated diodes and two low-voltage-rated diodes, whereas conventional ZVS FB converter needs only high-voltage-rated diodes. Note that low-voltage diodes have advantages such as much lower ON-state voltage and better recovery characteristics compared with high-voltage diodes.

LEE AND MOON: HB ZVS FB CONVERTER WITH REDUCED CONDUCTION LOSS FOR EV BATTERY CHARGERS

Fig. 13.

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Operating waveforms (ip1 (t), ip2 (t), and Vp1 (t)) of the proposed converter: (a) when VO = 250 V, (b) when VO = 350 V, and (c) when VO = 450 V.

Fig. 14. Operating waveforms (Vp1 (t) and Vrec (t)) of the proposed converter: (a) when VO = 250 V, (b) when VO = 350 V, and (c) when VO = 450 V.

The conventional ZVS FB converter uses the four highvoltage diodes during entire modes of the battery recharging process in Fig. 3. On the other hand, in the proposed converter, the two low-voltage diodes of D3 and D4 , which feature good performance, are mainly used to transfer the power required in the output port while the section of HB powering is increasing over the battery recharging process, particularly during constant-current mode or constant-voltage mode with lightload conditions. Due to this operation, the conduction loss of rectifier stage can be greatly reduced and the conversion efficiency can be further improved. G. Filter Requirement The output filter inductor’s values can be derived from the rectifier output voltages in Fig. 6 using (10), (11), and (15) as (16) and (17) V = LO

IPP ΔT

(15)

where IPP is the peak-to-peak value of current flowing through LO , and V is the voltage applied to LO during the time of ΔT . For the proposed converter LO_proposed =

VO TS (1 + 0.5α − Mnorm (D)) IPP 0.5α × 1− . Mnorm (D)

(16)

Fig. 15. Output filter inductor: (a) for the conventional ZVS FB converter (1.3 mH, CM400060) and (b) for the proposed converter (700 μH, CM343060).

For the conventional ZVS FB converter LO_Conventional =

VO TS (1 − Mnorm (D)) . 2IPP

(17)

For a quantitative analysis, these two parameters are used. • IPP = 500 mA. • Switching frequency fS = 100 kHz. Since in EV battery chargers, the current ripple of LO is worst when the battery voltage is lowest, VO is given as 250 V in Fig. 3.

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Fig. 16. ZVS waveforms of lagging-leg switches: (a) when VO = 250 V, (b) when VO = 350 V, and (c) when VO = 450 V.

Fig. 17. ZVS waveforms of leading-leg switches: (a) when VO = 250 V, (b) when VO = 350 V, and (c) when VO = 450 V.

Fig. 12 shows the required filter inductance in function of normalized voltage conversion ratio. With Figs. 8 and 12, it is known that the output inductor’s value of the proposed converter is much smaller than that of the conventional converter. The use of the output filter inductor with smaller inductance is very beneficial in terms of size and weight as EV battery chargers. H. Transformer Utilization T.U. In [20], the sections of FB and HB powering are separated over the switching period, which lower transformer utilization for EV battery chargers. On the other hand, in the proposed converter, HB part always helps powering of the FB part over the entire switching period, as shown in Fig. 7(a). This improves transformer utilization over the battery recharging profile in Fig. 3. For example, maximum HB powering occurs at the beginning of recharging (A) in Fig. 3 and can be simply estimated as 250 V × IBattery . As the battery voltage increases, FB powering will gradually increase, but it always receives the help of HB powering with 250 V × IBattery . Thus, maximum FB powering occurs at the end of constant-current charging (B) in Fig. 3 and can be calculated as 200 V × IBattery . Then, the transformer’s utilization, T.U. can be calculated as follows: T.U. = =

Peﬀective_power PActually_designed_power 450 V · IBattery = 1.0. (18) 250 V · IBattery + 200 V · IBattery

Consequently, it is known that the proposed converter has better T.U. than the topology in [20] consisting of FB and HB converters. IV. E XPERIMENTAL R ESULTS In order to verify the performance of the proposed converter as EV battery chargers, it is realized with the battery charger specification given below and the battery recharging profile in Fig. 3: • input voltage VIN = 400 V; • maximum output power PO(max) = 2 kW; • switching frequency fS = 100 kHz. Considering the fact that the power-handling capacity of the HB converter is generally smaller than that of the FB converter, the HB transformer was designed to be able to process only about 33% of the total power by adjusting the turns ratio. A prototype converter was built using the components listed in Table I, and the magnetizing inductance of the HB transformer T2 was set as 100 μH for extending the ZVS range of laggingleg switches. Figs. 13 and 14 show key waveforms of the proposed converter during constant-current mode in Fig. 3. As shown in the figures, all measured waveforms are well following the theoretical waveforms described in Fig. 5 and the primary current of HB inverter ip2 (t) exactly corresponds to Fig. 9. We can also see that the circulating current existing in the conventional ZVS FB converter is removed in the proposed converter from ip1 (t) in Fig. 13. In addition, due to the staircase output voltage

LEE AND MOON: HB ZVS FB CONVERTER WITH REDUCED CONDUCTION LOSS FOR EV BATTERY CHARGERS

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Fig. 19. Efficiency over the battery charging profile in Fig. 3: (a) efficiency during constant-current mode and (b) efficiency during constant-voltage mode.

Fig. 19 shows efficiency values measured during battery recharging process. As shown in the figures, the efficiency of the proposed converter is advanced over that of the conventional ZVS FB converter during constant-current mode. This is the result of reduced primary-conduction loss coming from no circulating current, lower turns ratio, and the same principle in [24]–[27]. The efficiency during constant-voltage mode is also over the conventional converter, which results from wider ZVS range. Fig. 18. RMS current stress measured during constant-current charging mode: (a) lagging-leg switches, (b) leading-leg switches, and (c) transformers.

V. C ONCLUSION waveform of the rectifier in Fig. 14, the output filter inductor’ size becomes smaller than that of the conventional ZVS FB converter, as shown in Fig. 15. Figs. 16 and 17 show ZVS waveforms of switches during constant-current charging mode. From Figs. 16 and 17, it is clear that all switches in the proposed converter are always turned ON with ZVS during the battery recharging process. Fig. 18 shows RMS current stress measured during constantcurrent charging mode in Fig. 3. It is verified that, as explained in Section III-E, the proposed converter has much smaller RMS current stress in the primary side in spite of wider ZVS range compared with the conventional ZVS FB converter.

In order to extend the driving range in EV mode, the power rating of battery chargers in EVs or PHEVs will continue to increase. As a result, research on high-power high-efficiency topologies is necessarily required. In this paper, a new softswitching dc–dc converter for EV battery on-board chargers has been proposed. From the analysis results, it is worth noting that the proposed converter has following advantages: 1) reduction of primary- and secondary-conduction losses; 2) no circulating current, less duty-cycle loss, and full ZVS capability; 3) smaller-sized output filter inductor.

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Due to these advantages, it is possible for the proposed converter to have smaller power loss and higher efficiency compared with conventional ZVS FB converters as battery chargers. From the experiment results of the proposed converter realized with a scale-downed 2-kW battery charger, all aforementioned advantages were verified and its effectiveness and feasibility were confirmed. Therefore, it can be said that the proposed converter is suitable for the dc–dc converter in high-power battery chargers for EVs or PHEVs. R EFERENCES [1] I. Aharon and A. Kuperman, “Topological overview of powertrains for battery-powered vehicles with range extenders,” IEEE Trans. Power Electron., vol. 26, no. 3, pp. 868–876, Mar. 2011. [2] J. Cao and A. Emadi, “A new battery/ultracapacitor hybrid energy storage system for electric, hybrid, and plug-in hybrid electric vehicles,” IEEE Trans. Power Electron., vol. 27, no. 1, pp. 122–132, Jan. 2012. [3] M. B. Camara, H. Gualous, F. Gustin, A. Berthon, and B. Dakyo, “DC/DC converter design for supercapacitor and battery power management in hybrid vehicle applications-polynomial control strategy,” IEEE Trans. Ind. Electron., vol. 57, no. 2, pp. 587–597, Feb. 2010. [4] M. Pahlevaninezhad, P. Das, J. Drobnik, G. Moschopoulos, P. Jain, and A. Bakhshai, “A nonlinear optimal control approach based on the controlLyapunov function for an AC/DC converter used in electric vehicles,” IEEE Trans. Ind. Inform., vol. 8, no. 3, pp. 596–612, Aug. 2012. [5] M. Yilmaz and P. T. Krein, “Review of charging power levels and infrastructure for plug-in electric and hybrid vehicle,” in Proc. IEVC, 2012, pp. 1–8. [6] M. Pahlevaninezhad, P. Das, J. Drobnik, P. Jain, and A. Bakhshai, “A novel ZVZCS full-bridge DC/DC converter used for electric vehicle,” IEEE Trans. Power Electron., vol. 27, no. 6, pp. 2752–2769, Jun. 2012. [7] B. Gu, J. Lai, N. Kees, and C. Zheng, “Hybrid-switching full-bridge DC–DC converter with minimal voltage stress of bridge rectifier, reduced circulating losses and filter requirement for electric vehicle battery chargers,” IEEE Trans. Power Electron., vol. 28, no. 3, pp. 1132–11441, Mar. 2013. [8] Y. Cho and J. S. Lai, “Digital plug-in repetitive controller for single-phase bridgeless PFC converters,” IEEE Trans. Power Electron., vol. 28, no. 1, pp. 165–175, Jan. 2013. [9] M. Pahlevaninezhad, P. Das, J. Drobnik, P. Jain, and A. Bakhshai, “A new control approach based on the differential flatness theory for an AC/DC converter used in electric vehicles,” IEEE Trans. Power Electron., vol. 27, no. 4, pp. 2085–2103, Apr. 2012. [10] F. Musavi, W. Eberle, and W. G. Dunford, “A high-performance singlephase bridgeless interleaved PFC converter for plug-in hybrid electric vehicle battery chargers,” IEEE Trans. Ind. Appl., vol. 47, no. 4, pp. 1833– 1843, Jul. 2011. [11] F. Musavi, M. Edington, W. Eberle, and W. G. Dunford, “Evaluation and efficiency comparison of front end AC–DC plug-in hybrid charger topologies,” IEEE Trans. Smart Grid., vol. 3, no. 1, pp. 413–421, Mar. 2012. [12] D. Zhang, F. Wang, R. Burgos, and D. Boroyevich, “Total flux minimization control for integrated inter-phase inductors in paralleled, interleaved three-phase two-level voltage-source converters with discontinuous space-vector modulation,” IEEE Trans. Power Electron., vol. 27, no. 4, pp. 1679–1688, Apr. 2012. [13] M. Pahlevaninezhad, P. Das, J. Drobnik, P. Jain, and A. Bakhshai, “A ZVS interleaved boost AC/DC converter used in plug-in electric vehicles,” IEEE Trans. Power Electron., vol. 27, no. 8, pp. 3513–3529, Aug. 2012. [14] J. A. A. Qahouq, L. Huang, and D. Huard, “Efficiency-based auto-tuning of current sensing and sharing loops in multiphase converters,” IEEE Trans. Power Electron., vol. 23, no. 2, pp. 1009–1013, Mar. 2008. [15] J. Zhang, F. Zhang, X. Xie, D. Jiao, and Z. Qian, “A novel ZVS DC/DC converter for high power applications,” IEEE Trans. Power Electron., vol. 19, no. 2, pp. 420–429, Mar. 2004. [16] T. Kim, S. Lee, and W. Choi, “Design and control of the phase shift full bridge converter for the on-board battery charger of electric forklifts,” J. Power Electron., vol. 12, no. 1, pp. 113–119, Jan. 2012.

[17] B. P. McGrath, D. G. Holmes, P. J. McGoldrick, and A. D. McIver, “Design of a soft-switched 6-kW battery charger for traction applications,” IEEE Trans. Power Electron., vol. 22, no. 4, pp. 1136–1144, Jul. 2007. [18] D. Gautam, F. Musavi, M. Edington, W. Eberle, and W. G. Dunford, “An automotive on-board 3.3 kW battery charger for PHEV application,” IEEE Trans. Veh. Technol., vol. 61, no. 8, pp. 3466–3474, Oct. 2012. [19] C. Liu, B. Gu, J. Lai, M. Wang, Y. Ji, G. Cai, Z. Zhao, C. Chen, C. Zheng, and P. Sue, “High-efficiency hybrid full-bridge-half-bridge converter with shared ZVS lagging leg and dual outputs in series,” IEEE Trans. Power Electron., vol. 28, no. 2, pp. 849–861, Feb. 2013. [20] W. Yu, J. Lai, W. Lai, and H. Wan, “Hybrid resonant and PWM converter with high efficiency and full soft-switching range,” IEEE Trans. Power Electron., vol. 27, no. 12, pp. 4925–4933, Dec. 2012. [21] B. Gu, C. Lin, B. Chen, J. Dominic, and J. Lai, “Zero-voltage-switching PWM resonant full-bridge converter with minimized circulating losses and minimal voltage stresses of bridge rectifiers for electric vehicle battery chargers,” IEEE Trans. Power Electron., vol. 28, no. 10, pp. 4657–4667, Oct. 2013. [22] E. Inoa and J. Wang, “PHEV charging strategies for maximized energy saving,” IEEE Trans. Veh. Technol., vol. 60, no. 7, pp. 2978–2986, Sep. 2011. [23] B.-Y. Chen and Y.-S. Lai, “New digital-controlled technique for battery charger with constant current and voltage control without current feedback,” IEEE Trans. Ind. Electron., vol. 59, no. 3, pp. 1545–1553, Mar. 2012. [24] M. Borage, S. Tiwari, S. Bhardwaj, and S. Kotaiah, “A full-bridge DC–DC converter with zero-voltage-switching over the entire conversion range,” IEEE Trans. Power Electron., vol. 23, no. 4, pp. 1743–1750, Jul. 2008. [25] M. Borage, S. Tiwari, and S. Kotaiah, “A passive auxiliary circuit achieves zero-voltage-switching in full-bridge converter over entire conversion range,” IEEE Trans. Power Electron., vol. 3, no. 4, pp. 141–143, Dec. 2005. [26] I. Lee and G. W. Moon, “Soft-switching DC/DC converter with a full ZVS range and reduced output filter for high-voltage applications,” IEEE Trans. Power Electron., vol. 28, no. 1, pp. 112–122, Jan. 2013. [27] I. Lee, S. Y. Cho, and G. W. Moon, “Improved phase-shift PWM converters for larger-sized PDP slim sustain power module,” IEEE Trans. Power Electron., vol. 28, no. 2, pp. 945–958, Feb. 2013.

Il-Oun Lee (S’10–M’13) received the B.S. degree in electrical and electronic engineering from Kyungpook National University, Daegu, Korea, in 2000; the M.S. degree in electrical engineering from Seoul National University, Seoul, Korea, in 2002; and the Ph.D. degree from Korea Advanced Institute of Science and Technology, Daejeon, Korea, in 2013. He was a Research Engineer in the Plasma Display Panel Development Group, Samsung SDI, Yongin, Korea, for five years beginning in 2003. Since 2008, he has been a Senior Engineer with the Power Advanced Development Group, Samsung Electro-Mechanics Company, Ltd., Suwon, Korea. His current research interests include dc–dc converters, powerfactor-correction ac–dc converters, LED driver, battery charger for electric vehicle, digital display power systems, and digital control approach of dc–dc converters.

Gun-Woo Moon (S’92–M’00) received the M.S. and Ph.D. degrees in electrical engineering from Korea Advanced Institute of Science and Technology (KAIST), Daejeon, Korea, in 1992 and 1996, respectively. He is currently a Professor with the Department of Electrical Engineering, KAIST. His research interests include modeling, design, and control of power converters; soft-switching power converters; resonant inverters; distributed power systems; power factor correction; electric drive systems; driver circuits of plasma display panels; and flexible ac transmission systems.