High-Power-Factor Electronic Ballast With Constant DC ... - IEEE Xplore

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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 13, NO. 6, NOVEMBER 1998

High-Power-Factor Electronic Ballast with Constant DC-Link Voltage Ricardo de Oliveira Brioschi and José Luiz F. Vieira

Abstract—This paper presents a high-power-factor (HPF) electronic ballast based on a single power processing stage with constant dc-link voltage. The switching frequency is controlled to maintain the dc-link voltage and the voltage across the switches constant, independently of changes in the ac-input voltage. This control method assures zero-voltage switching (ZVS) for the specified ac-input-voltage range. Besides, with an appropriate design of the fluorescent lamps’ drive circuit, the lamps’ power can be kept close to the rated value. The power-factor-correction (PFC) stage is formed by a boost converter operating in discontinuous conduction mode, which naturally provides HPF to the utility line. The fluorescent lamps are driven by an unmodulated sinewave current generated from an LC parallel resonant converter, which operates above the resonant frequency to perform ZVS. Theoretical analysis and experimental results are presented for two series-connected 40-W fluorescent lamps operating from 127 V 015% to +10% 60-Hz utility line. The switching frequency is changed from 25 to 45 kHz to maintain the dc-link voltage regulated at 410 V, which leads to a constant output power. The experimental results confirm the high efficiency and HPF of this electronic ballast. Index Terms—Electronic ballast, high power factor, soft switching.

I. INTRODUCTION

F

LUORESCENT lamps are usually preferred to replace the incandescent lamps because they inherently have a longer lifetime and yield higher efficacy [1]. However, these lamps require high striking voltage during starting and current limiting control after starting because they have negative impedance characteristics. Traditional magnetic ballasts, operating at line frequency, have been used to solve these problems. In spite of their low cost, these ballasts present flickering, high size and weight, and hum and stroboscopic effects [2]–[4]. When operating at high frequency, fluorescent lamps present the following characteristics [1]–[4]: the luminous efficacy increases by about 10%, which reduces the energy consumption, flickering as well as stroboscopic effects can be eliminated, and the audible noise falls to unnoticeable levels. To obtain these benefits, as well as smaller size and weight, electronic ballasts are used instead of magnetic ballasts. The preferred method to drive the fluorescent lamp is with an unmodulated sine-wave current with a minimal ripple for the operating content. The current crest factor Manuscript received April 15, 1997; revised March 23, 1998. Recommended by Associate Editor, K. D. T. Ngo. The authors are with the Departamento de Engenharia Elétrica, Universidade Federal do Esp´ırito Santo, 29060-970, Vitória, ES, Brazil (e-mail: [email protected]). Publisher Item Identifier S 0885-8993(98)08226-X.

condition should be as low as possible, not exceeding 1.7 parallel resonant converter has been attractive [1]. The for this application because it ensures a sine-wave current for the lamp with low crest factor as well as establishes an appropriate voltage during the ignition process and maintains a steady-state-rated current. A compact high-power-factor (HPF) electronic ballast inparallel resonant converter can be obtained corporating an when high frequency is used. However, at high frequency, soft commutation techniques are recommended to achieve high efficiency [5], [6]. The utility line can be more efficiently utilized when HPF with low total harmonic distortion (THD) is achieved. The advantages of the HPF with low THD include reduction in the rms line current and in the line current harmonic distortion [7]. HPF can be obtained using two power processing stages. The first one is an HPF preregulator stage, which converts the ac-input voltage to a dc voltage. The second stage transforms the dc voltage to a high-frequency ac voltage to drive the fluorescent lamps. Active power factor correction (PFC) can be performed by a boost preregulator operating in continuous conduction mode, where the inductor boost current must follow a sinusoidal reference waveform. This method provides nearly unity power factor with THD less than 5% [2], [4], [8]. When a boost converter operates as a preregulator stage in discontinuous conduction mode, the input current follows naturally the sinusoidal waveform of the input voltage, providing HPF to the utility line [9]. Two power processing stages increase the final cost, besides reducing the electronic ballast reliability. An interesting option to avoid these problems are the HPF electronic ballasts based on a single power processing stage [5]–[7], [10]–[15]. This paper presents an HPF electronic ballast based on a single power processing stage in which a boost converter operates in discontinuous conduction mode [16]. Besides that, by controlling the switching frequency, this ballast maintains parallel constant the dc-link voltage and output power. An resonant converter is used to drive the fluorescent lamps by an unmodulated sine-wave current. To obtain zero-voltage switching (ZVS), the electronic ballast operates above the resonant frequency. This electronic ballast has been designed to drive two series-connected 40-W fluorescent lamps operating from 127 V 15% to 10% 60-Hz utility line. To maintain the dclink voltage regulated in 410 V, the switching frequency is changed from 25 to 45 kHz.

0885–8993/98$10.00  1998 IEEE

DE OLIVEIRA BRIOSCHI AND VIEIRA: BALLAST WITH CONSTANT DC-LINK VOLTAGE

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(a)

(b) Fig. 1. Electronic ballast diagram: (a) power stage and (b) control circuit.

II. CIRCUIT DESCRIPTION In order to obtain a simple HPF electronic ballast, the PFC stage and the inverter output stage are combined in a single stage. This is accomplished by allowing both stages to share the two switches of the electronic ballast. The dc-link voltage and, consequently, switch voltages are kept at a fixed value, independently of the ac-line-voltage variations. This is obtained by controlling the electronic ballast switching frequency. By employing this control method, ZVS can be ensured for the specified ac-input-voltage range. In addition, with an appropriate design of the fluorescent lamps’ drive circuit, the lamps’ power can be maintained close to the rated value. The complete diagram of this electronic ballast is shown in Fig. 1. Fig. 1(a) shows the power stage diagram. As can be seen, the PFC stage is formed by a boost converter, which operates in discontinuous conduction mode. The fluorescent parallel resonant stage, which lamps are driven by an operates above the resonant frequency to perform ZVS. The control circuit, shown in Fig. 1(b), is based on a pulsewidth modulation (PWM) regulator IC (UC 3525). The ballast switching frequency is adjusted by altering the equivalent resistance of the internal oscillator circuit of the UC 3525 IC.

Fig. 2. Two-switch boost converter.

III. PRINCIPLE

OF

OPERATION

This electronic ballast can be viewed as being composed of two simplified independent converters. The first one is obparallel resonant converter is considered tained when the . The resulting converter is the twoas a resistive load to switch boost preregulator stage shown in Fig. 2, in which the main waveforms are shown in Fig. 3. The second one is large enough to be is obtained when the capacitance considered a voltage source, and the dc blocking capacitor is also large enough, resulting in a negligible impedance at the switching frequency. Hence, the boost converter can

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Fig. 5. Main waveforms of the

LC parallel resonant converter. (3)

Fig. 3. Main waveforms of the two-switch boost converter.

The instantaneous mean value of the boost inductor current, considering that the rectified input voltage remains constant in a switching period, is given by (4)

Fig. 4. Simplified

LC

Due to the high-frequency input filter, the ac-line current should be given by the instantaneous mean value of the boost inductor current, according to the following [9]:

parallel resonant converter.

be replaced by the voltage source . This results in an parallel resonant converter, shown in Fig. 4, for which the main waveforms are shown in Fig. 5. This converter has been widely used in electronic ballasts.

(5) where

IV. RELEVANT ANALYSIS

(6)

The relevant characteristics of the electronic ballast are defined by the input current, power factor, and THD. The main parameters to be determined are the boost inductance and resonant parameters.

(7) ac-peak voltage; dc-link voltage.

A. Input Current The inductor boost peak current during a switching period, obtained from Fig. 3, is given by (1) The linear increasing and decreasing times are given by the following: (2)

B. Input Power The input power is obtained from the following: (8) where (9)


Fig. 6. Power factor as a function of

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. Fig. 8. Normalized output characteristics of the boost stage as a function of fs =fs min .

E. Normalized Output Characteristics of the Boost Stage The instantaneous mean value of current Fig. 3, is given by

, obtained from

(14) Integrating (14) in a rectified input-voltage cycle results in (15) Fig. 7. THD as a function of

.

With the substitution of (14) into (15), one obtains (16)

From (5) to (9), the following expression results: (10)

where (17)

C. Power Factor (18)

The power factor is defined by minimum switching frequency. (11)

PF

Considering that the input voltage does not have harmonic components, the power factor can be given by

The normalized output characteristics of the boost stage as a function of the switching frequency are shown in Fig. 8. As can be seen, by controlling the switching frequency, the can be kept constant, independently of the dc-link voltage ac-input-voltage variations.

(12)

PF

F. Resonant Parameters The proposed electronic ballast power factor as a function of , given by (12), is shown in Fig. 6. D. Total Harmonic Distortion

parallel resonant circuit, shown in Fig. 4, is a The second-order low-pass filter, which can be described by the following [2], [17]: undamped natural frequency

(19)

Considering unit displacement factor, the THD can be defined by characteristic impedance THD

PF

The THD as a function of

PF is shown in Fig. 7.

(20)

(13) quality factor at the natural frequency

(21)

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Fig. 9. The jZi j=Zo characteristics as a function of fs =fo , with QL as a parameter.

for

Fig. 10. Low-frequency voltage and current: input voltage vac (50 V/div) and input current iac (0.5 A/div); time scale: 2 ms/div.

resonant frequency. (22)

parallel circuit is fed by a high-frequency squareThe of magnitude . Its fundamental wave voltage source component, obtained from the Fourier analysis, is given by (23) parallel resonant circuit can be designed to operate The , selected as at the undamped natural frequency the middle point of the operating frequency range. At this frequency, the fundamental component amplitudes of voltage and current through are, respectively [2] across (24) (25) When the fluorescent lamps are off, they can be considered as an open circuit. Therefore, the quality factor at startup is very high. As shown by (24), the voltage across the lamps will be high enough for striking them. At steady state, the parallel circuit operates above the resonant frequency , providing ZVS. The lamps’ power, obtained from (25), is (26) G. Input Impedance of the LC Parallel Resonant Circuit The input impedance of the given by [2]

Fig. 11. Rectified ac-input voltage vin (100 V/div) and the boost inductor current iLb (1 A/div); time scale: 2 ms/div.

where (29)

(30) characteristics as a function of Fig. 9 shows the , with as a parameter. As can be seen, for values , as well close to 1.0, the normalized input impedance parallel resonant circuit, do not as the input current of the . significantly change with

parallel resonant circuit is

V. DESIGN PROCEDURE A electronic ballast prototype has been built to meet the following specifications.

(27) A. Input Data which can be rewritten as (28)

ac-input voltage: 127 V rms 15% to 10% 60 Hz; W; output power: 45 kHz; switching frequency: 25 kHz A; fluorescent lamp-rated current:


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(a) Fig. 12.

(b)

MOSFET’s commutations: (a) vM 1 (100 V/div) and

iM 1

(0.5 A/div). (b)

Fig. 13. Resonant current iLr (0.5 A/div) and the high-frequency voltage: vM 1 (100 V/div); time scale: 5 s/div.

fluorescent lamp ignition voltage: %. efficiency:

Parameter, Power Factor, and THD

V) that also For the lowest input voltage ( results in the smallest value for the switching frequency kHz), a value of is obtained from (6). ( and THD From Figs. 6 and 7, we have FP %. D. Boost Inductance From Fig. 8, for a normalized current kHz, from (17), for V, and mH is obtained.

The lamps’ equivalent resistance is . From (26), results. For

iM 2

(1 A/div); time scale: 5

s/div.

Fig. 14. Resonant current iLr (0.5 A/div) and fluorescent lamps’ voltage: vFL (100 V/div); time scale: 5 s/div.

kHz (mean operation frequency) and using (19) and (20), one nF and mH. Equations (21) and obtains and kHz. (22) result in

As the boost stage must operate in discontinuous conduction has to be larger than twice the maximum mode, the voltage V. In this case, V ac-peak voltage has been selected.

E. Resonant Parameters

(100 V/div) and

V;

B. Selection of the DC-Link Voltage

C.

vM 2

, and, A,

VI. EXPERIMENTAL RESULTS An electronic ballast prototype has been built to meet the input data specifications, whose parameters and components are: mH, 140 turns on core EE 30/14, IP6—Thornton; mH, 112 turns on core EE 30/7, IP6—Thornton; mH, 60 turns on core EE 20/10, IP6—Thornton; Pulse transformer: 15/15/15 turns on core EE 20/10, IP6—Thornton; – , 1N4004; Input diode bridge: : SK3GF04 (Semikron); Fast diode, : IRF 840 (international rectifier); nF/630 V. Experimental waveforms have been obtained for V, A, kHz, and V. The ac-input current and voltage, which demonstrate the HPF of this electronic ballast, are shown in Fig. 10. The rectified input and the boost inductor current, presenting a 120-Hz voltage envelope, are shown in Fig. 11.

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ELECTRONIC INPUT

Fig. 15.

TABLE I OUTPUT CHARACTERISTICS

AND

Output power versus input voltage.

Fig. 17. Switching frequency versus input voltage.

efficiency, and Fig. 17 exhibits the switching frequency, also as a function of the input voltage. This electronic ballast exhibits a power factor larger than 0.99 and a THD lower than 10% for the entire inputvoltage range. VII. CONCLUSION

Fig. 16.

Efficiency versus input voltage.

The MOSFET’s commutations showing the ZVS can be seen in Fig. 12, whereas Fig. 13 shows the resonant current and the high-frequency voltage . The fluorescent and the resonant current are shown lamps’ voltage in Fig. 14. The electronic ballast input and output characteristics, obV, are presented in Table I, from which tained for Figs. 15–17 have been obtained. The output power versus the input voltage is shown in Fig. 15. As can be seen, the output power is kept close to the rated value (72 W) within 3% deviation for the entire input-voltage range. Fig. 16 shows the

An HPF electronic ballast, based on a single power processing stage, with constant dc-link voltage was introduced in this paper. By controlling the switching frequency, dclink voltage and voltage across the switches are maintained constant, independently of changes in the ac-input voltage. This control method ensures ZVS for the specified ac-inputvoltage range. Besides, with an appropriate design of the resonant converter, the lamps’ power is kept close to the rated value. A simple HPF electronic ballast has been obtained when the PFC stage and the inverter output stage were combined in a single stage. This was accomplished by allowing both stages to share the two-switch of the electronic ballast. The PFC stage is formed by a boost converter operating in discontinuous conduction mode. This operation mode ensures that the input current naturally follows the sinusoidal waveform of the input voltage. The fluorescent lamps are driven by an unmodulated parallel resonant sine-wave current generated from an converter, which operates above the resonant frequency to perform ZVS. Theoretical analysis and experimental results are presented for two series-connected 40-W fluorescent lamps


operating from 127 V 15% to 10% 60-Hz utility line. The switching frequency is changed from 25 to 45 kHz to maintain the dc-link voltage regulated at 410 V, which leads to a constant output power. The experimental results confirm the high efficiency and the HPF of this electronic ballast. ACKNOWLEDGMENT The authors would like to express their gratitude to “Thornton Inpec Eletrônica Ltda” for contributing the magnetic core for this project.

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[12] C. Licitra, L. Malesani, G. Spiazzi, P. Tenti, and A. Testa, “Single-ended soft-switching electronic ballast with unit power factor,” in IEEE-APEC Proc., 1991, pp. 953–957. [13] E. Deng and S. Cuk, “Single switch, unit power factor, lamp ballasts,” in IEEE-APEC Proc., 1995, pp. 670–676. [14] C. Blanco, M. Alonso, E. López, A. Calleja, and M. Rico, “A single stage fluorescent lamp ballast with high power factor,” in IEEE-APEC Proc., 1996, pp. 616–621. [15] W. C. F. C. Lee and T. Yamauchi, “An improved charge pump electronic ballast with low THD and low crest factor,” in IEEE-APEC Proc., 1996, pp. 622–627. [16] P. N. Wood, “Electronic ballasts using the cost-saving IR2155 driver,” Application Notes AN-995, International Rectifier, 1994. [17] T. H. Yu, H. M. Huang, and T. F. Wu, “Self excited half-bridge series resonance parallel loaded fluorescent lamp electronic ballast,” in IEEE-APEC Proc., 1989, pp. 657–664.

REFERENCES [1] E. E. Hammer and T. K. McGowan, “Characteristics of various F40 fluorescent systems at 60 Hz and high frequency,” IEEE Trans. Ind. Applicat., vol. IA-21, no. 1, pp. 11–16, 1985. [2] M. K. Kazimierczuk and W. Szaraniec, “Electronic ballast for fluorescent lamps,” IEEE Trans. Power Electron., vol. 8, no. 4, pp. 386–395, 1993. [3] E. C. Nho, K. H. Jee, and G. H. Cho, “New soft-switching for high efficiency electronic ballast with simple structure,” Int. J. Electron., vol. 71, no. 3, pp. 529–542, 1991. [4] L. Laskai and I. J. Pitel, “Discharge lamp ballasting,” in IEEE-PESC’95, Atlanta, GA, 1995. [5] J. L. F. Vieira, Márcio A. Có, and L. D. Zorzal, “High power factor electronic ballast based on a single power processing stage,” in IEEEPESC Proc., 1995, pp. 687–693. [6] M. A. Có, D. S. L. Simonetti, and J. L. F. Vieira, “High power factor electronic ballast operating at critical conduction mode,” in IEEE-PESC Proc., 1996, pp. 962–968. [7] E. Deng and S. Cuk, “Single stage, high power factor, lamp ballast,” in IEEE-APEC Proc., 1994, pp. 441–449. [8] J. Spangler and A. K. Behera, “Power factor correction used for fluorescent lamp ballast,” in IEEE-IAS Proc., 1991, pp. 1836–1841. [9] K.-H. Liu and Y.-L. Lin, “Current waveform distortion in power factor correction circuits employing discontinuous-mode boost converters,” in IEEE-PESC Proc., 1989, pp. 825–829. [10] I. Takahashi, “Power factor improvement of a diode rectifier circuit,” in IEEE-IAS Annu. Meeting Proc., 1990, pp. 1289–1294. [11] L. Laskai, P. Enjeti, and I. J. Pitel, “A unity power factor electronic ballast for metal halide lamps,” in IEEE-APEC Proc., 1994, pp. 31–37.

Ricardo de Oliveira Brioschi was born in Muniz Freire-ES, Brazil, in 1968. He received the B.E. and M.S. degrees in electrical engineering in 1991 and 1997 from the Federal University of Esp´ırito Santo, Vitória, Brazil. Since 1992, he has been a Professor at the Federal Technique School of Esp´ırito Santo. His research interests include lighting systems and switchingmode power supply.

José Luiz F. Vieira was born in Muqui-ES, Brazil, in 1958. He received the B.E. degree from the Federal University of Esp´ırito Santo, Brazil, in 1981, the M.S. degree from the Federal University of Rio de Janeiro, Brazil, in 1986, and the Ph.D. degree from the Federal University of Santa Catarina, Brazil, in 1993, all in electrical engineering. He is presently a Titular Professor in the Electrical Engineering Department, Federal University of Esp´ırito Santo, Vitória, Brazil, where he has been since 1982. He is a Member of the Power Electronics and Electric Drives Laboratory, where he develops research on power electronics. His main research interests include high-efficiency power converters, switching-frequency power supply, and lighting systems.