High Pressure Sodium Lamp High Power Factor ... - IEEE Xplore

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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 22, NO. 3, MAY 2007

High Pressure Sodium Lamp High Power Factor Electronic Ballasts Using AC–AC Converters Geraldo C. R. Sincero and Arnaldo José Perin, Member, IEEE

Abstract—Electronic ballasts usually present two stages, one for feeding the high pressure sodium (HPS) lamp and the other for power factor correction (PFC). This paper presents electronic ballasts using ac–ac converters without a PFC stage. The use of an indirect ac–ac converter, or an indirect frequency changer, to feed the HPS lamp is investigated. Modulation strategies are proposed to improve the input and output characteristics of the converter. Further studies lead to a novel HPS electronic ballast which uses an ac chopper capable of achieving PFC and supplying high frequency to the lamp. Different converter topologies, bidirectional switches, and a modulation strategy for the ac chopper are presented. Due to the output characteristic of both topologies, the lamp current is modulated at the frequency of the voltage source and the consequences of this operation are discussed. The experimental results for both electronic ballasts supplying 250-W HPS lamps are presented. Index Terms—AC–AC converter, electronic ballast, modulation strategy, power factor correction (PFC), soft switching.

I. INTRODUCTION

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HE large number of lamps required to provide light to public areas and the requirement for energy efficient equipment demands the use of lamps with high luminous efficacy and low maintenance cost. These requirements have made high pressure sodium (HPS) lamps the main source of light for outdoor illumination. The attraction of HPS lamps is their luminous efficacy, about 140 lm/W, and their long life, 24 000 or 32 000 h. In order to connect the HPS lamp to the mains, an interfacing element is required. This element is called a ballast and it is used to limit the lamp’s current due to its negative impedance characteristic [1]. The common practice has been to use electromagnetic ballasts to feed the HPS lamps. However, these elements are heavy and bulky, present poor regulation and a low power factor [2], and can present audible noise due to their low frequency operation. Recently, electronic ballast research has received great attention since they can eliminate many of the unfavorable characteristics of the electromagnetic ballasts. This is accomplished by high frequency operation which allows the magnetic components to be reduced and the audible noise to be eliminated if the switching frequency is high enough. It is also Manuscript received February 24, 2006; revised July 24, 2006. This work was supported by International Rectifier and the Brazilian National Council of Technological and Scientific Development (CNPq). Recommended for publication by Associate Editor J. M. Alonso. G. C. R. Sincero is with the Electrical Engineering Department, Université Laval, Sainte-Foy, QC G1V 4C7, Canada. A. J. Perin is with the Department of Electrical Engineering, Power Electronics Institute, Federal University of Santa Catarina, Florianópolis 88.040970, Brazil (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TPEL.2007.896453

possible to achieve high power factor (HPF) at the input and to control the power of the lamp. The primary advantage of the electronic ballast over the electromagnetic version is the possibility of saving energy since the electronic ballast is capable of controlling the power of the lamp. In fact, even the higher initial cost of the electronic ballast is compensated in the long run by the economy achieved through energy savings. High frequency operation can, unfortunately, generate acoustic resonance in HPS lamps, which is a serious problem that can cause light output variation, arc instability and could cause the arc tube to crack [1]. Furthermore, electronic ballasts should supply only ac current to the lamp in order to avoid the cataphoresis phenomenon. Another concern is the power quality of the electronic ballast input current [3]. There are limits set by international standards on the total harmonic distortion (THD), the amount of individual harmonics allowed and the power factor which can all contribute towards an increase in the cost of energy when not followed. Therefore, HPF is a requirement for ballasts in general. Typically, a PFC stage is used to meet these requirements [4], [5], resulting in an electronic ballast composed of two stages: a boost rectifier and an inverter. The first converter is responsible for providing a dc link to the inverter while sustaining a sinusoidal input current with a low THD. However, this extra stage increases the complexity of the control and the cost and ends up reducing the overall efficiency of the system. With the objective of improving the characteristics mentioned above, electronic ballasts without a PFC stage have been proposed [6]–[11]. The idea of these topologies is to obtain a single stage electronic ballast using an ac–ac converter to supply high frequency ac current to the HPS lamp and to achieve HPF at the input. This paper presents ac–ac converter topologies for HPS lamps. First, the indirect ac–ac converter, or indirect frequency changer, will be briefly presented. The main idea behind this topology is to achieve HPF by using a full bridge diode rectifier with a low output capacitance value and a full bridge inverter. This structure applies high frequency current modulated at low frequency to the load [6]–[11]. Two high frequency modulation strategies will be presented for the voltage inverter to improve the input and output current characteristics. Two possible output filters will also be discussed. Experimental results will be analyzed to complete the study of this converter. An electronic ballast that uses an ac chopper will be investigated [11]. This converter presents HPF at the input and supplies a high frequency ac current to the HPS lamp by using soft switching which increases efficiency. A clamping circuit to protect the switches is not necessary, even during the lamp’s ignition. As with the other proposed structures, this ballast will

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SINCERO AND PERIN: HPS LAMP HIGH POWER FACTOR ELECTRONIC BALLASTS

Fig. 1. Indirect frequency changer circuit diagram.

apply to the lamp high frequency current modulated at low frequency. The bidirectional switches, modulation strategy, input filter and output current characteristics will be discussed. To filter is used at the output. limit the lamp’s current, an Experimental results of a prototype will be presented to prove that the proposed solution is suitable for complying with the international standards. II. INDIRECT FREQUENCY CHANGER The main characteristics of the electronic ballast based on an indirect frequency changer are the absence of a PFC stage, the low frequency output current behavior, and the power control capability. These characteristics can decrease volume, weight, and costs besides increasing efficiency in comparison with other electronic ballasts. The capability of controlling the power of the lamp makes this structure more attractive than electromagnetic ballasts. A circuit diagram of the indirect frequency changer used as an electronic ballast is shown in Fig. 1. The structure is cominput filter to improve the quality of the posed of an input current, a full bridge diode rectifier with a low capacitance which allows HPF, a high frequency inverter, and value for an output filter. A. Input Stage The input stage is composed of the input filter and the full bridge diode rectifier. The input filter reduces the harmonic content of the current drawn from the source by the rectifier. In this case a low-pass – filter is used. The rectifier has a high input power factor due to the low capacitance value of output capacitor . The bus capacitor is used only to supply energy to the inverter at the zero crossing of the line voltage source. To verify the proposed circuit, simulation results for a resistive load without output filter are presented. The simulation pa1 F, 220 V , 12 mH, rameters are: 120 nF, 174 . The input current and the voltage across are shown in Fig. 2. It can be seen that the bus capacitor current has a sinusoidal shape, which guarantees a HPF to the structure. B. Inverter The inverter should operate at high frequency and supply ac current to the HPS lamp. In order to achieve these requirements the full bridge inverter shown in Fig. 1 was chosen. The full bridge configuration was chosen because it allows the maximum value of the input voltage to be applied to the load, thus,

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Fig. 2. Simulated input current and C voltage for resistive load.

Fig. 3. Electronic Ballast with a resonant filter—L

C

.

Fig. 4. Electronic Ballast with a series resonant filter—LC

C

.

supplying rated power to the lamp. Another advantage of this topology is the capability of implementing soft switching. C. Output Filter HPS lamps have negative impedance characteristics, therefore, a current limiting element is necessary. The filter used at the output of the inverter should stabilize the voltage across and current through the lamp, control the current at the zero crossing of the source voltage, provide an ignition pulse to start the lamp and allow the soft switching of the inverter switches. In order to meet these requirements, two topologies are used: a resonant and a series resonant filter . Figs. 3 filter and 4 shows the converter with both output filter structures. D. Modulation Strategies For each of the output filters, a modulation strategy was developed. This was necessary to improve the input current waveform and to decrease the re-ignition time of the HPS lamp after every half-cycle of the input voltage period. The purpose of both modulation strategies is to warm-up the lamp at the zero crossing of the source voltage to decrease the re-ignition time. 1) Frequency Modulation: The first implemented modulation strategy was frequency modulation [6], [9]. The idea is simple: apply a variation in the switching frequency during a half-cycle of the source voltage. Fig. 5 presents the modulation function utilized. Note that the switching frequency varies from

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Fig. 5. Frequency modulation function.

Fig. 6. Modified sinusoidal PWM.

27 to 42 kHz. These limits were used to reach the rated power of the lamp. The reason for decreasing the switching frequency near to the zero crossing of the input voltage is to supply higher power to the lamp to make the re-ignition time shorter. This strategy will resonant filter. be used with the 2) Modified Sinusoidal Pulse-Width Modulation (PWM): To warm-up the lamp at specific moments, variations in the dutycycle will be implemented every half-cycle of the input voltage. The objective is to use sinusoidal PWM; however, some modifications must be made in order to obtain the desired characteristics, the most significant being that a larger pulse will be placed near to the zero crossing of the source voltage [6]–[10]. In order to avoid the cataphoresis phenomenon the dc component of the lamp’s current must be eliminated, which is done and by inverting the drive signals of switches every half-cycle of the source voltage. Fig. 6 shows the pulse-width variation, the modified reference signal used for the comparison with the triangular carrier . The switching signal and the drive signal for switches frequency is constant and, in this illustration, is lower than the frequency used in reality to facilitate comprehension. E. Experimental Results Fig. 1 presents the electronic ballast implemented in the laboratory. The values for the components of the input filter and are the same for both circuits issuing output filters, 13 mH, 120 nF and 1 F. 1) Resonant Output Filter: The first implemented prototype resonant filter of Fig. 3. Frequency modulation employs the

Fig. 7. (a) Low frequency variations of the voltage across and current through the lamp and (b) input current and voltage.

was used to improve the input and output current characteristics. A PIC16F873 microcontroller from Microchip was used to generate the drive signals for the switches. This allows for an easy implementation of the modulation strategy presented input previously. The ballast parameters are: 60 Hz, 220 V voltage; switching frequency varying from 27 to 42 kHz; 363 H; 94 nF; 100 H. The high voltage pulse when is turned on. and is applied to the lamp by are coupled inductors and is a thyristor that along with and , resistor and diode form the capacitors ignition circuit. The high voltage pulse is approximately 3.5 kV. The voltage across and the current through the lamp are shown in Fig. 7(a). It is easy to see the low frequency behavior of the lamp caused by the previously explained characteristics of the ballast. The input voltage and current are shown in Fig. 7(b). It can be seen that the electronic ballast achieves a HPF. This can be further verified by a harmonic analysis of the input current presented in Fig. 8. Table I presents the main parameters obtained from the experimental prototype.


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Fig. 8. Harmonic analysis of the input current.

TABLE I INPUT AND OUTPUT DATA

The efficiency of the prototype is 91.6%. A HPF was obtained with this structure; however, some of the lamps that were tested presented acoustic resonance. This occurred due to the frequency range used in the frequency modulation. As was shown in [26], the radial longitudinal acoustic resonance appears at a power frequency of 80 kHz with a low amplitude. Another problem is the difficulty involved in controlling the power of the load as the lamp ages, which is brought on by the modulation strategy used. 2) Series Resonant Filter: A second prototype was implemented, this time using the series resonant filter of Fig. 4. The modulation strategy used was the modified sinusoidal PWM (Fig. 6). A PIC16F873 microcontroller from Microchip was used to generate the drive signals of the input switches. The ballast parameters are: 60 Hz, 220 V 217 H; voltage; switching frequency of 40 kHz; 1 F; 72 nF. Note in the figure that the auxiliary ignition circuit was removed. The output filter is capable of applying a high pulse to the lamp, which is sufficient to ignite it if the switching frequency is limited to a narrow range around its resonance frequency [12]. A control methodology for igniting the lamp must be implemented using the microcontroller in order to protect the structure in case of failure. The lamp ignition process will be detailed in the ac chopper section. Fig. 9(a) shows the voltage across and current through the lamp. Once again the low frequency characteristic explained previously can be observed. In comparison with the previous implementation, this one presents an improved current characteristic during the time interval when the lamp current was low. The time required for the lamp to re-ignite is important because it is the interval during which the input current presents greater distortion. This fact is illustrated by Fig. 9(b). The input current and voltage are shown in Fig. 10. A small peak current can be observed, which is caused by the re-ignition of the lamp. The harmonic analysis of the input current is

Fig. 9. (a) Low frequency variation of the voltage across and the current through the lamp and (b) lamp voltage and input current.

Fig. 10. Input current and voltage.

presented in Fig. 11. Note that the harmonic content is smaller than in the case of the previous implementation.

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Fig. 11. Harmonic analysis of the input current with the LC filter.

TABLE II INPUT AND OUTPUT DATA

Table II presents the main parameters obtained from the experimental prototype. The efficiency of the prototype is 92%. A HPF of approximately 0.978 was achieved. Neither the harmonic analyses shown in Figs. 11 and 8 meet the IEC 61000-3-2 Class C limits. However, meeting the limits of the standard was not the objective of the work up to this point. By simply modifying the input filter, however, the prototype can be made compliant with the Class C limits. The efficiency and the PFC performance of the series resonant filter were increased when compared to the resonant filter since the modulation strategy used allowed the re-ignition time to be decreased which was responsible for improving the input current waveform. Acoustic resonance was not observed in this prototype. A contributing factor to this was the characteristic of the modulation strategy. The PWM strategy used can spread the harmonic content of the lamp’s power, so that the value of some harmonic components is low enough to avoid acoustic resonance [13], [14], [25], [26]. Another cause for concern was the crest factor of approximately 2 observed in the current of the lamp. This subject will be discussed in the ac chopper section. III. AC CHOPPER The ac–ac converters are well-known structures in the technical literature. Many topologies were proposed to improve the input current quality, the regulation, the commutation, and to eliminate the need for clamping circuits [15]–[18]. In comparison with other converters used in electronic ballasts, the proposed topology presents fewer components and high power density. The topology chosen for this application is presented in Fig. 12(a) [10], [11]. The regulation capacity and HPF at the input are important characteristics that were taken into account when choosing this topology. It is also important to note that only two semiconductors will be conducting at the same time,

Fig. 12. (a) AC chopper and possible bidirectional switch configurations, (b) conventional IGBTs and diodes, and (c) RB–IGBT pair.

Fig. 13. Proposed ac–ac converter topology.

which increases the efficiency. The converter uses two bidirectional switches ( and ) formed by two pairs of metal oxide semiconductor field effect transistors (MOSFETs) ( and ), Fig. 12(a). and could also be composed of two conventional IGBTs and two diodes [Fig. 12(b)] or a pair of reverse blocking IGBTs (RB–IGBT) [Fig. 12(c)] [19], [20]. With the RB–IGBT pair, only one semiconductor is active during each operating stage. In the end, the MOSFET switches were used due to their availability, to the possibility of using a low cost bootstrap integrated circuit driver, and to the advantage of obtaining soft-switching commutation. A. Modulation Strategy The bidirectional switches are composed of MOSFETs (voltage unidirectional switches) which require a proper commutation strategy in order to avoid shorting the voltage source or causing a discontinuity in the inductive load current. To overcome these difficulties some modulation strategies are proposed [17], [18] for this converter. A suitable and simple solution is two step modulation, which is based on the PWM technique and depends on the polarity of the input voltage. This modulation strategy allows soft-switching operation and the use of switching modules and the bootstrap drivers. However, in order to use bootstrap drivers a simple modification in the converter configuration is necessary. Fig. 13 presents the implemented ac–ac converter. The two step modulation drive signals and the input and output voltages are shown in Fig. 14, where low frequency operation is assumed for better understanding. Note that while


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Fig. 14. Input and output voltage and drive signals for two step modulation.

Fig. 15. Main waveforms of the ac chopper over a switching period.

TABLE III SPECIFICATIONS OF THE ELECTRONIC BALLAST

the input voltage is positive, switches are kept on while operate in a complementary fashion based switches on the PWM modulation. When the input voltage becomes remain on throughout the entire interval negative, operate in a complementary fashion based on the and PWM modulation. B. Description of the Operating Stages Based on the two step modulation strategy presented above, the operation stages are described. It is assumed that the switching frequency is high enough to consider voltage constant during the entire switching period. The input filter is neglected in order to simplify the analysis. Also, it is assumed that the filter was properly designed, so that, the output filter current is lagging . The analysis considers that the input voltage is positive, therefore, and remain on during the entire switching period.

Fig. 16. AC–ac converter operating stages: (a) stage 1, (b) stage 2, (c) stage 3, (d) stage 4; (e) transition from stage 2 to stage 3; and (f) transition from stage 4 to stage 1.

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is equal to input voltage , consequently Fig. 16(a). the magnitude of the output filter current decreases. When the load current reaches zero at time this stage ends. Stage 2 : At the output filter current becomes positive, therefore, switch and diode conduct the load current Fig. 16(b). Voltage is equal to input voltage and the load current increases. This stage ends when is turned off. Stage 3 : This stage begins at when is turned on. However, the output filter current is positive, therefore, diode and switch conduct the load current Fig. 16(c). is equal to zero, consequently the magnitude of the output filter current decreases. When the load current reaches zero at time this stage ends. : At the output filter current becomes Stage 4 and diode conduct the negative, therefore switch load current Fig. 16(d). Voltage is equal to zero and is the load current increases. This stage ends when turned off. The main waveforms for a switching period are presented in Fig. 15. The description of the operating stages can be extended to the negative half-cycle. C. Commutation Analysis The ac–ac converter configuration of Fig. 13 is capable of complete soft-switching operation when using two step modulation. This characteristic is achieved by means of parallel capacitors and the inductive characteristic of the output filter. Note that at and the load current crosses zero which consequently produces a soft transition from stage 1 to stage 2 and from stage 3 to stage 4. In the other stage transitions softswitching is achieved by charging and discharging the parallel capacitors. Fig. 16(e) and (f) show the zero voltage switching (ZVS) transitions. D. Design of the Electronic Ballast Using an ac Chopper

M

V

M

Fig. 17. (a) Output voltage , output filter voltage and drive signals of and ; (b) input voltage and current, (c) lamp’s voltage and current at high during the lamp’s frequency; and (d) lamp’s voltage and voltage across ignition.

M

Stage 1 : This stage begins at when is turned on. However, the output filter current is negative, therefore, diode and switch conduct the load current

After presenting the ac chopper’s main characteristics, it is necessary to adjust the converter for the proposed application. When driving a HPS lamp certain aspects should be considered. It is necessary to use an output filter to limit the lamp current, an input filter to reduce the input current harmonic levels and an ignition circuit to provide a high voltage pulse to the lamp. The design of these elements will be presented in following section. The first step is to determine the specifications of the electronic ballast, which are presented in Table III. 1) Output Filter: The filter utilized at the output of the ac chopper should stabilize the voltage and the current of the lamp, control the current at the zero crossing of the source voltage, provide a high voltage pulse for the lamp ignition and allow soft-switching operation. The filter is the first part to be designed due to the importance to determinate the lamp’s current and the ignition system used. series In order to perform all of these functions, the , which resonant filter is used and is formed by inductor limits the lamp current, capacitor , which filters the dc voltage component, and capacitor , responsible for applying the high voltage pulse. The filter was presented in Fig. 13.


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Fig. 18. AC chopper electronic ballast implemented in the laboratory.

The design methodology of the filter is well-known in the technical literature and is presented again in [10], [23]. Equation (1) defines the filter parameters. For the electronic ballast specifications presented above the output filter parameters 127.8 H; 195 nF; 106.8 nF are (1) (2) (3) 2) Lamp Ignition: Note that in Fig. 13 there is no auxiliary circuit for igniting the lamp. The filter is responsible for applying a high voltage pulse to the lamp when the switching frequency is close to the resonance frequency of the filter [24]. A control methodology for the lamp ignition must be implemented to protect the structure in case ignition fails. This should be the next step after designing the output filter in order to determine the maximum value of the voltage used to ignite the lamp, which is important for the input filter design. The first step of the ignition procedure is to detect the zero crossing of the source voltage, then, 3.5 ms later, the switching frequency is changed to a value close to the resonance frequency of the filter. Zero detection and a delay time are necessary when applying the high voltage pulse whenever the instantaneous input voltage is larger than 90% of its rated rms value. The resonance switching frequency is used for 1 ms. This is a safe time interval during which the voltage across the lamp will reach a value capable of starting the lamp but not dangerous enough to damage the structure. Afterwards, the switching frequency is set at its rated value. This procedure is the same as was used for igniting the lamp of the indirect frequency changer ballast using the output filter [6]. 3) Input Filter: The input stage is composed of a simple lowpass filter, and in Fig. 13. The input filter has the function of adjusting the harmonic level of the current flowing through the source which is produced by the high frequency operation of the ac–ac converter. The voltage across capacitor determines the maximum voltage level across the switches of the ac chopper. Consequently, it is necessary to specify, in the input filter design, a low ripple value for the voltage across .

In steady-state operation this is not a huge problem. However, during ignition, when a high voltage pulse is necessary to break the initial inertia of the HPS lamp plasma, the voltage across the switches can reach damaging values of up to 1,000 V. This will is low and the inductance of occur if the capacitance of is high. Thus, the design of the low-pass filter should present a capacitance for high enough to avoid creating high voltages across the switches. For the design of the input filter it is assumed that the converter behaves as a resistive load as defined by (4). The input filter transfer function is presented in (5) in the domain and in (6) in the domain. The parameters that define the filter characteristics, cutoff frequency and damping factor are obtained from this equation and are shown in (7) and (8). After defining these parameters, inductor and capacitor are defined by (9) and (10). To achieve proper voltage clamping of the switches, the cutoff frequency is defined as 2 , where 3.1 kHz. For the electronic ballast specifications presented, the input filter parameters are: 1 mH and 2.66 nF (4) (5) (6) (7) (8) (9) (10) 4) Numerical Simulation: A numerical simulation was performed to verify the study presented above. The circuit presented in Fig. 13 was used in the simulation. The HPS lamp is modeled as a resistance. The simulation parameters are: 220 V , 1 mH, 2660 nF, 60 kHz, 127, 8 H, 108 nF, 195 nF, 33 . Fig. 17(a) presents the output voltage and output

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Fig. 19. Microcontroller output signal.

Fig. 21. Lamp’s voltage and current and voltage across ignition.

Fig. 20. Drive signals of

M

and

M

during the lamp’s

M.

current of the ac chopper and the drive signals of MOSFETs and using two step modulation. Fig. 17(b) shows the input voltage and current. Note that the converter achieves a HPF at the input; however, it is necessary to remember that in the simulation the HPS lamp is modeled as a load resistance, thus, the lamp’s low frequency behavior is not analyzed. The lamp voltage and current waveforms at high frequency are shown in Fig. 17(c). To prove the efficiency of the input filter at clamping the voltage across the switches, a simulation of the lamp’s ignition is performed. Fig. 17(d) shows the lamp’s . Note that the voltage voltage and voltage across switch does not exceed 500 V. across 5) Experimental Results: A prototype of the electronic ballast presented in Fig. 18 was implemented in the laboratory. The prototype was designed for a 250-W HPS lamp according to the specifications presented in Table III. In order to generate two step modulation, a PIC16F716 microcontroller from Microchip and four OR logic ports implemented by the 7432 integrated circuit were used. Fig. 19 shows the microcontroller output signals and the logic that generates the two step modulation. The ac chopper topology allows the use of a low cost bootstrap driver. The bootstrap circuit is necessary to generate the virtual voltage reference for the switches. In the experiment the IR2110 integrated circuit was used. The Mosfet switches are IRFP27 N60 K. and , Fig. 20 shows the drive signals for switches where two step modulation was used. The lamp’s voltage and during the ignicurrent waveforms and the voltage across tion of the lamp are presented in Fig. 21. It is important to note does not exceed 500 V which that the voltage across switch

Fig. 22. Lamp’s voltage and current at (a) low frequency and (b) high frequency.

demonstrates the ability of the input filter to clamp the voltage during ignition. across The lamp’s voltage and current are shown in Fig. 22(a)—low frequency and Fig. 22(b)—high frequency. The resistive behavior of the lamp is observed at high frequency. The lamp’s


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is 94% and the power factor is 0.965. Acoustic resonance was not observed during experimentation. The ac chopper allows the use of modified sinusoidal PWM which spreads the harmonic content of the lamp’s power and reduces the crest factor and the THD by decreasing the lamp’s re-ignition time. If the values of some harmonic components are low enough, acoustic resonance can be avoided [13], [14], [25], [26]. The frequency used here is higher than that used when acoustic resonance is more pronounced [26]. Dimming control and the regulation of the lamp’s power can also be implemented. IV. CONCLUSION

Fig. 23. Input voltage and current.

Fig. 24. Harmonic analysis of the input current.

TABLE IV INPUT AND OUTPUT DATA

voltage and current present the same low frequency behavior as when electromagnetic ballasts are used. This characteristic was expected due to the low frequency modulation at the output of the ac chopper. Note that the lamp current in this case and in the case of the indirect frequency changer electronic ballast presents a crest factor of approximately 2, which for fluorescent lamps would be a problem. However, recent studies show that for high frequency operation a crest factor higher than 1.7 is not a concern for HPS lamps [21]. In fact, the electrode erosion, under this condition, is lower than that obtained for line frequency operation. For low power lamps there is a strategy that can reduce the crest factor [22]; however, this strategy cannot be applied to 250-W lamps because the energy storage elements would be too large. The input voltage and current are presented in Fig. 23. Note that the peak current occurs during the lamp re-ignition. The harmonic analysis of the input current reveals a low harmonic content equal to 21.4%, as shown in Fig. 24. Table IV presents the main results of this prototype. The efficiency of the structure

An electronic ballast using an indirect frequency changer to achieve HPF for HPS lamps was analyzed. With the objective of improving the quality of the input and output currents, two modulation strategies were proposed and both presented good results. An advantage of these modulation strategies is the possibility of avoiding acoustic resonance, but this characteristic should be further investigated. Both output filters presented a HPF, but the combination of the PWM strategy and the series resonant filter presented the best results of all. A novel electronic ballast using an ac chopper was proposed and the possible converter topologies and bidirectional switch implementations were analyzed. The modified ac chopper allows the use of a bootstrap driver. The analysis presented two step modulation, the operating stages and soft switching operation. Some considerations for the input and output filter design were taken into account so that the ac–ac converter could be used for the proposed application. A detailed description of the ignition procedure was presented. The circuits used to generate the modulation strategy were presented. The crest factor, which was an initial concern, might not be a problem for the HPS lamp in high frequency operation after all. However, the crest factor can be reduced, if necessary, by using an appropriate filter, but this strategy is normally only used in low power applications. Simulation and experimental results were presented for the prototype of the electronic ballast for a 250-W HPS lamp. Acoustic resonance was not observed and dimming control is possible as well as lamp power regulation. When compared to the traditional PFC approach to electronic ballasts, the proposed structures presented better results in terms of efficiency and PFC performance. Furthermore, the proposed electronic ballasts have a lower cost and reduced implementation complexity. However, the absence of a dc link is a problem if a low crest factor is required, since the circuitry necessary to improve this characteristic would inevitably increase the overall cost. REFERENCES [1] J. J. Groot and J. A. J. M. Vliet, The High Pressure Sodium Lamp. London, U.K.: MacMillan, 1986. [2] C. Blanco, M. Alonso, E. López, A. Calleja, and M. Rico, “A single stage fluorescent lamp ballast with high power factor,” in Proc. IEEE APEC, 1996, pp. 616–627. [3] Limits for Harmonic Current Emissions, Eur. Std. EN 61000-3-2, 1999. [4] C. Brañas, F. J. Azcondo, and S. Bracho, “Evaluation of an electronic ballast for HID lamps with passive power factor correction,” in Proc. IECON’02, 2002, vol. 1, pp. 317–376.

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[5] F. J. Ferrero, M. Rico, J. M. Alonso, C. Blanco, and J. Ribas, “Analysis and design of an ac–ac resonant converter as a high pressure sodium lamp ballast,” in Proc. IECON’98, 1998, vol. 2, pp. 947–952. [6] G. C. R. Sincero, S. A. Wuerges, and A. J. Perin, “A 250 W high pressure sodium lamp high power factor electronic ballast using an indirect frequency changer,” in Proc. Brazilian Power Electron. Conf. (COBEP’05), Jun. 2005, pp. 332–337. [7] F. S. Reis et al., “Low cost high power factor electronic ballast for high pressure sodium lamps,” in Proc. Induscon’04, Oct. 2004, pp. 1000–1005. [8] C. Licitra, L. Malesani, G. Spiazzi, P. Tenti, and A. Testa, “Single-ended soft-swithcing electronic ballast with unity power factor,” in Proc. IEEE IAS Annu. Meeting, 1991, pp. 953–958. [9] S. A. Wuerges, “Electronic Ballast for a 250 W High Pressure Sodium Lamp Without an Active Power Factor Correction Stage,” M.S. thesis, Univ. Fed. Santa Catarina (UFSC), Florianópolis, Brazil, 2004. [10] G. C. R. Sincero, “A 250 W High Pressure Sodium Lamp High Power Factor Electronic Ballast Using an ac Chopper,” M.S. thesis, Univ. Fed. Santa Catarina (UFSC), Florianópolis, Brazil, 2005. [11] G. C. R. Sincero, A. S. Franciosi, and A. J. Perin, “A 250 W high pressure sodium lamp high power factor electronic ballast using an ac chopper,” in Proc. Eur. Conf. Power Electron. Appl. (EPE’05), Sep. 2005, [CD ROM]. [12] E. I. Pereira, C. B. Nascimento, and A. J. Perin, “Electronic ballast for fluorescent lamps with the PFC stage integrated with the resonant inverter,” in Proc. 35th Annu. IEEE Power Electron. Spec. Conf., Jun. 2004, vol. 5, pp. 4050–4056. [13] H. Faehnrich and E. Rasch, “Electronic ballast for metal halide lamps,” J. Illum. Eng. Soc., pp. 191–140, 1988. [14] L. Laskai, P. N. Enjeti, and I. J. Pitel, “White-noise modulation of high-frequency high-intensity discharge lamp ballast,” IEEE Trans. Ind. Appl., vol. 34, no. 3, pp. 597–605, May/Jun. 1998. [15] J. C. Bowers, S. J. Garret, and H. A. Nienhaus, “A solid state transformer,” in Proc. Power Conv. Int., May/Jun. 1980, pp. 56–65. [16] N. Marjanovic, “Semiconductor auto-transformer with continuous voltage transformation ratio variation,” PCI Proc., pp. 262–267, Sep. 1982. [17] P. N. Enjeti and S. Choi, “An approach to realize higher power PWM ac controller,” in Proc. 8th Annu. Appl. Power Electron. Conf. Expo (APEC’93), Mar. 7–11, 1993, pp. 323–327. [18] J. H. Kim and B. H. Kwon, “Three-Phase ideal phase shifter using ac choppers,” Proc. Inst. Elect. Eng., vol. 147, no. 4, pp. 329–335, Jul. 2000. [19] M. Takei, T. Naito, and K. Ueno, “The reverse blocking IGBT for matrix converter with ultra-thin wafer technology,” in Proc. ISPSD’03, 2003, pp. 156–159. [20] A. Odaka et al., “An application technique of a novel IGBT with reverse blocking capability for a direct linked type converter,” in Proc. EPE’03, Sep. 2003, [CD ROM].

[21] W. Kaiser, A. F. Correa, and R. P. Marques, “Electrode erosion in pulse operated high-pressure-sodium lamps,” in Proc. 39th Annu. Ind. Appl. Soc. Conf., 2004, vol. 2, pp. 1362–1367. [22] L. Malesani, L. Rossetto, G. Spiazzi, and P. Tenti, “High efficiency electronic lamp ballast with unity power factor,” in Proc. IEEE Ind. Appl. Soc. Annu. Meeting, 1992, vol. 1, pp. 681–688. [23] M. K. Kazimierczuk and W. Szaraniec, “Electronic ballast for fluorescent lamps,” IEEE Trans. Power Electron., vol. 8, no. 4, pp. 386–395, Oct. 1993. [24] M. K. Kazimierczuk, N. Thirunarayan, and S. Wang, “Analysis of series-parallel resonant converter,” IEEE Trans. Aerosp. Electron. Syst., vol. 29, no. 1, pp. 88–99, Jan. 1993. [25] J. Correa, Ponce, M. Arau, J. Sanchez, and J. M. Alonso, “Evaluation of frequency modulation techniques to avoid acoustic resonances in electronic ballast for HID lamps: Analysis and methodology,” in Proof. IEEE CIEP’04, Oct. 2004, pp. 245–250. [26] H. L. Witting, “Acoustic resonances in cylindrical high-pressure arc discharges,” J. Appl. Phys., vol. 49, pp. 2680–2683, May 1978. Geraldo C. R. Sincero was born in Maringá, Brazil, in 1980. He received the B.E. and M.Sc. degrees in electrical engineering from Universidade Federal de Santa Catarina, Florianopolis, Brazil, in 2003 and 2005, respectively, and is currently pursuing the Ph.D. degree in the Electrical Engineering Department, Université Laval, Sainte-Foy, QC, Canada. From 2005 to 2006, he was with the Power Electronics Institute (INEP), Florianopolis, as a Researcher on the Active Filter Project. His research interests include power electronics, electronic ballast systems, ac converters, power factor correction, and machine drives.

Arnaldo José Perin (M’86) was born in Nova Prata, Brazil, in 1953. He received the B.E. degree in electronic engineering from the Pontificia Universidade Catolica do Rio Grande do Sul, Porto Alegre, Brazil, in 1977, the M.Sc. degree in electrical engineering from Universidade Federal de Santa Catarina, Florianopolis, Brazil, in 1980, and the Dr.Ing. degree from the Institut National Polytechnique de Toulouse (INPT), Toulouse, France, in 1984. He joined the Electrical Engineering Department, Universidade Federal de Santa Catarina (UFSC) in 1980 and is now engaged in education and research on power electronics analysis and design. His research interests include power electronics, modulation, ac converters, and power factor correction. Since 1993, his research interests have concentrated more specifically on electronic ballast to use with fluorescent lamps and with HID lamps.