Insulated-Gate Bipolar Transistor Rectifiers - IEEE Xplore

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Aug 19, 2014 - When we say IGBT rectifier, we mean a PWM rectifier built with IGBTs (Fig- ure 1). A PWM rectifier (sometimes also called an active rectifier) is ...
Insulated-Gate Bipolar Transistor Rectifiers ©istockphoto.com/Jess_Yu

Why They Are Not Used in Traction Power Substations

Vitaly Gelman

Digital Object Identifier 10.1109/MVT.2014.2333762 Date of publication: 19 August 2014

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I

nsulated-gate bipolar transistors (IGBTs) are widely used in high-power converters. The definite advantages of IGBT rectifiers [also called pulsewidth modulation (PWM    ) rectifiers] are zero reactive power, low harmonics, and inherent power recuperation capability. However, stationary traction rectifiers are built with either thyristors or diodes, not with IGBTs. This article compares IGBT and thyristor rectifiers and analyzes the factors precluding the use of IGBT rectifiers at traction power substations. When we say IGBT rectifier, we mean a PWM rectifier built with IGBTs (Figure 1). A PWM rectifier (sometimes also called an active rectifier) is just a PWM inverter used to transfer power from the ac side to dc. It has the same power circuit as a PWM inverter, but the difference is in the control—instead of controlling the ac voltage value and frequency, we are regulating the dc voltage by controlling the ac. To operate at unity power factor, the ac side current in each phase should be proportional to a product of the primary phase voltage and

1556-6072/14©2014ieee IEEE vehicular technology magazine | SEPTEMBER 2014

required dc current. Furthermore, the reactive component can be added to the ac current to generate reactive power if needed. PWM rectifiers offer many advantages: the ability to regulate the dc voltage, inherent energy recuperation, unity power factor, and low harmonics. PWM rectifiers have been used successfully for four decades for train propulsion with ac catenaries [1], first with silicon-controlled rectifiers (SCRs) with forced commutation then with gate turn-off thyristors (gate turn-off SCRs) and later with IGBTs. In recent years, there has been substantial progress in IGBTs, and now the IGBT is the device of choice for converters with dc link voltage of interest for traction rectifiers (700–1,500 Vdc) [6], so we will use the terms PWM rectifier and IGBT rectifier interchangeably. However, despite technological advances, the IGBTs are not used for stationary (or wayside) traction rectifiers, although some academic papers have investigated the concept [2], [3]. What is the difference between a rolling stock rectifier and stationary traction rectifier that makes IGBTs widely accepted in the first case and prevents their use in the second case? In our opinion, the major factor is a high-impedance, single-phase railroad ac catenaries power system with wide voltage regulation [1, Sections II and III-A]. As a result, the railroads place stringent requirements on both harmonics distortion and reactive power [1, Section II] necessitating active ac shaping of the converter rectifier. This can be done with a PWM rectifier or an earlier boost type converter [1, Figure 4]. A single-phase line-commutated SCR rectifier generates high harmonics and consumes high reactive power, precluding its use for rolling stock. The stationary traction rectifiers operate with multiphase incoming power, which is better regulated and has lower impedance. Therefore, we have two factors acting in the same direction: the rectifier generates lower harmonics and reactive power while, at the same time, the power system can tolerate a higher level of reactive power and harmonics. The second distinction is the environment, namely, the cooling methods: rolling stock converters are liquid cooled. (“Fluid cooling is inevitable for high-power applications” [4, p. 15].) The stationary traction rectifier uses air cooling that is more robust and less expensive. The rest of this article examines the details of a controlled rectifier for traction application.

The losses occur in the IGBT and a parallel diode, the losses can be divided into static losses and switching losses. regulation of 6%, limits the nominal rectifier voltage to be the same as the nominal train voltage [5, Section II]. The constraint of the diode rectifier voltage in turn (together with voltage regulation at higher current) limits the traction power substation (TPSS) spacing to about 1 mi for 750-V systems. We have more flexibility with the selection of regulated output voltage because a regulated rectifier can compensate for changes in both incoming line voltage and load current. The U.S. transportation authorities typically specify the output voltage of 825–850 Vdc for the systems with 750 Vdc trains. The increase of the nominal voltage and the ability to keep a constant output voltage with a current up to 150–200% load gives more margin to compensate for rails resistance voltage drop (e.g., see Figure 2 from [5] for a comparison of diode and thyristor rectifiers’ loading curves), thus allowing one to increase the spacing between the TPSSs. The additional cost of a regulated rectifier is a small incremental portion of the installed TPSS cost. Therefore, using regulated rectifiers allows us to cut the total system cost [5].

Requirements for Regulated Traction Rectifiers The traction rectifiers typically operate with trains having a nominal voltage of either 750 or 1,500 Vdc; in the United States, it is mostly 750 Vdc. Consequently, for regulated rectifiers, the nominal voltage is typically specified as 825–850 Vdc for systems with 750-V trains. The train starting current requires a very high overload capacity; for diode rectifiers, it is 450% of the rated current. For regulated rectifiers, the maximum current is specified at lower values, from 300 to 400%. The overloads continue for 15–30 s or even longer. The rectifier must be able to withstand the short circuits caused by faults inside the TPSS and outside. The fault duration is at least 85 ms (five cycles), most likely 100–250 ms to allow for selectivity (the dc breaker

+

Controlled Traction Rectifier Versus Diode Rectifier A majority of stationary traction rectifiers are unregulated diode rectifiers: they are robust, simple, and well understood by the end users. Their drawback is the inability to compensate for the incoming voltage and load changes. Modern trains can operate with the voltage up to 120% nominal, and this condition, together with the incoming voltage regulation of 5–10% and load

SEPTEMBER 2014 | IEEE vehicular technology magazine

-

Figure 1 An IGBT rectifier.



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Insulated-gate bipolar transistors (IGBTs) are widely used in high-power converters. should clear the external fault first). Ideally a regulated rectifier should limit the fault current to reduce the breaker and rectifier transformer stresses to extend their service lives. The rated rectifier power is 1–5 MW, while the typical value for heavy rail application is 3 MW. To compare the regulated traction rectifier alternatives, let us assume that we have a rectifier with rated voltage 825 Vdc, 3-MW rated power, and 300% overload capacity. The rated current of 3 MW at 825 Vdc is 3.63-kA dc; the current at 300% is 10.9-kA dc. We will compare a PWM IGBT-based rectifier with a phase-controlled, thyristor-controlled rectifier (TCR). In sizing the IGBT rectifier, we need to consider the overload capability and short-circuit withstanding capability.

IGBT Rectifier Requirements for Overload Current To regulate the output dc voltage, the IGBT rectifier must have an incoming ac voltage with peak value below the dc voltage. Furthermore, we need additional margins to compensate for the transformer leakage inductance voltage drop and the incoming ac voltage regulation. We estimate that the output dc voltage, Vd, needs to be 20% above the incoming ac rated voltage peak. Then,

Rectifier Loading Curves

assuming negligible losses in the IGBT rectifier, from the power being equal at ac and dc sides, we get the following:

3$

Vm I m = Vd I d 2

or

I m = 2 Vd I dMax = 2 1.2 10.9 = 15.1 kA pk,  3 Vm 3 1

where Vm and I m are peak values of the IGBT rectifier ac voltage and current, and Vd and I d are the rectifier dc voltage and current. The secondary transformer voltage

Vd V 1 m V= = .2 = 825 = 486 Vac.  2 2 1.2 $ 2

To provide 15.1 kA, we will need to connect several IGBTs in parallel. Presently, the biggest available IGBT modules are rated at 3,600-A dc; the actual ac is lower because of the switching losses and cooling limitation. If we connect six IGBTs in parallel, we will need to provide peak ac per IGBT of 15.1/6 = 2.53 kA pk; with eight IGBTs, we will need 1.9 kA pk. In both cases, we assumed equal current sharing. Table 1 shows the losses for three 3,600-A 1,700 V IGBTs, see “IGBT Losses for Sinusoidal Current” for the calculation details. For the high-efficiency forced-air heatsink, we can assume a thermal impedance of 40 °C/kW per IGBT module [7]. With an ambient temperature of 40 °C and 2-kW total IGBT losses, we get the heatsink temperature

850



800

40 cC + 2 kW $ 40 cC/kW = 120 cC. 

750 Voltage

700 650 600 550

Diode TCR

500 450 450

0

50

100 150 200 250 300 350 Load (%)

400

Figure 2 The voltage regulation of a diode rectifier and a TCR [5]. Table 1  IGBT losses per IGBT module (in kilowatts). I Peak 1.9 kA (eight IGBTs) 2.53 kA (six IGBTs)

Carrier Frequency ABB

Hitachi

Infineon

500 Hz

1.547

1.461

1.331

2,000 Hz

2.681

2.747

2.390

500 Hz

2.288

2.107

1.941

2,000 Hz

3.798

3.820

3.350

From the maximum junction temperature of 150 °C and the temperature difference between the junction and heatsink of 10 °C, we get the margin of only 20 °C (150 − 120 − 10 = 20). From Table 1, we can see that we get IGBT module losses close to 2 kW with a 500-Hz carrier frequency and ideal current sharing with six devices in parallel (2.53 kA pk). However, this does not leave us additional margins for the current imbalance, additional losses due to switching frequency ripples, and other contingencies. It would be safer to go with eight parallel devices (1.9 kA pk) and a 500–1,000-Hz carrier frequency, with the losses below 2 kW per IGBT module.

IGBT Rectifier Requirements for Withstanding Short Circuit The IGBT rectifier has a built-in diode rectifier (see Figure 1), so, even if we turn off all IGBTs, the diodes can still conduct the current. Therefore, the IGBT rectifier, unlike the TCR, cannot limit the short-circuit current but rather has to withstand it. We should allow the feeder breaker to open first to isolate the fault, so the IGBT

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IGBT Losses for Sinusoidal Current

r

#

The losses occur in the IGBT and a parallel diode, the losses can be divided into static losses and switching losses.

0

i = I m sin z;

= =

Let us further assume the modulation to be sinusoidal with a phase shift a with respect to the current

1 + m $ sin (z + a) 2 I m sin 2 {d{ 2

# 0



Static Losses We calculate the static losses following [9]. Let us assume that the ac i is a sine wave with amplitude I m and duty cycle c .



r

ci 2 d{ =

r

2 Im 2

#

sin 2 { d{ +

0

rI 2m

4

+

r

2 mI m 2

#

sin ({ + a) $ sin 2 {d{

0

2mI 2m cos a. 3

The last term in the previous equation is evaluated as r

#

{ = wt

r

sin ({ + a) $ sin 2 {d{ = cos a

0

#

sin 3 {d{

0 r



1 + m $ sin ({ + a) c= , 2



#

+ sin a

r

= cos a

where ~ is the line frequency and m is the modulation index. In real converters, the duty cycle c is not sinusoidal to achieve better dc voltage utilization, but to get a losses estimate, we will neglect that effect. We get the losses by averaging the losses during the positive half wave of the current. Let us assume that the IGBT forward voltage drop VIGBT and parallel diode voltage drop VD can be written as

2 VTH -IGBT I m r Im 2m cos a rm r c + m ` 1 + 4 cos a j + IGBT 2r 2r 4 3 2 V I r I = TH -IGBT m ` 1 + rm cos a j + IGBT m c 1 + 8m cos a m . 2r 4 8 3r

DPIGBT =

We can perform similar calculations for the diode losses, but, since the only difference between the equations for the diode and IGBT losses is the use of a factor 1− c instead of c and this corresponds to using modulation index −m instead of m , we can get the equation for the diode losses

Similarly, the diode static losses are PD ({) = (1 - c) VD $ i = (1 - c) (VTH -D i + rD $ i 2) .



To get the average losses per mains frequency period, we need to integrate the IGBT and diode losses over the positive half wave of the current, 0 < { < r, and divide the result by the period (2r) DPIGBT = 1 2r =

r

# 0



# 0

PIGBT ({) d{ = 1 2r

VTH -IGBT 2r r

cid{ =

r

# 0

= Im 2

r

#

cid{ +

0

r

# 0

rIGBT 2r



c ^VTH -IGBT i + rIGBT $ i 2 h d{

r

0

sin { d{ + mI m 2

0

= I m ` 1 + rm cos a j 4

r

#

sin ({ + a) $ sin {d{

0

SEPTEMBER 2014 | IEEE vehicular technology magazine

VTH -IGBT I m r I 2m 8 . r c1 + m ` 1 + 4 j + IGBT 2r 8 3r

r I2 DPD = VTH -D I m ` 1 - 0.8r j + D m c 1 - 8 $ 0.8 m 2r 4 8 3r rD I 2m V . 6 4 TH -D I m = (1 - 0.2r) + c1 m. 2r 8 3r

Switching Losses We calculate the switching losses in a simplified way, assuming that they are linearly changing with current. Every time the IGBT turns on or off, there are switching energy losses EON and EOFF . These losses depend on the current, bus voltage, and temperature. As a first approximation, we can assume the losses EswIGBT to be proportional to the current

ci 2 d{

1 + m $ sin (z + a) I m sin {d{ 2

#

DPIGBT =

For the diode, we can assume that m = 0.89 and cos a = 0.9

r

#

r I2 DPD = VTH -D I m ` 1 - rm cos a j + D m c 1 - 8m cos a m . 2r 4 8 3r

To provide the unity power factor, we need to compensate for the voltage drop on the rectifier transformer impedance and, thus, the cos a is less than unity, but, even at 300% load and 15% transformer impedance, it would still be about 0.9. For the IGBT losses estimates, we can assume the worst case m = 1 and cos a = 1; then we get the static losses

PIGBT ({) = cVIGBT $ i = c (VTH -IGBT + rIGBT $ i) i = c (VTH -IGBT i + rIGBT $ i 2) .



sin 3 {d{ = 4 cos a. 3

Combining the equations together we arrive at

where VTH–IGBT and rIGBT are the IGBT threshold voltage and dynamic (or slope) resistance, VTH -D and rD are the parallel diode threshold voltage and dynamic resistance, and i is the current through the device. The IGBT static losses are



# 0

VIGBT = VTH -IGBT + rIGBT $ i VD = VTH -D + rD $ i,



cos { $ sin 2 {d{

0





E swIGBT (i) = E ON + E OFF =

i E (I ), I swIGBT swIGBT swIGBT

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where I swIGBT is the current selected to better approximate the switching losses. Similarly, the diode switching energy losses during turn-off can be presented as



E swD (i) = E OFF = i E swD (I swD) . I swD

To get the average switching losses per mains frequency period, we need to integrate the IGBT and diode switching losses over the positive half wave of the current, 0 < { < r, and divide the result by the period (2r) r

DPswIGBT = 1 2r



#

fE swIGBT ({) d{

0 r

= 1 2r

#

f

0

i E (I ) d{ I swIGBT swIGBT swIGBT

fE swIGBT (I swIGBT) E (I ) = 1 2I m = I m f swIGBT swIGBT . I swIGBT I swIGBT r 2r Similarly, for the diode switching losses, we get



DPswD = 1 2r

r

# 0

fE swD ({) d{ = 1 2r

r

# 0

f i E swD (I swD) d{ I swD

E (I ) = I m f swD swD . I swD r

IGBT Parameters and Losses To estimate the losses, we need to use the actual IGBT parameters (Table S1). We will use a 3,600-A 1,700-V IGBT from

three leading manufacturers: ABB’s SNA5SNA 3600E170300, Hitachi’s MBN3600E17F, and Infineon’s FZ3600R17HP4_B2 [10]–[12]. All of these devices are specified to operate at a 150 °C junction temperature. To estimate the threshold voltage and dynamic resistance for both the IGBT and diode, we used typical curves for on-state characteristics at 150 °C. We used best straight-line approximation for the on-state voltage at 1,200, 2,400, and 3,600 A. To estimate the switching losses for all IGBTs, we used E swIGBT at current 3,600 A, we used 3,600 A for the diodes too, except for MBN3600E17F (used 3,000 A). The losses curves as a function of the current for the IGBTs are concave so the actual losses are lower than our estimate. For the diodes, the curve is convex, so, at a lower current, the losses are above our estimate (at higher current–lower), and we increase the safety margin at high current where it matters. Now we can calculate the losses using the formulas for carrier frequency of 500 and 2,000 Hz. All three devices have similar characteristics; their losses are within ±10% of each other for the operating conditions of interest to us. Furthermore, it appears that the Infineon device has the best losses out of all of them. However, it is a new device, and its data sheet is marked “preliminary” (the same is true for the Hitachi IGBT), so we have to wait to see if the specs will be confirmed; additionally, its present cost is much higher.

Table S1  IGBT parameters. ABB 5SNA3600E170300

Parameter

Hitachi MBN3600E17F

Infineon FZ3600R17HP4

VthIGBT (V)

1

1.03

0.933

rdynIGBT (mOhm)

0.583

0.375

0.417

VthD (V)

1

0.966

0.833

rdynD (mOhm)

0.208

0.375

0.229

EswIGBT (J)

3

3.7

2.8

IswIGBT (kA)

3.6

3.6

3.6

EswD (J)

1.5

1.17

1.4

3.6

3

3.6

IswD (kA) AC Current (A pk) 1,900 2,530

Carrier Frequency (Hz)

Losses (kW)

500

1.547

1.461

1.331

2,000

2.681

2.747

2.39

500

2.288

2.107

1.941

2,000

3.798

3.82

3.35

633

1,000

0.527

0.548

0.477

1,267

1,000

1.169

1.178

1.038

1,900

1,000

1.925

1.889

1.684

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rectifier needs to withstand the short-circuit current for at least 85 ms, preferably 100–250 ms. For our case, with a 3-MW rectifier with a secondary voltage of 486 Vac and a rated current of 3.56-kA ac, we can consider using a transformer with a high-value short-circuit impedance of 10–15% to reduce the short-circuit current. At 10% impedance (the worst case), the steady-state short-circuit current will be 35.6-kA ac. With six IGBTs in parallel, the current through each diode will be 5.93-kA ac, or 8.4 kA pk (assuming ideal current sharing). ABB provides data on a 3,600 -A 1,700 -V IGBT (5SNA3600E170300) diode’s surge withstanding capability [8, Section 3.1.1]. Using thermal model parameters and the diode parameters, we estimated the peak overheat temperature reach of 210 °C for the specified surge current of 20 kA pk at 50 Hz, and 154 °C for the specified surge current of 13.5 kA pk at 16.7 Hz. Using the same IGBT parameters, we got only a 90 °C overheat for half wave 60-Hz current with 8.4 kA pk (short circuit with six IGBTs in parallel). Therefore, we have plenty of margins to withstand the IGBT rectifier short-circuit current for 250 ms. The margins increase even more if we increase the transformer impedance or go with eight IGBTs in parallel.

The IGBT rectifier must have an incoming ac voltage with peak value below the dc voltage. introduce a phase shift on the carrier frequency and practically eliminate both the carrier frequency harmonics and power frequency harmonics from the rectifier transformer primary [1]. We can fuse each bridge on the ac side: if an IGBT fails, we can disable its bridge and operate at reduced load with the remaining bridges. An additional benefit of the multiple bridges approach is the ability to connect bridges both in parallel and in series, thus easily achieving higher-voltage operation, e.g., for a 1,500-Vdc train system. These advantages come at a cost: each bridge needs to have its own current sensors and some controls, and the transformer cost with multiple secondary windings will also be higher. We can also consider a halfway approach: four bridges with two parallel devices in each bridge.

IGBT Rectifier Transformer Impedance There is a tradeoff in selecting the transformer impedance: on the one hand, we want to keep it high to reduce the carrier frequency harmonics and fault currents; on the other hand, we need to keep it low to reduce the secondary ac voltage needed to operate at unity power factor. The secondary voltage, V, is determined by an equation

Paralleling Devices or Multiphase System? We see that, for both overload and, to a smaller extent, short-circuit conditions, we need to parallel eight IGBTs to build a 3-MW IGBT rectifier with a 300% overload. We can achieve this either by using a three-phase rectifier with eight parallel devices or by having multiple secondarys’ windings connected to their respective bridges and then paralleling the bridges’ outputs. In the first case (paralleling devices), we have the simplest control circuit design and a simple transformer design. However, this approach has the following disadvantages. ■■ We need to fuse every IGBT to isolate the failing devices. The fuse selection might be difficult because the IGBTs can conduct a short-circuit current only for 10 μs, then they need to be shut down; the fuse must melt during this short time to isolate the failed device. An alternative approach is to place a fuse in the ac line, which removes the limitation on the fuse clearing time; however, we will need to shut down an IGBT in the same leg as the failed device, complicating the control. ■■ There is no way to reduce both dc and ac harmonics by creating a phase shift between the switching of parallel IGBTs: all parallel devices are switched together. ■■ There is the potential problem of unequal current sharing between parallel devices, both for conduction and switching. An alternative is to use multiple secondary windings and connect each winding to a separate bridge. Since each bridge is controlled independently, we can

SEPTEMBER 2014 | IEEE vehicular technology magazine



V = 1 + jx i ,  V0 i rated

where V and V0 are the secondary voltages at secondary current i and no-load secondary voltage, respectively; i and i rated are secondary currents, actual and rated; and x is the transformer impedance (%)

V = V0

1 +c x

i 2 . m i rated

For a 15% impedance transformer and 300% load current, we get a voltage increase factor of

V = 1 + (0.15 $ 3) 2 = 1.096.  V0

The 15% impedance looks like a good compromise; in general, we can consider the transformer impedance between 10 and 20%.

Switching Frequency The switching (or carrier) frequency presents another compromise: increasing the switching frequency reduces the carrier harmonics and transformer losses, but it increases the IGBT module switching losses. Since we have multiple IGBT bridges, we can introduce the phase shift between their switching and, thus, almost completely cancel the switching frequency from the rectifier



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All of these devices are specified to operate at a 150 °C junction temperature. transformer primary [1], [3, Section IV-B]. We can assume the switching frequency to be between 500 and 2,000 Hz.

IGBT Rectifier Losses The rectifier losses are important for two reasons: 1) the cooling cost (fans and heatsinks) and 2) lost energy costs. In addition to direct energy loss, we have the additional air conditioning cost to remove the power losses from the substation, especially for underground substations. To estimate the IGBT rectifier losses, we use the formulas from “IGBT Losses for Sinusoidal Current” and assume that we have eight IGBTs in parallel with switching frequency of 1,000 Hz. We calculate the losses for the IGBTs from three

manufacturers (ABB, Hitachi, and Infineon; see “IGBT Losses for Sinusoidal Current”) and used the average losses for the three brands as the IGBT module losses, repeating the calculations for the loads of 100%, 200%, and 300%.

Comparison with TCR The TCR also provides the capability to regulate the output voltage and, with an additional reversible bridge, can recuperate the breaking energy into the power grid (Figure 3). If we use a 100-mm SCR with 2,800-V ratings (e.g., ABB 5STP45N2800), we will need two devices in parallel for the forward bridge and a single device for the reverse bridge, with 36 devices in total. The SCR threshold voltage is 0.86 V, and the slope resistance is 0.07 mΩ. The total SCR losses are (we have two parallel bridges with two parallel devices in each)

DPSCR = 2 c VthSCR + rdynSCR I d m I d .  4

The results are shown in the last column of Table 2; the TCR losses are almost four times lower than the IGBT rectifier losses. Table 3 shows the results of the comparison of the IGBT rectifier and TCR.

Reverse Bridge

+ Y

-

Figure 3 A reversible 12-pulse TCR. Table 2 The IGBT rectifier and TCR losses. IGBT Module Losses

Total IGBT Losses

Total TCR Losses

Load

I(AAC)

100%

633

0.517 kW

25 kW

7 kW

200%

1,267

1.128 kW

54 kW

15 kW

300%

1,900

1.833 kW

88 kW

22 kW

Table 3 The IGBT rectifier versus the TCR. Parameter

IGBT Rectifier

TCR

Reactive power (power factor)

Excellent

Moderate

Harmonics

Low

Moderate

Constant dc voltage range

Excellent

Good

Fault current limiting

No

Yes

Power losses

High

Low

Acoustic noise

Higher

Lower

Cost

High

Low

Power Quality Comparison We will consider four characteristics: power factor, harmonics, constant dc voltage regulation range, and fault current limiting. Power factor (or actually reactive power consumption) is an important parameter directly affecting voltage regulation in the power grid. The IGBT rectifier can operate with zero reactive power (unity power factor). The TCR has to consume reactive power; at a high load (when it matters), the TCR’s reactive power is close to that of a diode rectifier [5]. The ac harmonics of the TCR are higher than those of a diode rectifier because of the delayed SCR firing. However, at a high current, the firing angle is minimal and the TCR harmonics are similar to those of the diode rectifier [5]. Both the calculations and actual measurements show the 12-pulse TCR ac voltage harmonics to be below IEEE 519 [13]. In addition, no harmonics-related problems were reported in the TCR installations. The dc harmonics of the TCR are lower than those of the diode rectifier due to the big output capacitor filter. The IGBT rectifier has low harmonics, both ac and dc. The constant voltage regulation range for a TCR is a tradeoff with the reactive power consumption. The practical range for the constant voltage range is 130–200% load. The IGBT rectifier can provide up to 300% load at a constant voltage with zero reactive power. The fault current can be limited by the TCR, thus facilitating the dc breaker current interruption and increasing the dc breaker service life; the IGBT rectifier cannot limit the fault current.

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The IGBT rectifier is superior in the first three categories: it can operate with unity power factor over the entire range of the 300% load, the ac harmonics are low, and it can keep the constant dc voltage over the load 0–300%. The advantages of the TCR cannot match all of those characteristics, but it is adequate for most traction applications. The advantages of the TCR are its limiting fault current and improvement of the dc breaker service life.

The application of an IGBT rectifier for traction can be justified only for special cases where the power grid is weak.

Author Information Vitaly Gelman received the M.S.E.E. from Moscow Power Engineering University in 1976. He designed variable frequency drives from 1976 to 1982, first in Moscow and from 1979 at Ramsey Controls in New Jersey. He worked for ABB in New Jersey from 1982 to 1983 designing and troubleshooting high-power rectifiers. Since 1984, he has worked for VG Controls developing various controllers and protection relays for traction and other heavy industrial applications. He is also a consultant for traction authorities and traction equipment manufacturers.

Acoustic Noise Pollution We expect the IGBT rectifier to have much higher acoustic noise for two reasons: intensive cooling and higher switching frequency. The noise pollution is important because many traction power substations are located in residential areas. The IGBT rectifier losses are a few times higher than the TCR’s; to remove the additional heat, we need to run proportionally more air through the heatsinks. This means higher air speed and higher noise. The IGBT switching frequency in the range of 500–1,000 Hz generates acoustic noise in the transformer windings with double frequency (1,000–2,000 Hz) that is in the maximum sensitivity range of human ears.

References

[1] H. Kielgas and R. Nill, “Converter propulsion systems with threephase induction motors for electric traction vehicles,” IEEE Trans. Ind. Applicat., vol. IA-16, no. 2, pp. 222–233, 1980. [2] L. Wang, G. Zhang, M. Shen, H. Quan, and Z. Liu, “A novel traction supply system for urban rail transportation with bidirectional power flow and based on PWM rectifier,” in Proc. Int. Conf. Energy Environment Technology, 2009, pp. 40–43. [3] X. Lu, Z. Liu, L. Wang, and M. Shen, “On the characteristics of a novel traction power supply system based on three-level voltage-source PWM rectifier,” in Proc. IEEE Vehicle Power Propulsion Conf., Harbin, China, Sept. 3–5, 2008, pp. 1–4. [4] A. Steimel, “Electrical railway traction in Europe,” IEEE Ind. Applicat. Mag., vol. 6, no. 2, pp. 6–17, Nov./Dec. 1996. [5] V. Gelman, “Thyristor controlled rectifiers (wTCR) for traction— Problems and solutions,” in Proc. 3rd Int. Conf. Electric Power Energy Conversion Systems, 2013, pp. 1–6. [6] H.-G. Eckel, M. M. Bakran, E. U. Krafft, and A. Nagel, “A new family of modular IGBT converters for traction applications,” in Proc. European Conf. Power Electronics Applications, Dresden, Germany, 2005, pp. 1–10. [7] Webra. (2013). Webra catalog. [Online]. Available: http://www.webra. se/HS_katalog2013.pdf [8] ABB. (2011, Mar.). Surge currents for IGBT Diodes. 5SYA2058-02 [Online]. Available: http://www.5scomponents.com/pdf/currents_for_ igbt_diodes.pdf [9] ABB. (2012, May). Applying IGBTs. 5SYA2053-04. [Online]. Available: http:// search-ext.abb.com/library/Download.aspx?DocumentID=5SYA2053-01 &LanguageCode=en&DocumentPartID=&Action=Launch [10] ABB. ABB Spec 5SYA 1414-05 08-2013. 5SNA 3600E170300. [Online]. Available: http://search.abb.com/library/Download.aspx?DocumentID=5SYA%20 1414-06&LanguageCode=en&DocumentPartId=&Action=Launch [11] Hitachi. Spec.No.IGBT-SP-10024 R0 P1. MBNF3600E17F. [Online]. Available:  http://www.hitachi-power-semiconductor-device.co.jp/ en/product/igbt/pdf/mbn3600e17f.pdf [12] Infineon. FZ3600R17HP4_B2. [Online]. Available: http://www.infineon. com/dgdl/ds_fz3600r17hp4_2_2_de-en.pdf?folderId=db3a304412b4 07950112b4095b0601e3&fileId=db3a30432313ff5e01235601c5db1610 [13] IEEE Recommended Practices and Requirements for Harmonic Control  in Electrical Power Systems, IEEE Standard 519, 1992.

Cost Comparison To compare the cost of an IGBT rectifier and a TCR, let us use the same 3-MW rectifier built with either 48 IGBTs or a reversible 12-pulse TCR built with 36 high-current SCRs (5,080 A, 2,800 V). The TCR has two devices in parallel for forward bridges and a single device for reverse bridges. At a price of US$1,300 for an IGBT and US$700 for an SCR, we get a total IGBT cost of US$1,300 # 48 = US$62,400, and the total SCR cost is US$25,200. Assuming that the converter cost is proportional to the power semiconductor cost, the IGBT rectifier is at least twice as expensive as the TCR.

Conclusions The IGBT rectifier offers many advantages for traction applications, such as inherent energy recuperation, low harmonics, and unity power factor. However, compared with a TCR, it has four times higher losses with related expenses on the cooling system and 2.5 times higher power semiconductor costs, although the cost difference most likely will decrease in the future. At the present time, the application of an IGBT rectifier for traction can be justified only for special cases where the power grid is weak and has a limited capacity to absorb the reactive power and harmonics.

Acknowledgments I would like to thank Prof. Bih-Yuan Ku of the National Taipei University of Technology, Taiwan for bringing the subject of using IGBTs in traction rectifiers to my attention.

SEPTEMBER 2014 | IEEE vehicular technology magazine



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