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Mar 26, 2014 - Byung-Moon Han, Senior Member, IEEE, Nam-Sup Choi, and Jun-Young Lee, Member, IEEE. Abstract—This paper proposes a new ...

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New Bidirectional Intelligent Semiconductor Transformer for Smart Grid Application Byung-Moon Han, Senior Member, IEEE, Nam-Sup Choi, and Jun-Young Lee, Member, IEEE

Abstract—This paper proposes a new bidirectional intelligent semiconductor transformer (BIST) for the smart distribution system and smart grid. The proposed BIST consists of high-voltage high-frequency ac/dc converter, bidirectional low-voltage dc/dc converter, and hybrid-switching dc/ac inverter. It features 1) inputto-output isolation with a high-frequency transformer; 2) bidirectional power flow; 3) small size and light weight; 4) capability of compensating voltage sag and/or swell; and 5) realization of three-phase structure based on single-phase module. The operational feasibility of proposed transformer was verified not only by computer simulation with PSCAD/EMTDC software but also by a hardware prototype with rating of 1.9 kV/127 V, 2 kVA, allowing a three-phase transformer of 3.3 kV/220 V, 6 kVA with three-phase construction. Index Terms—Bidirectional dc/ac converter, bidirectional intelligent semiconductor transformer (BIST), high-voltage ac/dc rectifier, hybrid-switching, PSCAD/EMTDC.

I. INTRODUCTION ONVENTIONAL transformer composed of coil and iron core can change only the magnitude of the ac voltage and the quality of supplying power is totally dependent on that of the input power. So, it cannot be applicable for the smart grid, in which the magnitude and frequency of the operation voltage are various and high-quality power is required. Intelligent semiconductor transformer or solid-state transformer was proposed by EPRI to replace the conventional transformer in railway systems and substations, in which light weight is mandatorily required [1]. Recently, EPRI has reported 100 kVA single-phase semiconductor transformer named intelligent universal transformer for distribution automation [2]. Intelligent semiconductor transformer can easily offer small size and light weight because it operates at much higher frequency with reduction of the magnetic component. It can supply not only the dc

C

Manuscript received June 19, 2013; revised July 18, 2013 and September 6, 2013; accepted September 20, 2013. Date of current version March 26, 2014. The work was supported in part by the Basic Science Research Program through NRF funded by the Ministry of Education, Science, and Technology under Grant 2012-001096 and in part by the Human Resources Development of the Korea Institute of Energy Technology Evaluation and Planning grant funded by the Korea Government Ministry of Knowledge Economy under Grant 20114010203030. Recommended for publication by Associate Editor P. Tenca. B.-M. Han and J.-Y. Lee are with the Department of Electrical Engineering, Myongji University, Yongin-Si, Gyeonggi 449-728, Korea (e-mail: [email protected]; [email protected]). N.-S. Choi is with the Department of Electrical, Electronic Communication and Computer Engineering, Chonnam National University, Yeosu 550-749, Korea (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TPEL.2013.2284009

Fig. 1.

Configuration of the BIST.

power, but also high-quality ac power to the customer by compensating the voltage sag, swell, and harmonics. So, it can be utilized for implementing the smart distribution system and the microgrid [3]–[5]. Various kinds of intelligent semiconductor transformers were already proposed. However, since the power flow in these transformers is unidirectional, it is not properly applicable for the dc distribution and microgrid [1], [2], [6]–[10]. One can find some studies on the semiconductor transformer topologies with bidirectional power flow capability [11]–[23]. In [11] and [12], bidirectional power flow can be achieved but the power factor is not controlled. The topology in [13] can compensate sag/swell voltage; however, it employs heavy and bulky line-frequency transformer for isolation. The semiconductor transformer in [14] has not only the bidirectional power flow functions but also voltage sag compensation where high-frequency dc/dc power conversion is employed. The circuit configuration in [14], however, shows too many active switching device counts, at least 18 IGBTs for implementing singlephase module. In [15]–[23], three-stage structure comprised of ac/dc converter, dual-active-bridge dc/dc converter, and inverter. These topologies provide power factor correction and reactive power compensation, but they suffer from heavy turn-off loss in dc/dc stage and complex control for voltage balancing. This paper proposes a new bidirectional intelligent semiconductor transformer (BIST) for the smart distribution system and microgrid. The proposed BIST consists of high-voltage part and low-voltage part, whose configuration is shown in Fig. 1. The high-voltage part is composed of several half-bridge ac/dc converters connected in series through high-frequency transformers to cope with high input voltage, while the low-voltage part is composed of bidirectional half-bridge dc/dc converter and dc/ac PWM inverter. In the prototype BIST, the input voltage on the high-voltage side is 1900 V and the output voltage on the low-voltage side is 127 V, in which the primary and secondary dc-link voltages are 320 V and 700 V, respectively. A three-phase 3.3 kV/220 V transformer can be built using three units of 1.9 kV/127 V single-phase module.

0885-8993 © 2013 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications standards/publications/rights/index.html for more information.

HAN et al.: NEW BIDIRECTIONAL INTELLIGENT SEMICONDUCTOR TRANSFORMER FOR SMART GRID APPLICATION

Fig. 3.

Fig. 2.

Bidirectional high-frequency ac/dc converter.

II. PROPOSED SEMICONDUCTOR TRANSFORMER A. High-Voltage Part Fig. 2 shows the power circuit of ac/dc rectifier, which converts single-phase ac voltage of 1900 V into full-bridge-rectified waveform of 320 V. The ac/dc converter has high-frequency transformers, which offer high-frequency resonance and input– output isolation. The input side works under high voltage, while the output side works under low voltage. So, the input side is designed with three half-bridge modules connected in series, in which two IGBT units are connected in series in the reverse direction. The output side is designed with three half-bridge modules connected in shunt. Whole system operates in bidirectional high-frequency resonance mode under a fixed frequency with 50% duty ratio to reduce system size and switching loss. Because the resonant stage is basically an LLC converter, the input-to-output gain of each resonant converter, that is defined by vlink /|vac1 |, is determined only by its transformer turns-ratio nT if the resonant frequency fr is equal to the switching frequency fsr [24], where vac1 is the input voltage of each resonant stage and it is equal to vac /3. Since the input and output filter capacitors of Cin and CL are much larger than Cr and parasitic capacitances of switches are much smaller than Cr , the resonant frequency fr , which is equal to fsr , is calculated as 1/[2π(2Lr Cr )0.5 ] with resonant inductor Lr and two resonant capacitors of Cr . Fig. 3 shows the switching pulses for each switch in a single-module of the bidirectional high-frequency ac/dc converter according to the polarity of the ac input voltage. The gating pulses for each switch are generated with same pattern regardless of the direction of power flow. Before explanation, it is assumed that the magnetizing inductance Lm is infinity. Mode 1: The direction of power flow is forward and the polarity of input voltage is positive as shown in Fig. 4(a). In the first stage, the primary current flows through the transistor in

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Driving signals of high-voltage resonant stage.

M1 and the diode in M2 when M1 turns ON. At this instance, the secondary current flows through diode in M5 . In the next stage, the primary current flows through the transistor in M3 and the diode in M4 when M3 turns ON. At this instance, the secondary current flows through the diode in M6 . Mode 2: The direction of power flow is forward and the polarity of input voltage is negative as shown in Fig. 4(b). In the first stage, the primary current flows through the transistor in M2 and the diode in M1 when M2 turns ON. At this instance, the secondary current flows through diode in M6 . In the next stage, the primary current flows through the transistor in M4 and the diode in M3 when M4 turns ON. At this instance, the secondary current flows through the diode in M5 . Mode 3: The direction of power flow is backward and the polarity of input voltage is positive as shown in Fig. 4(c). In the first stage, the secondary current flows through transistor in M5 when M5 turns ON. At this instance, the primary current flows through the diode in M1 and the transistor in M2 . In the next stage, the secondary current flows through the transistor in M6 when M6 turns ON. At this instance, the primary current flows through the diode in M3 and the transistor in M4 . Mode 4: The direction of power flow is backward and the polarity of input voltage is negative as shown in Fig. 4(d). In the first stage, the secondary current flows through transistor in M6 when M6 turns ON. At this instance, the primary current flows through the transistor in M1 and the diode in M2 . In the next stage, the secondary current flows through the transistor in M5 when M5 turns ON. At this instance, the primary current flows through the transistor in M3 and the diode in M4 . B. Low-Voltage Part The low-voltage part consists of the dc/dc converter and the dc/ac inverter connected in cascade as shown in Fig. 5. The dc/dc converter changes the full-bridge rectified waveform of 320 V into the constant dc voltage of 700 V and the dc/ac inverter changes the constant dc voltage of 700 V into the single-phase ac voltage of 127 V. The dc/dc converter and dc/ac inverter use a hybrid switch with IGBT and MOSFET connected in parallel. The dc/dc converter and dc/ac inverter are composed of two half-bridges connected in cascade. The dc/dc converter operates to control the power factor and the dc-link voltage, while the dc/ac inverter operates to control the output voltage. As the switching frequency in IGBT increases, the switching loss increases due to tail-current, which critically reduces the

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Fig. 5.

Configuration of the bidirectional dc/ac converter.

Fig. 6.

Gating method for hybrid switch.

system efficiency. In order to improve this switching loss, a MOSFET is connected in parallel to implement a hybrid switch. Fig. 6 shows how to supply the gating signal to the hybrid switch. The MOSFET turns ON a few microseconds ahead when the IGBT switch turns OFF. After the MOSFET turns ON, the IGBT turns OFF immediately and the MOSFET turns OFF at the instant that the IGBT is originally to turn OFF. Hybrid switching offers reduction of recovery loss due to tail-current. If a diode is connected in series with MOSFET, MOSFET destruction due to counter electromotive force can be protected. If resistance is connected in parallel with diode, ringing phenomenon can be reduced. C. Zero-Voltage-Switching (ZVS) Operation

Fig. 4. Current path for each operation mode in the ac/dc rectifier. (a) Mode 1: forward power flow_ positive input voltage. (b) Mode 2: forward power flow_ negative input voltage. (c) Mode 3: Reverse power flow_ positive input voltage. (d) Mode 4: Reverse power flow_ negative input voltage.

Since the magnetizing inductance Lm cannot have infinity value in real transformer, operational modes are somewhat different from that explained in Fig. 4 and it is helpful to achieve soft-switching of switches. All modes in Fig. 4 have same ZVS operation so that operational mode analysis is explained based on mode 1 of forward power flow with positive input voltage. Fig. 7 shows ZVS operation in mode 1 when the magnetizing inductance is not infinity. Before explanation, it is assumed that the resonant frequency fr is equal to the switching frequency fsr . Mode A: The magnetizing current charges collector–emitter capacitance of M3 Cce,M 3 and discharges collector–emitter capacitance of M1 Cce,M 1 . Thus, collector–emitter voltage of M3 vce,M 3 increases and collector–emitter voltage of M1 vce,M 1 decreases. When vce,M 3 exceeds the source voltage or vce,M 1

HAN et al.: NEW BIDIRECTIONAL INTELLIGENT SEMICONDUCTOR TRANSFORMER FOR SMART GRID APPLICATION

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Fig. 8. Simulated waveform of ZVS operation in mode 1 of forward power flow with positive input voltage.

Fig. 9. Operational waveform of the resonance converter during half of the line cycle.

Fig. 7. ZVS operation in mode 1 of forward power flow with positive input voltage. (a) Mode A. (b) Mode B. (c) Mode C.

crosses zero, the body diode of M1 starts to conduct the current. At this instant, the mode A starts. During mode A, the secondary resonant current iL r s begins to flow with resonant manner and it is divided into half and each half currents flow through the two resonant capacitors as shown in Fig. 7(a). The primary resonant current iL r p is the sum of the magnetizing current iM and secondary resonant referred to the primary side, which can be expressed as iL r s /nT . Since iL r s /nT is smaller than iM , the primary resonant current iL r p is negative so that it flows through body diode in M1 and transistor in M2 from the negative peak value of the magnetizing current—IM ,pk . This mode continues until iL r s /nT is equal to iM . Mode B: After iL r s /nT is greater than iM , iL r p flows through transistor in M1 and body diode in M2 . Since the resonant

frequency fr is equal to the switching frequency fsr , iL r s is nearly reduced to zero at the end of this mode. Mode C: When M1 is turned OFF, only the magnetizing current remains on the primary side and it can be assumed that the magnetizing current is constant because mode C is a short dead-time period. As shown in Fig. 7(c), all switches of M1 and M3 are in turn-off state so that they can be modeled as their collector–emitter capacitances of Cce,M 1 and Cce,M 3 . Accordingly, the magnetizing current flows through two paths of Lm , Cin , collector–emitter capacitance of M1 Cce,M 1 , body diode in M2 and Lm , Cin , transistor in M4 , collector–emitter capacitance of M3 Cce,M 3 . Therefore, Cce,M 1 is charged from zero to vac1 by the half of the magnetizing current and Cce,M 3 is discharged from vac1 to zero by the half of the magnetizing current. If mode C operation is completed before M3 is turned ON, ZVS of M3 can be accomplished. ZVS of M1 has the same manner as that of M3 . Fig. 8 is computer simulation of ZVS operation in mode 1 of forward power flow with the positive input voltage. It shows that the simulation waveforms are similar to those shown in Fig. 7. D. Transformer Design Fig. 9 is the operational waveform of the LLC resonance converter during half of the line cycle. To analyze the proposed

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circuit, it is necessary to derive the resonant waveform expressions. The input voltage vac1 and link voltage vlink during nth switching period can be written as vac1 [n] = Vac1,pk sin ωnTsr vlink [n] =

Vac1,pk | sin ωnTsr | nT

(Vac1,rm s /nT )2 Po

(2)

(3)

with the output power of each resonant converter Po . Referring to Fig. 9, the resonant current waveforms of both sides during nth switching period can be written as √ iLrp [n] = 2ILrp,rm s [n] sin(2πfsr (t − (n − 1)Tsr ) − φ[n]) (4) iLrs [n] =

πVac1,pk sin(ωnTsr ) sin(2πfsr (t − (n − 1)Tsr )). nT Rb (5)

The primary resonant current iLrp has a small delay to the secondary resonant current iLrs because of the magnetizing current iM . Since vac1 and vL are assumed to be constant during switching period, RMS current of iLrp and the delayed angle of each switching period can written as follows [25]:  n4T Rb2 Vac1,pk sin(ωnTsr ) π2 (6) + ILrp,rm s [n] = 2 2 2 nT Rb 128Lm fsr 2   Vac1,pk sin(ωnTsr )Tsr −1 √ (7) φ[n] = sin 8 2Lm ILrp,rm s [n] where Lm is the magnetizing inductance referred to the primary side. The IGBT collector–emitter capacitance Cce varies according to the magnitude of collector–emitter blocking voltage vce . The relationship can be expressed by the following equation: cce (vce ) = 

Cceo 1 + vce /Vo

Poff =

(1)

where ω means angular frequency of vac1 and Tsr is resonant converter switching period. nT means transformer turns-ratio. The effective resistor model Rb of secondary stage is simply written as Rb =

side during half of the line cycle TL can be obtained as

(8)

where Cceo is Cce at vce = 0 and Vo is the potential barrier that is dependent on the construction of the device. Referring to this equation, the output capacitance is strongly nonlinear and it is rapidly increasing as the blocking voltage decreases. Thus, the magnetizing inductance Lm used for ZVS in the primary switches should have a small value at a low line voltage, i.e., a low blocking voltage. However, too small Lm generates heavy conduction loss so that some tradeoffs are needed to obtain a minimum loss. To do this, switching and conduction losses of the primary switches should be calculated under some ZVS range. The turn-off switching loss of IGBT’s on the primary

kr 1  vac1 [n] 2Eoff (Im ,pk , VT ) TL n =1 VT

(9)

where Eoff (Im ,pk , VT ) is turn-off switching energy at the test voltages VT and Im ,pk , and kr is the total switching number of the resonant converter during half of the line cycle TL , which is ∗ calculated by int(TL /Tsr ). Fig. 9 shows ZVS start voltage Vac1 and it is defined as a minimum line voltage from which ZVS operation can be achieved. If the angle of ZVS start voltage is θ, the switching number from zero to θ becomes xn = int(kr θ/π). Because turn-on switching loss happens during θ, the loss can be written as follows: xn 1  vac1 [n] 2Eon (Im ,pk , VT ) (10) Pon = TL n =1 VT where Eon (Im ,pk , VT ) is turn-off switching energy at the test voltages VT and Im ,pk . Conduction losses on the primary side can be calculated with the resonant current expression of (4) and (6) and it can be written as follows: √ kr 2 2 Tsr  ILrp,rm s [n](Vce,sat1 + Von1 ) (11) Pcond = TL n =1 π where Vce,sat1 is the saturation voltage of IGBT and Von1 is on-drop voltage of the antiparallel diode of IGBT on the primary side. During the dead-time Tdead , the magnetizing current is used for the displacement current in the resonance capacitor and for the ZVS current of primary-side switch. As explained before, collector–emitter capacitance of IGBT varies according to the applied collector–emitter voltage. Unfortunately, the applied collector–emitter voltage is equal to the input voltage that has sinusoidal waveform so that there are many solutions of magnetizing inductance to meet ZVS. Furthermore, since collector– emitter capacitance is increased as the applied collector–emitter voltage is decreased, small magnetizing inductance is required to obtain a wide ZVS range. Therefore, some ZVS range which can be specified by ZVS start voltage Vac∗ should be determined considering power loss, circuit reliability, etc. After determining Vac ∗, the magnetizing inductance Lm for ZVS operation under ∗ can be obtained by (12) [25] the given ZVS start voltage Vac1 Lm =

Tdead ∗ )n2 ] + Cce2 (nT Vac1 T

∗ ) 16fsr [Cce1 (Vac1

(12)

where, Cce1 and Cce2 are collector–emitter capacitances of pri∗ and Lm mary and secondary switches, respectively. Using Vac1 calculated with (12), losses can be estimated from the loss equa∗ tions of (9)–(11). Then, Vac1 and Lm at the minimum loss are the optimal design points for the resonant converter. III. COMPUTER SIMULATION Computer simulation with PSCAD/EMTDC software was carried out to confirm the circuit operation and control performance of the proposed semiconductor transformer. Fig. 10 shows simulation results to check the operation of the proposed transformer under power flow reversal. The first graph shows the input voltage and current waveform. The second graph shows

HAN et al.: NEW BIDIRECTIONAL INTELLIGENT SEMICONDUCTOR TRANSFORMER FOR SMART GRID APPLICATION

Fig. 10.

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Simulation analysis in forward and reverse power flow.

the 50-kHz resonant current for ZVS when the power flow is reversed from forward to backward at 0.18 ms. The third graph shows the output voltage waveform with the full-bridge rectified voltage and current. The fourth graph shows the dc-link voltage, which maintains constant voltage of 700 V with negligible ripples. The fifth graph shows the output voltage and current, which are almost sinusoidal with negligible harmonics. Through the simulation results, it is confirmed that the proposed semiconductor transformer operates properly as analyzed with the theoretical approach. Fig. 11 shows simulation results to check the operation of the proposed semiconductor transformer under input voltage sag in forward power flow and reverse power flow. Fig. 11(a) shows the input voltage and current, rectified voltage and current, dc-link voltage, and output voltage and current when sag occurs in the forward power flow. The input current and rectified current slightly increase during sag to maintain same input power. The dc-link voltage is maintained with 700 V through the voltage control of the dc/dc converter. Fig. 11(b) shows the input voltage and current, rectified voltage and current, dc-link voltage, and output voltage and current when sag occurs in the reverse power flow. The input current and rectified current slightly increase during sag to maintain same power. The dc-link voltage is maintained with 700 V through the voltage control of dc/dc converter.

Fig. 11. Simulation analysis against input voltage sag. (a) Forward power flow. (b) Reverse power flow. TABLE I SPECIFICATIONS OF BIST

IV. EXPERIMENTAL SETUP AND RESULTS Hardware prototype with the specifications in Table I was built and tested to confirm the operation of the actual system. Fig. 12 shows the hardware structure of the proposed semiconductor transformer. Table II shows the switching device technical data for the proposed BIST. Fig. 13 is the loss calculation of ∗ to single IGBT on the primary side and ZVS start voltage Vac1 select the magnetizing inductance Lm with transformer turns∗ is ratio nT = 2. Turn-on losses are negligibly small but Vac1 drastically increased, which expands hard-switching region and generates heavy switching noise on the high-voltage primary

side due to hard-switching. It affects the operational reliability and device voltage stresses. In this design, we have selected Lm as 1 mH compromising these two factors. Fig. 14(a) and (b) are operational waveforms according to power flow directions. These figures show that the measured waveforms are almost identical to the waveforms in simulation results. Also, Fig. 14(c) shows measured switching waveform and its simulation result. To show the soft-switching operation, direct measurement of the switch current and switch voltage should be performed. However, measurement of the switch current is somewhat

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Fig. 12.

IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 8, AUGUST 2014

Hardware structure of BIST. TABLE II SWITCHING DEVICE TECHNICAL DATA FOR BIST

Fig. 14. Experimental results of hardware prototype. (a) Forward power flow. (b) Reverse power flow. (c) Measured and simulated switching waveforms of the high-frequency ac/dc converter under forward power flow. Fig. 13. Loss calculation of single IGBT on the primary side and ZVS start voltage V a∗c 1 for L m design.

difficult because of the implemented circuit structure and abnormal switch failure caused by dummy line inductance for measuring purpose. By comparing with simulation result, ZVS operation can be observed indirectly. The first waveform of the simulation result is collector–emitter voltages of M1 and M3 and resonant currents on the primary and secondary sides, which is

identical to experimental waveform, and second and third waveforms are collector–emitter voltage and switch current waveforms of M1 and M3 , respectively. Investigating the simulation result and measured waveform, it can be seen that ZVS operation of prototype hardware is successfully accomplished. Also, because the relatively small magnetizing inductance is used for

HAN et al.: NEW BIDIRECTIONAL INTELLIGENT SEMICONDUCTOR TRANSFORMER FOR SMART GRID APPLICATION

Fig. 15.

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Experimental waveforms for power flow reversal. Fig. 17.

Calculated loss analysis under bidirectional power flow.

flow and reverse power flow, respectively. In Fig. 16(a), the input current and the rectified current slightly increase during sag to maintain constant output power. The dc-link voltage is maintained at 700 V through the voltage control of the dc/dc converter. The output voltage and current are maintained as a constant value in the forward power flow regardless of the input voltage sag. In Fig. 16(b), the input current and the rectified current slightly increase during sag to maintain constant output power. The dc-link voltage is maintained at 700 V through the voltage control of the dc/dc converter. The output voltage and current are maintained as a constant value in the reverse power flow regardless of the input voltage sag. Finally, the measured efficiency has been recorded as 83.6% at the rated condition regardless of power flow directions. Fig. 17 is the calculated loss analysis under bidirectional power flow. It shows that switching loss is a dominant factor due to high-frequency operation of 50 kHz of the ac/dc converter and 20 kHz of the dc/ac converter. V. CONCLUSION

Fig. 16. Experimental waveforms against input voltage sag. (a) Forward power flow. (b) Reverse power flow.

design, phase delay of primary resonant current is somewhat large. Fig. 15 shows the waveforms to verify the operation of proposed semiconductor transformer when the power flow reversal occurs. The input current and the rectified current are reversed when the power flow is reversed. However, the dc-link voltage is maintained at 700 V through the voltage control of the dc/dc converter, even though some transient exists. Fig. 16 shows experimental results to check the operation of the proposed transformer under input voltage sag in forward power

In this paper, a new configuration of the BIST was proposed, which has rating of 1.9 kV/127 V, 2 kVA. The transformer consists of the high-voltage high-frequency ac/dc rectifier, and low-voltage dc/dc and dc/ac converters. The operational feasibility of the proposed transformer was verified by computer simulation with PSCAD/EMTDC software. Based on the simulation results, a hardware prototype with rating of 1.9 kV/127 V, 2 kVA was built and tested in the lab to confirm the feasibility of hardware implementation. Using three units of this transformer, a three-phase transformer with rating of 3.3 kV/220 V, 6 kVA can be built. The proposed transformer could be applicable for implementing the smart grid. REFERENCES [1] “Feasibility assessment for intelligent universal transformer,” EPRI, Palo Alto, CA, USA, EPRI Rep. TR-1001698, Dec. 2002. [2] M. Arindam, S. Ashok, G. Mahesh, B. Simon, and D. Shoubhik, “Intelligent universal transformer design and applications,” in Proc. 20th Int. Conf. Exhib. Elect. Distrib., 2009, pp. 1–7.

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[3] X. She, R. Burgos, G. Wang, F. Wang, and A. Q. Huang, “Review of solid state transformer in the distribution system: From components to field application,” in Proc. IEEE Energy Convers. Congr. Expo., 2012, pp. 4077–4084. [4] S. Falcones, R. Ayyanar, and X. Mao, “A DC–DC multiport-converterbased solid-state transformer integrating distributed generation and storage,” IEEE Trans. Power Electron., vol. 28, no. 5, pp. 2191–2203, May 2013. [5] G. Oritz, D. Bortis, J. W. Kolar, and O. Apledoorn, “Soft-Switching techniques for medium-voltage isolated bidirectional DC/DC converters in solid state transformers,” in Proc. 38th Annu. Conf. IEEE Ind. Electron. Soc., 2012, pp. 5233–5240. [6] L. Heinemann and G. Mauthe, “The universal power electronics based distribution transformer, an unified approach,” in Proc. IEEE 32nd Annu. Power Electron. Spec. Conf., 2001, pp. 504–509. [7] J. W. Merwe, T. Du, and H. Mouton, “The solid-state transformer concept: A new era in power distribution,” in Proc. APRICON’90, 2009, pp. 1–6. [8] E. Ronan, Jr., S. Sudhoff, S. Glover, and D. Galloway, “A power electronicbased distribution transformer,” IEEE Trans. Power Deliv., vol. 17, no. 2, pp. 537–543, Apr. 2002. [9] J. S. Lai, A. Maina, A. Mansoor, and F. Goodman, “Multilevel intelligent universal transformer for medium voltage applications,” in Proc. Ind. Appl. Conf., 2005, pp. 1893–1899. [10] T. Zhao, L. Yang, J. Wang, and A. Q. Huang, “270 kVA solid state transformer based on 10 kV SiC power devices,” in Proc. IEEE Elect. Ship Technol. Symp., 2007, pp. 145–149. [11] M. Kang, P. Enjeti, and I. Pitel, “Analysis and design of electronic transformers for electric power distribution system,” IEEE Trans. Power Electron., vol. 14, no. 6, pp. 1133–1141, Nov. 1999. [12] H. Qin and J. W. Kimball, “AC-AC dual active bridge converter for solid state transformer,” in Proc. IEEE Energy Convers. Congr. Expo., 2009, pp. 3039–3044. [13] E. C. Aeloiza, P. N. Enjeti, L. A. Moran, and I. Pitel, “Next generation distribution transformer: To address power quality for critical loads,” in Proc. IEEE Annu. Power Electron. Spec. Conf., 2003, pp. 1266–1271. [14] Z. Tiefu, Z. Jie, S. Bhattachaya, M. E. Baran, and A. Q. Huang, “An average model of solid state transformer for dynamic system simulation,” in Proc. IEEE Power Energy Soc., 2009, pp. 1–8. [15] X. She, A. Q. Huang, S. M. Lukic, and M. E. Baran, “On integration of solid-state transformer with zonal DC microgrid,” IEEE Trans. Smart Grid, vol. 2, no. 2, pp. 975–985, Jun. 2012. [16] H. Fan and H. Li, “High-Frequency transformer isolated bidirectional DC–DC converter modules with high efficiency over wide load range for 20 kVA solid-state transformer,” IEEE Trans. Power Electron., vol. 26, no. 12, pp. 3599–3608, Dec. 2011. [17] J. Shi, W. Gou, H. Yuan, T. Zhao, and A. Q. Huang, “Research on voltage and power balance control for cascaded modular solid-state transformer,” IEEE Trans. Power Electron., vol. 28, no. 4, pp. 1523–1532, Apr. 2013. [18] T. Zhoa, G. Wang, S. Bhattacharya, and A. Q. Huang, “Voltage and power balance control for a cascaded H-Bridge converter-based solid-state transformer,” IEEE Trans. Power Electron., vol. 26, no. 4, pp. 1154–1166, Apr. 2011. [19] S. H. Hwang, X. Liu, J. M. Kim, and H. Li, “Distributed digital control of modular-based solid-state transformer using DSP+FPGA,” IEEE Trans. Ind. Electron., vol. 60, no. 2, pp. 670–680, Feb. 2013. [20] X. Liu, H. Li, and Z. Wang, “A Start-Up scheme for a three-stage solidstate transformer with minimized transformer current response,” IEEE Trans. Power Electron., vol. 27, no. 12, pp. 4832–4836, Dec. 2012. [21] X. She, A. Q. Huang, and G. Wang, “3-D space modulation with voltage balancing capability for a cascaded seven-level converter in a solid-state transformer,” IEEE Trans. Power Electron., vol. 26, no. 12, pp. 3778– 3789, Dec. 2011. [22] G. Oritz, H. Uemura, D. Bortis, J. W. Kolar, and O. Apledoorn, “Modeling of soft-switching losses of IGBTs in high-power high-efficiency dualactive-bridge DC/DC converters,” IEEE Trans. Electron. Devices, vol. 60, no. 2, pp. 587–597, Feb. 2013. [23] G. Oritz, M. Leibl, J. W. Kolar, and O. Apledoorn, “Medium frequency transformers for solid-state-transformer applications—design and experimental verification,” in Proc. 10th Int. Conf. Power Electron. Drive Syst., 2013, pp. 1285–1290.

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[24] J. Y. Lee, Y. S. Jeong, and B. M. Han, “An isolated DC/DC converter using high-frequency unregulated LLC resonant converter for fuel cell applications,” IEEE Trans. Ind. Electron., vol. 58, no. 7, pp. 2926–2934, Jul. 2011. [25] B. Lu, W. Liu, Y. Liang, F. C. Lee, and J. D. van Wyk, “Optimal design methodology for LLC resonant converter,” in Proc. 21st Annu. IEEE Appl. Power Electron. Conf. Expo., 2006, pp. 533–538.

Byung-Moon Han (SM’00) received the B.S. degree in electrical engineering from Seoul National University, Seoul, Korea, in 1976, and the M.S. and Ph.D. degrees from Arizona State University, Phoenix, AZ, USA, in 1988 and 1992, respectively. He was with the Westinghouse Electric Corporation as a Senior Research Engineer in the Science & Technology Center, Pittsburg, PA, USA. He is currently a Professor in the Department of Electrical Engineering, Myongji University, Seoul, Korea. His current research interests include power electronics applications for FACTS, custom power, distributed generation, and microgrid.

Nam-Sup Choi received the B.S. degree in electrical engineering from Korea University, Seoul, Korea, in 1987, and the M.S. and Ph.D. degrees in electrical engineering from the Korea Advanced Institute of Science and Technology, Taejeon, Korea, in 1989 and 1994, respectively. He is currently a Professor in the Division of Electrical, Electronic Communication and Computer Engineering, Chonnam National University, Yeosu, Korea. His research interests include the modeling and analysis of power conversion systems, matrix converters and multilevel converters for renewable energy systems, and microgrid applications.

Jun-Young Lee (M’12) received the B.S. degree in electrical engineering from Korea University, Seoul, Korea, in 1993, and the M.S. and Ph.D. degrees in electrical engineering from Korea Advanced Institute of Science and Technology, Taejon, Korea, in 1996 and 2001, respectively. From 2001 to 2005, he worked as a Manager in Plasma Display Panel Development Group, Samsung SDI where he was involved in circuit and product development. From 2005 to 2008, he worked as a Faculty Member in the School of Electronics and Computer Engineering, Dankook University. In 2008, he joined the School of Electrical Engineering, Myongji University, Seoul, as an Assistant Professor. His research interests include the areas of power electronics which include converter topology design, soft-switching techniques, display driving system, and battery charger system.

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