Series-parallel load-resonant converter for controlled ... - IEEE Xplore

Series-parallel load-resonant converter for controlled-current arc welding power supply H. Polloclk J.O. Flower

Indexing terms: Power supplies, Resonant converter, Arc welding supply, Power control, DC power supply, Constant current power supply

Abstract: An arc welding power supply requires the ability to limit the load current under short circuit conditions and limit the load voltage under open circuit conditions. The series-parallel loadresonant converter is a circuit configuration which can achieve this. The potential for high frequency operation in these converters minimises the size of the magnetic components in the converter and reduces output current ripple. The paper presents the theoretical and practical aspects of the design and construction of a seriesparallel load-resonant converter, for use as a controlled current arc welding power supply. A converter with an output voltage of 25V and output current of 200A operating at 85kHz is designed and constructed. A novel method of control of the resonant converter is described. Output power control is achieved by using an active rectifier modulating the D C supply voltage to the converter. Practical results of the converter in operation supplying 200A to a low inductance load are presented.

1

Introduction

I. I Requirements of an arc welding power supply Fig. 1 shows a block diagram of a present welding

size of both of these magnetic components depends on the frequency of operation of the circuit. At 50 or 60 Hz supply frequencies they can become large, especially as the power requirements increase. With the use of electronics, operation at frequencies of several kHz significantly reduces the size of the transformer and increases the speed at which the circuit can respond to current and voltage fluctuations. There are several different conditions which arise during the welding process, such as starting the arc, short circuit of the arc and open circuit of the arc. Ideal arc starting involves a rapid initial current rise to establish the arc quickly [4] and a quick response once the arc has been established to reduce the current levels to those required for welding. The rapid initial rise of current is most commonly achieved by bringing the welding electrode into contact with the workpiece. If the current rise, once the arc is established, is not controlled burnback occurs as the arc creeps up the tip of the electrode and excess welding wire is deposited. A fast arc ignition is therefore important. This is predominantly determined by the time constant of the power supply circuit and the accuracy with which the arc current can be monitored and controlled. Under normal welding conditions the electrode can come into contact with the workpiece and the arc becomes a short circuit. The power supply must limit the load current under these conditions. The smaller the short circuit load current, the less spatter occurs [5-71.

power supply [l]. The power supply is supplied from the AC mains through an isolation transformer, a rectifier which may be controlled if required and a smoothing inductor to the welding arc [2, 31.

(7)

cGGJ-J-n---q[ transformer

(optional)

Fig. 1 Block diagram of welding power supply 10

An isolation transformer is present because the workpiece forms part of the electrical circuit and must be isolated from the power supply for safety reasons. A smoothing inductance is present to reduce the current ripple on the supply current to the welding arc. The 0IEE, 1996 IEE Proceeding,s online no. 19960257 Paper first received 24th July 1995 and in revised form 5th December 1995 The authors are with The Department of Engineering, University of Warwick, Coventry CV4 IAL,UK IEE Proc.-Electr. Power A p p l . , Vol. 143, No. 3, May 1996

I I I I I 150 200 250 300 350 weld current,A Current and voltage curves for consumable and non-consumable I

50

Fi 2

100

weBing

If the arc is extinguished during the welding process an open circuit occurs across the power supply. The 21 1

Table 1: Explanation of Fig. 2 Curve

Welding t Y Pe

Electrode type

Polarity of Electrode

Shielding gas

Arc length (inches)

(1)

TIG

AI

+ve

0.16

(2)

TIG

cu

Ar

+ve

0.4

(3)

TIG

cu

Ar

+ve

0.3

(4)

TIG

cu

Ar

+ve

0.025

(5)

TI G

AI

He

+ve

0.08

(6)

TI G

AI

He

+ve

0.16

(7)

MIG

AI

Ar

8.3 (per minute)

(8)

MIG

AI

Ar

8.3 (Der minute)

performance of an arc welding power supply under these conditions is defined in the British Safety Standard 638 [8]. The no load voltage cannot exceed a DC value of 113V or an AC value of 68V peak (48V RMS). A voltage reducing device must automatically reduce the output voltage to these levels when the resistance of the external circuit exceeds 200R. The voltage must be reduced within 0.3 seconds. The representative current and voltage curves for TIG and MlG welding under normal conditions are shown in Fig. 2. These curves were obtained from experimental evaluation by Cook and Merrick [9]. See Table 1 for explanation. From this data a model of the welding arc was created which represented the arc as a voltage source and series resistance. A model based on this has been used for all the analysis and development of the new power supply described in this paper.

cantly reduces the magnitude of the current ripple in the load and the size of the isolation transformer. The series-parallel configuration also satisfies the basic requirements of a welding power supply circuit. The load has components in series and in parallel with it. The current can therefore be limited under short circuit conditions and the voltage limited under open circuit conditions. 1

1

1

Empc J I

controller

s2

O2

i"

I

Ls cs -----iFull series-parallel load-resonant converter I

Fig.3

ct

1.2 Existing welding power supplies Conventional power sources used in arc welding are a 50Hz stepdown transformer feeding a controlled rectifier [IO]. Electronic constant current power supplies have more recently been developed. These are inverters which are switched at around 20kHz and produce sine wave current in the transformer at a frequency of about 1kHz [ll-151. Higher frequency of operation in a welding power supply would improve the performance and quality of welding which can be achieved and reduce the physical size and weight of the unit. Traditional power supplies have a poor power factor. The current drawn from the supply is not in phase with the supply voltage. This aspect of the power supply design is becoming increasingly important, not only because increased efficiency is obtained when near unity power factor is achieved, but also because of forthcoming, stringent EMC legislation which restricts the harmonic content of the current drawn from the utility.

1.3 Proposed power supply The power supply presented in this paper is a true constant current power supply drawing near unity power factor. It is a series-parallel load-resonant converter, operating at 85 kHz, supplied by an active rectifier. High frequency operation is achieved in the seriesparallel load-resonant converter by using resonant switching techniques. The converter topology is such that the current through the power switches naturally commutates to zero. The converter is controlled so that the power switch is turned on or off at the instant the current is zero, minimising the switching losses. The high frequency of operation of the converter signifi212

Fig.4 Simpllfied model of series-parallel load-resonant converter

2

2. I

Series-parallel load-resonant converter

The converter

Fig. 3 shows the full series-parallel load-resonant converter. The converter is a halfbridge inverter supplying a resonant circuit. The load is the welding arc R L O A D connected to the circuit by an isolation transformer. Fig. 4 shows a simplified model of the series-parallel resonant circuit. The isolation transformer has been removed and replaced by an inductor, L L , which includes the magnitude of the leakage inductance of the transformer. This inductor is in series with a capacitor, CL, and the value of the load resistance, RLoADreferred to the primary side of the isolation transformer is R,. The series leg of the circuit and parallel legs of the circuit consist of an inductor, L,, and capacitor, C,, and an inductor, L,, and capacitor, C,, respectively. The resonant circuit, although simple in construction, is IEE Proc.-Electr. Power Appl., Vol. 143, No. 3, M a y 1996

complex 1.0 design in order to achieve the required current gain, resonant frequency and output voltage requirements.

2.2 Design analysis The circuit in Fig. 4 was analysed using simple AC theory. The impedance, ZTOT, of the circuit in Fig. 3 at an angular frequency w was first derived. This is given in eqn. 1. (alw”Uaw3 +asw) +J (bid + b W 4 + b y 2 + b 4 ) =

ZTOT

(qw5

+c p 3 +c3w)

(1)

where

al = c ~ c ; L ~ , c ; R ~

-2CSCpLpC;R~

04

a3

= CsC; RL

bl

1

b2

= C~C;CL(-L’$- ~ L s L L -~

+ L$LL + L s L i + L ~ L+~~ , L ~ L

CsC;C?(LpLi

~ L ~ ) L P LL 2LsLp)

+ cscpC;(-L: - 2LPLL 2LSLL + c;C;(-L: - 2LPLL - L2,) + c,c;c;R~,(L~ +L ~ ) -

b3

-

2LsLp)

T o obtain an effective design method for the seriesparallel resonant circuit and to ensure that the resonant frequencies of the circuit could be specified prior to the design stage it was decided that the earlier equations needed to be reversed. The designer would specify the three resonmt frequencies of the circuit, the referred resistance of the load R L , the load inductance, LL, the load capacitance, C, and the desired resistance of the complete circuit RTOT at the resonant frequency of operation. Solution of the appropriate system of equations would allow the remaining four component values to be calculated. Four independent equations were required in order to determine values for C,, L,, C,, L, so that the resulting circuit had the desired characteristic. Three equations were derived from the impedance, ZTOT, of the circuit given in eqn. 1. The coefficients of the imaginary part of this expression were shown to be related to the resonant frequencies of the circuit in eqn. 3. To simplify the mathematics, the reactance of each leg of the circuit was introduced into the equations replacing the capacitive terms. The reactance of each leg of the circuit is related to the component values in that leg and the operating frequency as follows:

+ ~ L L C+L ~ L s C I , C; Ri) + CsC;(LL + L s ) -

X L = W L L-

= -CsCp - CSCL- C;

e1 = c,C;(c;Lz,

-

(4)

~

WCP 1 ~

WCL

Eqn. 4 was rearranged to give equations for C,, C,, CL which where substituted into the coefficients of the imaginary part of eqn. 1 to give an equation for each of k, ,kb, k,. in terms of reactance, inductance and primary operating resonant frequency coo.

+ c ; c L ( - c I , R ~ + ~ L +L 2 ~ p ) + 2CPC3LL + L p ) b4

1

xp = w L p -

= CSCP(LPCP i- LSCP i2LPCL

2 C p c ~- C:

+ 2Lpc;LL + L $ c ; )

~2

= CsC;(CZRi - ~ C L L-L 2 L p c ~ )

e3

+ csCp(-2C;LL 2LpCi)‘ = csc; + 2CSCPCL + CSC?

1

-

The impedance equation has a real and imaginary part. At a resonant frequency of the circuit, the circuit appears totally resistive and the imaginary numerator of the impedance equation equals zero.

+

+

+

w6 k a W 4 k b W 2 k , = 0 (2) In eqn. 2 the coefficients k, ,kb, k, are related to the coefficients in the general impedance equation as follows: k.= h / b i kb = b 3 / h

I%, = -w; - wf - w,” = w;w;

+ wfwz” + w;w;

I

Lp--woxp)(w;LL-woxL

- (U;

k~ b 4 / h The imaginary part of the numerator of the impedance equation may be regarded as a polynomial of the sixth order in w or a cubic polynomial in w2 The roots are the resonant frequencies of the circuit and are defined as w = -+coo, w = f w l and w = +w2. The resonant frequencies of the circuit and the coefficients of eqn. 2 are therefore related: kb

kb=

)Z

(wii’L L - w o x L )(U; Lp--woXp)2 - (wip L ~ - w o x ~ ) ( w ~ L S - w o x ~ , ) ~ LS-WOXS )(WiLP - w o x P l2 2 (LJ,” Ls-Woxs)(LJ~LpiJoxp)(W~ L.r,-Wox~) k, = _-_ L p L i L2pLL LsL2, LSL%+2LSLPLL -

(4 +

+

+

(7) These were the first three system equations. T o solve for the fourth component in the circuit a further equation was required. The resistance of the circuit at resonance was

(3)

kc = -wo”wl”w,” IEE Proc.-Electr. Power Appl., Vol. 143, No. 3, M a y 1996

213

Eqns. 5-8 were now solved simultaneously to obtain the required reactances and subsequently for the component values. Eqns. 5-7 were rearranged into three equations for Ls,the value of the series inductance, of the converter.

3 Design and construction of the series-parallel load-resonant converter

A series-parallel resonant converter was designed using the analysis in section two. The upper resonant frequency at which the circuit would run was chosen as 8SkHz. The other resonant frequencies were defined as W O = 2~ 85 000

67 500 = 2ir 40 000

~1 = 2~ ~2

+ ( U i L P --WoXp) ( W i L L - W O x L ) 2 + ( U i L L -WOxL)(w;LP -WOXP)2 -WO xs (WO L L - w ~ X- w ~O x) s(w0 ~2Lp -uoXp)2 -2 w O x s(U: Lp w O X p (U: ) L L - w O x L) -

I

The resistance of the circuit at resonance was specified by designing for a peak current of 200A in the welding arc and assuming a load resistance of 0.12Sohms. The isolation transformer was wound with a turns ratio of 11: 1 resulting in a referred resistance of the welding arc of 18.6!2, taking into account the distortion of the rectifier. The peak output power = 12R = 2002 0.125 = 5 kW. The half bridge inverter had a maximum DC supply of 600V when the circuit was delivering rated power. This applied a f300V square wave across the circuit. The RMS value of the fundamental component of the square wave voltage supplied across the resonant circuit was

2 1

Vorrns = 600--

Tz/z

= 270 V

To achieve power balance and assuming 80% efficiency with this voltage the required total resistance of the resonant circuit at resonance, RTor was found to be 11.7Q. The component values calculated using the mathematical analysis from these initial conditions were C, = 87nF, Ls= 112pH, L, = 34m, C, = 4SnF and LL = 8OpH. Fig. 5 shows the magnitude and phase plot of the switch current, parallel leg current and load current against frequency resulting from the design procedure. The circuit was simulated to investigate the short circuit and open circuit characteristics and then constructed and tested.

80 120 160 f , kHz Fig.5 Magnitude and phase of switch current, parallel leg current and load current against frequency

0

An expression for X , was obtained by equating the two equations for X,. This resulted in a cubic polynomial in X,. The three solutions to this polynomial were found using symbolic analysis in Mathernatica. A Mathematica program was written based on this system of equations. A system specified in terms of X,, LL,RL, RTon oo,col, w2, could be solved to find the required component values L,, C,, L, and C,. Mathernatica was chosen as the programming language so that the analysis could remain in a symbolic format. 214

40

Switch current (solid line); load current (broken line); switch current phase (dashes); C, = 87nF; L, = 112pH; L,, = 34pH; C,, = 45nF; LL = 8OpH;

RL = 18.6!J

3.7 Short circuit and open circuit simulation of converter The series-parallel resonant converter design was modelled and calculated using the Saber simulation software under short circuit and open circuit conditions of the load. Saber contains a ready-made library of common components but also allows models of devices or sections of circuits to be described using a language I E E Proc.-Electr. Power Appl., Vol. 143, No. 3, May 1996

called MAST. To speed up the simulation the circuit was used with ideal models of the power switches and the circuit was controlled by algorithms written in the MAST development language. The modelling of the circuit using ideal devices ignored the diode recovery of both the freewheeling diodes and the rectifier diodes, the switching characteristics of the IGBT and the gate drive requirements of the IGBT. This allowed greatly reduced simulation times while having a minimal effect on the accuracy of the results. The two-winding transformer, in the model, was an ideal transformer. This model had negligible magnetising inductance or leakage inductance. An extra inductor was placed in series with the primary winding of the transformer to simulate the leakage inductance of the high frequency transformer which was a resonant component in the load leg of the circuit. For the simulation of the open circuit and short circuit conditions two additional switches were placed in the converter model. One across the load allowing a short circuit to occur, the other in the load leg causing an open circuit. The circuit was controlled to run at a revised frequency of the circuit 3 0 p after a short circuit condition or open circuit condition occurred. This was the method of control considered in practice.

-4001 200

,

, 300

I

,

I

,

I

,

,

400

500 600 t, ws Short circuit simulation of load (switch current)

Fig. 6

,

700

When the open circuit condition occurred the circuit is driven at 47kHz. The voltage across the load leg reduced and fell to 68V within 0.3 seconds, which is within the British Standard requirements.

3.2 Construction of converter Fig. 3 showed the full series-parallel load-resonant converter. The converter was constructed using IGBTs. These were chosen due to their low on-state losses and their availability in a module, the SKMSOGBlOOD. This module contains two series connected 1000V, 50A IGBTs and two freewheeling diodes in parallel with each IGBT. Its construction reduces the inductance between the two power switches to a minimum and controls the current ringing and voltage overshoot during switching. The resonant components in the circuit were polypropylene capacitors, inductors wound on ferrite cores with air gaps to prevent saturation and the leakage inductance of the isolation transformer. Polypropylene capacitors were used due to their relatively constant value of capacitance over a wide frequency range and low value of inductance. The isolation transformer had a turns ratio of 11: 1 and was a U and I core combination of 3C85 material. The secondary coil of the isolation transformer was connected LO a diode bridge rectifier. Power diodes BYV54V200 were used in this bridge. The bridge supplied a resistive load with unidirectional current. The actual load had a resistance of 0.12552 and was constructed from 11 1.2R high power resistors in parallel. The resistors had a very low value of inductance.

I -.....,.,.,..-... ,.* * ,,

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c.,

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