electronics Article
Soft Switching DC Converter for Medium Voltage Applications Bor-Ren Lin Department of Electrical Engineering, National Yunlin University of Science and Technology, Yunlin 640, Taiwan;
[email protected]; Tel.: +886-912312281 Received: 12 November 2018; Accepted: 17 December 2018; Published: 18 December 2018
Abstract: A dc-dc converter with asymmetric pulse-width modulation is presented for medium voltage applications, such as three-phase ac-dc converters, dc microgrid systems, or dc traction systems. To overcome high voltage stress on primary side and high current rating on secondary side, three dc-dc circuits with primary-series secondary-parallel structure are employed in the proposed converter. Current doubler rectifiers are used on the secondary side to achieve low ripple current on output side. Asymmetric pulse-width modulation is adopted to realize soft switching operation for power switches for wide load current operation and achieve high circuit efficiency. Current balancing cells with magnetic component are used on the primary side to achieve current balance in each circuit cell. The voltage balance capacitors are also adopted on primary side to realize voltage balance of input split capacitors. Finally, the circuit performance is confirmed and verified from the experiments with a 1.44 kW prototype. Keywords: soft switching; asymmetric pulse-width modulation (APWM) converter; current doubler rectifier
1. Introduction Medium voltage dc–dc converters have been proposed and implemented to achieve high power density and high efficiency advantages for dc light rail transportation systems [1,2], dc microgrid systems [3,4], or industry power converters [5,6]. In those applications, the high side dc bus voltage is normally at 750 V. The 1200 V SiC or Insulated Gate Bipolar Transistor (IGBT) power switches can be used to convert 750 V dc bus voltage to low voltage output through high-frequency link dc–dc converters. However, SiC devices are expensive, and the switching frequency of IGBT devices is less than 60kHz. The series-connected switches [7,8] and series-connected dc–dc converters [9,10] can be used for medium voltage converters with 600V Metal-Oxide-Semiconductor Field-Effect Transistors (MOSFETs) power devices. These two approaches can reduce the voltage stress on power devices. However, the voltage stresses on each power switch are difficult and unbalanced. Therefore, power devices still have an unbalanced voltage stress problem. Three-level pulse-width modulation converters or resonant converters have been presented in [11–14] to lessen the voltage rating and switching loss on power devices. Modular converters with series or parallel connection have been developed in [15–17] for high voltage or current applications. However, the current balance in each circuit modular should be controlled well in order to distribute equal power in each modular. To solve the current balance issue in each circuit modular, the current balance control approaches have been discussed in [18,19] by using the passive magnetic component. This paper presents a high voltage dc–dc converter with three cascade half-bridge circuits on primary side to reduce the voltage and current ratings of power devices for high voltage and medium power applications, such as dc light trail transportation systems and three-phase ac–dc power converters. Voltage balance capacitors are also employed on the primary side in order to balance Electronics 2018, 7, 449; doi:10.3390/electronics7120449
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converters. Voltage balance capacitors are also employed on the primary side in order to balance To prevent current imbalance on each half bridge circuit, the magnetic coupling input split voltages. To (MC) current balance components are employed between each half bridge circuit. If the primary-side currents are unbalanced, then the primary-side and secondary-side voltages of MC component will decreased, or increased, in order to compensate the imbalance in primary-side currents. Asymmetric pulse-width modulation modulation approach approach is adopted to realize the soft switching turn-on characteristic for switches. Therefore, Therefore,the theswitching switchinglosses lossesofofpower power devices high frequency operation power switches. devices at at high frequency operation cancan be be reduced. Current doubler rectifiers are used onvoltage low voltage side in order to reduce the output reduced. Current doubler rectifiers are used on low side in order to reduce the output ripple ripple current. Theispaper is organized as follows. The circuit and diagram and operating principle are current. The paper organized as follows. The circuit diagram operating principle are presented presented in The Section 2. The circuit characteristics of theconverter proposedare converter are in discussed inIn Section 3. in Section 2. circuit characteristics of the proposed discussed Section 3. section In experiments Section 4, experiments are provided to demonstrate the effectiveness the developed circuit. Then, 4, are provided to demonstrate the effectiveness of the of developed circuit. Then, the the conclusion of the presented circuit is discussed in Section conclusion of the presented circuit is discussed in Section 5. 5. 2. Circuit Circuit Diagram Diagram and and Principles Principles of of Operation Operation 2. The developed developedhigh-frequency high-frequencylink linkdc–dc dc–dc converter is illustrated in Figure 1 to realize the The converter is illustrated in Figure 1 to realize the main main benefits soft switching operation, low switching loss, low ripple outputcurrent ripple current the benefits of softofswitching operation, low switching loss, low output and the and balance balance primary-side and secondary-side currents. Three half-bridge circuits with primary-series primary-side and secondary-side currents. Three half-bridge circuits with primary-series secondarysecondary-parallel configuration areproposed used in the proposed converter realize lowVvoltage stress parallel configuration are used in the converter to realize low to voltage stress in/3 on power V /3 on power switches S ~S , and low current stress I /6 on power diodes D ~D . Therefore, o 1 6 1 6 in switches S1~S6, and low current stress Io/6 on power diodes D1~D6. Therefore, 600 V power MOSFETs 600 used V power MOSFETs on the high primary side tooperation achieve high operation low are on the primaryare sideused to achieve frequency and frequency low conduction loss. and Voltage conduction loss. Voltage capacitors on Cf1the andprimary Cf2 are employed on the primary side to achieve balance capacitors Cf1 andbalance Cf2 are employed side to achieve a split voltages balance at a split voltages balance at V /3. Half-bridge circuits are operated under asymmetric pulse-width Vin/3. Half-bridge circuits arein operated under asymmetric pulse-width modulation. Therefore, the modulation. soft switching turn-on of powerand switches can be achieved, and the circuit soft switchingTherefore, turn-on ofthe power switches can be achieved, the circuit efficiency is improved. The efficiency coupling is improved. Thebalance magnetic balance [19] on cells, MC2to , are used 1 and side magnetic current [19]coupling cells, MCcurrent 1 and MC 2, are used theMC primary achieve on the primary to achieve currentcircuit. balance for each half-bridge Therefore, themodular current current balance side for each half-bridge Therefore, the currentcircuit. unbalance issue of unbalance is issue of modular converters is overcome. current rectifiers on the converters overcome. The current doubler rectifiersThe are used on doubler the secondary sideare to used accomplish secondary side to accomplish low ripple current on load side. low ripple current on load side.
vCin2
vCin1
Vin Cin1
vCf1 CS2 f1 Cin2 S3 vCf2
vCin3
S1
Cin3
S4 Cf2 S5 S6
CS1 a CS2
vC1 C1
b CS3 c CS4
vC2
ip1 vMC1,p MC1
ip2 v MC1,s C2 vMC2,p
T1
L1 D1
L2 T1
L3
D3 d T2 CS5 MC2 L5 vC3 ip3 e CS6 C vMC2,s D5 3 f T3
D2 L4
T2
Vo Co
Io Ro
D4 L6
T3
D6
1. Circuit configuration of the developed modular converter for medium FigureFigure 1. Circuit configuration of the developed modular converter for medium voltagevoltage applications. applications.
Figure 2 provides the voltage and current waveforms of the studied converter in a switching cycle.Figure Based2 on the pulse-width Figureconverter 2, eight operating steps provides the voltagemodulation and currentwaveforms waveformsshown of theinstudied in a switching can be observed in each switching cycle under steady state. Power switches S , S , and S have the 1 3 5 cycle. Based on the pulse-width modulation waveforms shown in Figure 2, eight operating steps can same gating signals, and S , S , and S have the same gating signals. The duty cycle of S , S , and 2 4cycle under 6 be observed in each switching steady state. Power switches S1, S3, and S5 have1the3same S denotes d and the duty cycle of S , S , and d. When S1duty , S3 , and S2 , S4 , 5 2 4 6 is 1 −signals. gating signals, and S2, S4, and S6 have the sameSgating The cycleS5ofare S1,active, S3, andthen S5 denotes and S are inactive. We can obtain that v = v and v = v . If S , S , and S are inactive, 1 3 Cin1 d and6the duty cycle of S2, S4, and S6 is Cf1 1 − d. When S1, Cf2 S3, andCin2 S5 are active, then5 S2, S4, and S6 and are S , S , and S are active, then v = v and v = v . For steady state operation, capacitor 2 4 Cin2 vCf2 = vCf2 inactive. We 6can obtain that vCf1 Cf1 = vCin1 and Cin2. If Cin3 S1, S3, and S5 are inactive, and S2, the S4, and S6 are voltagesthen vCin1vCf1 = v=Cin2 vCin3vCf2 = v=Cf1 = v. Cf2 Vin /3. state Beforeoperation, the systemthe analysis, thevoltages circuit parameters active, vCin2=and vCin3 For=steady capacitor vCin1 = vCin2on =
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vCin3 = vCf1 = vCf2 = Vin/3. Before the system analysis, the circuit parameters on the proposed circuit are the proposed circuit are follows: (1) thecapacitances same voltage Cf1 input = Cf2 = Cf , assumed as follows: (1) assumed the same as voltage balance Cf1balance = Cf2 = capacitances Cf, (2) the same split (2) the same input split capacitances C = C = C = C , (3) the same output capacitances of power in1 same in2 output in3 capacitances in capacitances Cin1 = Cin2 = Cin3 = Cin, (3) the of power switches CS1 = … = CS6 = switches CS1same = . . .dc= block CS6 = capacitances Coss , (4) the same C1 =turns C2 = ratio C3 = n C1c ,=(5) turns Coss, (4) the C1 = dc C2 block = C3 = capacitances Cc, (5) the same n2 the = n3same = n, (6) the ratio n = n = n = n, (6) the identical magnetizing inductances L = L = L = L , (7) the same m1 m2 m3 m 1 magnetizing 2 3 identical inductances Lm1 = Lm2 = Lm3 = Lm, (7) the same leakage inductances Llk1 = Llk2 = Llk3 leakage Llk1 =output Llk2 = Lfilter same output filter inductances L1 = . . . =and L6 lk3 = L lk
