Transformer-less converter concept for a grid-connection of thin-film photovoltaic modules Ulrich Boeke
Heinz van der Broeck
Philips Research Laboratories Philips Technologie GmbH Aachen, Germany
[email protected]
Institute of Automation Engineering University of Applied Science Cologne, Germany
Abstract— A transformer-less converter concept for gridconnected photovoltaic systems is proposed that combines a DC/DC converter front-end with a DC/AC inverter. The converter system has an earth-connected DC input, as required from many thin-film photovoltaic modules. The DC/DC converter increases the positive photovoltaic DC-bus voltage by its negative DC output voltage to supply a grid-connected 3-phase inverter. This architecture extends today’s power electronic converter topologies for thin-film photovoltaic modules considering their special requirements with the ambition to realize higher power conversion efficiencies at lower cost. Thin-film photovoltaic, DC-DC, DC-AC, converter, earth connection, efficiency
I. INTRODUCTION Photovoltaic power systems are discussed to contribute in a mix with other renewable energy sources to the energy supply of the 21st century. However, costs of electricity from photovoltaic modules are still higher compared with utility bulk power especially due to high prices of photovoltaic modules. A significant cost reduction of photovoltaic modules is expected especially from a mass production of thin-film photovoltaic modules that require less, however still complex, production steps than modules with crystalline solar cells [1]. The most interesting application of photovoltaic (PV) power systems are grid-connected power systems. In a mix with other grid-connected renewable energy technologies such as wind or biomass power plants, photovoltaic power systems generate electricity especially in summer times when wind is typically low and grids are often loaded additionally by airconditioning and other cooling systems [2]. Grid connected photovoltaic systems require DC/AC converters to feed electricity form a DC bus of photovoltaic modules into an AC utility grid. A converter has to load photovoltaic modules with an optimum impedance to operate modules in their maximum power point (MPP). It is important that a converter does not generate AC voltages between the photovoltaic module DC bus and earth to avoid earth-leakage currents. Photovoltaic modules and also a converter in a grounded cabinet may have significant parasitic capacitances to earth. Today’s most efficient photovoltaic converters for crystalline solar cell modules operate with high European Efficiencies of up to 97 % [3]. The term “European Efficiency” is explained in the Appendix.
Converters for thin-film photovoltaic systems have, on the other side, an additional requirement that limit European Efficiencies of state of the art converters to 95%. This requirement is an earth-connection of a photovoltaic DC-bus minus-pole to avoid negative electric fields in the photovoltaic modules to earth. That is discussed to avoid a transport of positive charged sodium ions from a module front glass into the transparent conductive oxide (TCO) layer of a thin-film module [4, 5]. Due to these requirements, such a DC-bus system is not able to supply directly a transformer-less, voltage-feed, grid-connected 3-phase inverter. Practical, DC/AC converter systems for thin-film photovoltaic systems are realized by combing either a DC/DC converter front-end with high frequency transformer with a grid-connected DC/AC inverter [6, 7] or by combining a DC/AC inverter with a grid frequency transformer depicted in Figure 1 [8]. Thus, these converters include two converter stages and both lowfrequency or high-frequency transformer stages reduce the converter efficiency by at least 2 % compared with transformer-less single-stage inverters for crystalline solar cells. Single-stage DC/AC converters for thin-film module applications without a transformer are also known that make use of the “flying inductor” concept [9, 10]. Products with this technology realize European efficiencies of 95 % too [11, 12]. To extend power electronic converter topologies for thin-film photovoltaic systems, this paper presents a transformer-less converter concept that promises a realization of higher conversion efficiencies. The basic idea of the proposed converter concept is to use a transformer-less DC/DC converter to increase a positive photovoltaic DC-bus voltage of e.g. +500 V by its output voltage of e.g. –350 V to a suited DC input voltage of a grid-connected 3-phase inverter illustrated with Figure 2. + L3
PV
Q1
Q3
L2 Q2
-
VAC C1
VDC Q4
L
L1 M
Grid frequency transformer
C2
~ N
Figure 1: Typical circuit diagram of a photovoltaic inverter with grid frequency transformer
978-1-4244-2279-1/08/$25.00 © 2008 IEEE
II. THIN-FILM PHOTOVOLTAIC MODULES A wide range of thin-film photovoltaic module technologies exists today that make use of different chemical composites. The proposed converter concept requires thin-film modules that are approved to realize DC bus voltages of at least 700 V. Table III in the Appendix documents modules that fulfill this requirement. A proper interaction of modules and converter requires the consideration of a module voltage range that is a function of temperature, solar radiation and load. A mathematical model to calculate that voltage range has been proposed in [13]. It considers the first module in Table III from Wuerth Solar as an example. For other modules in Table II it is estimated that they operate with output voltage between 55 % and 100 % of Voc(-10°C) too. The index “oc” stands for open-cathode as an acronym for an unloaded solar cell. Voc(-10°C) is often considered as worst case operation point where a photovoltaic module generates its maximum output voltage. The last column in Table III informs about the number of series connected modules per string that generate the required DC input voltage range for the proposed converter. These numbers of modules in series result first in MPP-voltages between 350 V…550 V and second DC-bus voltages of unloaded modules of maximum 700 V. Multiple strings can be connected in parallel to design systems with different nominal power levels. New thin-film modules using amorphous silicon technology degrade in the first weeks of operation. Manufacturers offer this as free over-power. That also increases the maximum module voltage, Voc up to 11 %. Thus it is preferred to install such photovoltaic systems in summer periods when module operating temperatures are not too low. III.
CONVERTER CONCEPT
The proposed converter concept consists of a DC/DC converter and a 3-phase DC/AC inverter depicted in Figure 2. The earth-connected DC voltage of a photovoltaic module string is the positive DC supply voltage of the inverter. Its value can change between a value slightly higher than the peak voltage of an AC line-to-ground voltage and a certain maximum value. The DC/DC buck-boost converter generates from the positive DC input voltage Vin = VDC.1 a second constant DC voltage VDC.2 that is the negative supply voltage for the inverter. That is similar to the converter concept in [14]. That converter concept, however, considers that PVmodules also generate the earth-connected negative DC supply voltage of an inverter that is not recommended for the use of thin-film PV modules as discussed above. Thus the converter concept in this paper extends the converter concept from [14] to fulfill the special requirements of thin-film PV-modules.
A 3-phase inverter has been chosen for the converter concept because of the lower current stress in buffer capacitors C4 and C5. Transformer-less 3-phase inverters for photovoltaic applications with crystalline solar cells and high efficiencies up to 98 % are known [15, 16]. Thus, the paper focuses in the following on the design of an efficient DC/DC buck-boost converter for the proposed converter concept. After that the operation of voltage-feed 3-phase inverter with two unbalanced DC supply voltages will be discussed. A. DC/DC Converter The converter system benefits from a realization of a DCDC converter that operates between zero and full-load with very high efficiency. Thus the use of a soft-switching activeclamped buck-boost converter is proposed that is depicted in Figure 3. The active clamping mechanism is realized by means of additional components switch Q2 with inverse diode D2 and snubber capacitor C2, clamping capacitor C6 as well as inductor L2. Classical PWM-controlled active-clamped DC/DC converters operate with fixed switching frequency [17, 18]. Hereby, the clamping voltage changes as a function of the relative power level and relative input to output voltage transfer ratio. One challenge of an active-clamped buck-boost converter design in the given application is a limitation of the clamping voltage Vclamp to a relative small value. A large clamping voltage results either in a low useful converter DC input voltage range or an unfavorable high voltage stress of the power semiconductors. To limit the clamping voltage a different control principle has been chosen for the activeclamped buck-boost converter in this paper. The buck-boost converter operates with two independent and constant regulated DC voltages VDC.2 and Vclamp at medium to maximum power levels. A variable switching frequency is a consequence of that operation “Mode 2”. Below a certain medium power level the converter is forced to operate with its maximum specified switching frequency in “Mode 1”. The clamping voltage is no longer regulated and drops below its regulated value of “Mode 2” that has been learned from an analytical converter model. In “Mode 1” the converter behavior has a floating transition from an activeclamping converter to a bi-directional soft-switching buckboost converter similar to the resonant-pole principle [19]. PV C1 D1
C4
Q1
Vin
IL1
L1
L2
IL2 Vinveter
Iplus PV
Q1 D1
Q3 D9
Q5 D5
Q7 D7
Vin +350 V.... +700 V .
C4 L3
L4
400 V 3-phase AC grid
C5
C3
D3
C2 D2
Q2
VDC.2
L1
Vinverter Q4 D4
VDC.2 +350 V
C5
Q6 D6
Q8 D8
C7
C8
C9
C6
D3
Vclamp DC/DC Converter
Iminus
DC/AC Inverter
Control
Figure 2: Principle circuit diagram of the proposed converter system
Figure 3: Principle circuit diagram of an active-clamped buck-boost converter
To study the converter behavior two converter models have been developed. An analytical converter model has been used to study the influence of the control principle with its switching frequency limitation in Mode 1 and regulated clamping voltage in Mode 2. That analytical model considers a loss-less, hard-switching converter. The second more complex converter model includes softswitching details e.g. capacitors C1, C2, C3, dead-time intervals between gate signals of Q1 and Q2 and current time functions in L1 and L2 during these dead time intervals. This second model can be calculated, however, only numerically. All details of the more complex model are considered in the converter time functions depicted in Figure 4 that considers data of the prototype documented in Chapter IV. A practical difference between both models is that the complex model considers a current drop in inductor L2 during the dead time after the turn-off of power semiconductor Q1. Due to that IL2(ta+tb) is slightly smaller than –IL2(t.a) in opposite to the definition of equation (9). Time interval tb is slightly shorter and the switching frequency is slightly higher than calculate with the analytical model. VGate(Q1)
On
VGate(Q2)
VGate(Q1)
Off 800 V VCE(Q1)
V(D3)
600 V
V(D3)
400 V 200 V
1 ⋅ Vin ⋅ t a L1 + L2 1 2 ⋅ [I 0 + ∆I] = ⋅ Vclamp ⋅ t b L2 1 2⋅ I0 = ⋅ [Vin + VDC.2 ]⋅ t c L2 1 2 ⋅ ∆I = ⋅ VDC.2 ⋅ (t b + t c ) L1 1 = ta + tb + tc fs 2 ⋅ ∆I =
30 A
(2) (3) (4) (5)
Pin = Vin ⋅ [I 0 ⋅ t a ⋅ f s − ∆I ⋅ t c ⋅ f s ]
(6)
Pout = VDC.2 ⋅ [I 0 ⋅ (t b + t c ) ⋅ f s + ∆I ⋅ t c ⋅ f s ]
(7)
Pin = Pout = P
(8)
I L 2 (t a + t b ) = −I L 2 (t a )
(9)
I0
Vin ⋅ VDC.2 2 ⋅ f s. max L1 ⋅ Vin + (L1 + L 2 ) ⋅ VDC.2 P 1 I0 = ⋅ L2 2 ⋅ ∆I ⋅ f s. max ⋅ Vin L1 + L 2 − Vin Vin + VDC.2 1
∆I =
I(L1)
∆I
∆I
I(L2) 10 A
0A
⋅
t a = (L1 + L 2 ) ⋅
-10 A
tc = -I(D3)
-20 A
tb =
ta 10
tb
1 f s. max
− ta − tc
20
30 time (µs)
Vclamp = L 2 ⋅
tc 40
50
2 ⋅ ∆I Vin
L2 ⋅ 2 ⋅ I0 Vin + VDC.2
-30 A
-40 A 0
(1)
In Mode 1, these equations have been solved to calculated the parameters ∆I, I, ta, tb, tc, Vclamp.
0V
20 A
1) Analytical DC/DC converter model The analytical converter model of a loss-less, hardswitching, active-clamped buck-boost converter considers a set of nine equations that includes: Difference equations (1), (2), (3), (4) of the effective • inductance in four time intervals ta, tb, tc, tb +tc of a switching frequency period, • The definition of the switching frequency (5), • The definition of input and output power (6), (7), (8), • A condition of the active-clamping mechanism to operate with constant average charge in a clamping capacitor (9).
2 ⋅ (I 0 + ∆I) tb
(10) (11)
(12) (13) (14) (15)
60
Figure 4: Numerical calculated time functions of an active-clamped buck-boost converter Vin = +350 V, VDC.2 = +350 V, Vclamp = +50 V, P = 3 kW, fs = 19.3 kHz
In Mode 2, these equations are used to calculate parameters fs, I, ∆I, ta, tb, tc as well as a current ratio A. Figure 5 till Figure 7 illustrate data calculated with parameters of the prototype documented in Chapter IV.
§V VDC.2 L 2 ¨ DC.2 + ¨ Vclamp Vin + VDC.2 © A= V L1 − L 2 DC.2 Vclamp
· ¸ ¸ ¹
Switching frequency (kHz) 60
(16)
50
Vin = 550 V
P ⋅ L1 + L 2 L2 − Vin Vin + VDC.2 ª A ⋅ (L1 + L 2 ) ⋅ Vclamp ⋅ (Vin + VDC.2 ) º + « » 2 « A ⋅ Vclamp ⋅ Vin ⋅ (Vin + VDC.2 ) » .......« » + ⋅ ⋅ ⋅ + + ⋅ ⋅ ( 1 A ) L V ( V V ) L V V 2 in in DC.2 2 in clamp » « « » A ⋅ Vclamp ⋅ Vin2 ⋅ (Vin + VDC.2 ) ¬ ¼
40
I0 =
30
Vin = 350 V 20 10 0
0
1000
(17)
2000
3000
Power (W) Figure 5: Calculated switching frequencies
∆I = A ⋅ I 0
(18) 2 ⋅ ∆I Vin
(12)
2 ⋅ (I 0 + ∆I) t b = L2 ⋅ Vclamp
(19)
t a = (L1 + L 2 ) ⋅
L2 ⋅ 2 ⋅ I0 tc = Vin + VDC.2
Clamping voltage (V)
60 50 Vin = 350 V
(13)
40 Vin = 550 V 30
Figure 5 till Figure 7 illustrate calculated operation parameters considering data of the prototype that is discussed in Chapter IV. The transition between Mode 1 and Mode 2 is at about 1200 W when operating with Vin = 350 V and at about 1750 W when operating with Vin = 550 V. The switching frequency is limited to 50 kHz in Mode 1. In Mode 2 the clamping voltage is regulated to 50 V and the switching frequency varies between 19 kHz and 50 kHz as function of input voltage and power level. Figure 7 illustrates that the converter operates also at noload with a peak current of 3.4 A in L2 at the time points when Q1 and Q2 are turned off. That allows soft- switching even at no-load.
20 10 0
0
1000
2000
3000
Power (W) Figure 6: Calculated clamping voltage
Inductor peak current (A) 25
20
Vin = 350 V 15
Vin = 550 V 10
5
0
0
1000
2000
3000
Power (W) Figure 7: Calculated inductor peak current in L1 and L2
2) DC/DC converter control The controller sub-circuit of the active-clamped buckboost converter, depicted in Figure 8, makes use of two PItype voltage regulators and a dual-peak current control circuit. The voltage controller of the buck-boost converter output voltage, VDC.2 generates the reference signal for the positive peak currents in inductors L1 and L2. Once the regulated current and energy levels in L1 and L2 has been reached comparator 2 generates a reset signal for a flip-flop that turnsoff Q1 and turns-on active-clamping switch Q2. The voltage controller of the clamping voltage, Vclamp generates the reference signal for the negative peak current of the inductor L2. Once this current level has been reached comparator 1 generates the set-signal for the flip-flop that turns-off Q2 and turns-on Q1. Additionally comparator 1 feeds a mono-stable circuit (a one-shot timer) that generates a pulse with an equivalent length equal to the time period of the maximum switching frequency. Thus the next set-signal for the flip-flop can be generated first if the inverted output of the mono-stable circuit (one-shot) is high again. Our converter prototype uses a LEM “LTSR 25-NP” current shunt to monitor the current time function in inductor L2. That current shunt generates with two primary turns an output signal of 0.05 V/A and it has a peak current range of ±40 A that is transferred into output signals of ±2.0 V. The output signals of the two voltage regulators are limited such that the peak current reference signals do not exceed signals equivalent to ±35 A. That is important e.g. during the start of the converter when capacitors C5 and C6 are charged with maximum allowed current to generate the nominal values of VDC.2 and Vclamp. The output signals of the flip-flop are feed first into a subcircuit that generates dead-time signals and second into a standard level-shifter (IR2213) to drive the two power semiconductors Q1 and Q2 of the buck-boost converter. The actual prototype uses constant dead-time signals to support the soft-switching of the converter. That can be principally extended to adapted dead-time generation that monitors the dV/dt of power semiconductor Q2 as given in [18].
Vclamp Comp 1
2.5 V
IL2
fs =
1 >> f g τ
(20)
Thus, only small filter components are required to sufficiently attenuate all high frequency harmonics of the PWM. As the grid frequency depending voltage drop at the filter components (Lf, RL ) is very low, the duty cycle of the PWM: V1 ( t ) =
t +τ / 2
V1 ( x) 1 ⋅ ⋅ dx τ t − τ³ / 2 Vin + VDC.2
(21)
almost follows the AC grid voltage:
(
ˆ ⋅ sin 2 ⋅ π ⋅ f ⋅ t Vg ( t ) = V g g
)
(22)
superposed by the lower DC link voltage UDC.2. Considering the modulation factor ˆ V g 0
