V-Band On-Chip Dipole-Based Antenna - IEEE Xplore

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 57, NO. 10, OCTOBER 2009

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V-Band On-Chip Dipole-Based Antenna I-Shan Chen, Student Member, IEEE, Hwann-Kaeo Chiou, Member, IEEE, and Nan-Wei Chen, Member, IEEE

Abstract—A V-band on-chip dipole-based antenna for 60 GHz wireless personal area network (WPAN) application is implepHEMT process. The fabricated mented using WIN 0.15 2 , including test pads. antenna has a compact size of 0.9 The antenna comprises a half-wavelength dipole element and two tilted and slotted dipole elements to realize a wider impedance bandwidth than conventional wire dipole antennas, and provides endfire radiation patterns with high front-to-back ratio. The antenna performance is characterized using S-parameter, two-antenna (identical), three-antenna, and radiation pattern measurement methods for return loss, transmission gain, absolute gain, and radiation patterns. Measurement results shows that the on-chip antenna achieves a fractional bandwidth of 24% (55 to 70 GHz), a transmission gain of (the separated distance ), an absolute gain of 3.6 dBi, a front-to-back ratio of 12 dB, and an half-power beamwidth of 60 in E-plane and H-plane. The measured and simulated results are shown in good agreements.

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mm

VSWR = 2 = 5 cm

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Index Terms—Dipole-based antenna, GaAs, millimeter-wave (MMW) antenna, on-chip antenna, pseudomorphic high electron-mobility transistor (pHEMT), V-band, wireless personal area network (WPAN).

I. INTRODUCTION

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HE growing demand for wireless personal area network (WPAN) technology has driven market toward 60 GHz band, which offers up to 7 GHz bandwidth (57–64 GHz) for high definition video streaming, intelligent transport system (ITS) devices for vehicles, wireless Gigabit Ethernet, and wireless point to multipoint connections, etc., [1], [2]. These millimeter-wave (MMW) applications call for the high integration level transceiver design to lower the form factor, weight, and production cost. Recently, GaAs pseudomorphic high electron-mobility transistor (pHEMT) technology has been developed for the high-volume production of 60 GHz integrated transceivers with high performance, small form factor, light weight, and acceptable production cost [3], [4]. Another important technology for millimeter application is on-chip antennas. On-chip antenna can be integrated with the transceiver to reduce the system size, and makes all needs components integrate on a chip. Moreover, on-chip antennas can be periodically arranged to realize a beam forming system

Manuscript received June 19, 2008; revised June 19, 2009. First published September 09, 2009; current version published October 07, 2009. This work was supported by the National Science Council under Contract NSC 96-2628-E008-001-MY3 and the 0.15 WIN pHEMT foundry service provided by Chip Implementation Center (CIC), Taiwan, R.O.C. The authors are with the Department of Electrical Engineering, National Central University, Jhongli City, Taoyuan County 32001, Taiwan, R.O.C. (e-mail: [email protected], [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TAP.2009.2031758

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that increase the sensitivity of the wireless systems. In the past, most V-band antennas are designed in a discrete manner on conventional dielectric substrates [5], [6]. Recently, researchers have employed multi-chip modules (MCM) to integrate V-band antennas into receiver [7]. However, the MCM needs bondwire to connect the receiver as a system-on-package (SoP) design. The bondwire inductance is sensitive to V-band circuit performance thus making production difficult. A recent study shows that an integrated MMW antenna on high-resistivity GaAs substrate can alleviate this problem [8]. On-chip antennas have already produced in 60 GHz radio front-ends based on pHEMT technology. For instance, Alping et al. demonstrate a pHEMT high gain active patch antenna for 60 GHz WLAN/WPAN applications [2]. However, this patch antenna is relatively large on thin film alumina substrate, and also requires bondwire to connect the power amplifier. Changyul et al. develop a V-band push-pull patch antenna with a self-oscillating mixer in 0.15 pHEMT technology [9]. Grindrod et al. demonstrate a V-band integrated patch antenna [10]. Popovic et al. present a 60 GHz monolithic active antenna array in pHEMT technology [11]. In aforementioned published papers, some important parameters of on-chip antennas, such as radiation patterns, and/or absolute gain, have not been measured due to the difficulty of measurement setup in V-band. This paper presents an on-chip dipole-based antennafabricated in WIN 0.15 pHEMT technology. The proposed antenna comprises a straight dipole element and two tilted and slotted dipole elements to realize endfire and directional patterns. The proposed tilted and slotted dipole elements are essential to the formation of the endfire and directional patterns. The authors use four measurement techniques, such as S-parameters, two-antenna (identical) method, three-antenna method, and radiation pattern measurement, to verify the on-chip antenna. This paper is organized as follows. Section II presents the design and analysis of the proposed antenna structures. Section III outlines the measurement techniques and experimental results for the antenna verifications. Section IV gives the conclusions. II. ANTENNA DESIGN AND ANALYSIS Fig. 1 illustrates the circuit configuration and substrate parameters of the on-chip antenna. The antenna comprises a halfwavelength dipole radiating element and two tilted and slotted dipole elements. A coplanar stripe line (CPS) feeds the antenna with two parallel radial open stubs, and this feed configuration is preferable for the fully differential circuits to avoid the common mode interference. The dimensions of the antenna are specified , , , as follow, , , , , , , , ,

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Fig. 1. Configuration of the dipole-based on-chip antenna with two tilted and slotted dipole elements (a) three dimensional view, (b) cross sectional view, and (c) top view.

, and . The following investigation employs the commercial software package Ansoft HFSS to characterize the performance of antenna. Unlike a conventional dipole antenna, the antenna configuration uses two tilted and slotted dipole elements connected to a dipole element to realize endfire radiation patterns. This design is essentially motivated by array theory. The antenna results in multi-path current flows on the antenna plane and hence the antenna bandwidth is increased. As shown in Fig. 1(c), the tilted and slotted dipole elements comprises two metallic stripes which are connected to the straight dipole element and the short-circuit end. The short-circuit end used in the antenna is a key to realize the high directivity endfire pattern and enhance the antenna bandwidth. Fig. 2(a) displays the current distribution on the antenna plane at 61 GHz. The closed path formed by the short-circuit end leads to multipath current flows. The current flows along the two parallel sections of each tilted and slotted element are in the same direction, whereas the current flow on the straight dipole element is in the opposite direction. Furthermore, the current density on the edges of both radial open stubs is relatively high and the direction of current flow is the same as the tilted and slotted dipole elements. As a result, the radiation characteristics of the antenna are mainly attributed to the straight dipole radiating element together with the two tilted dipole radiating elements. The radiation pattern is partially affected by the radial open stubs, especially at the E-plane. Namely, the radiation from the tilted and slotted dipole elements is similar to an endfire-like radiation pattern of Vee dipole antenna, but Vee dipole antenna has high backside radiation. In contrast, radiation from the straight dipole element forms a doughnut-shape pattern. Fig. 2(b) and (c) show the normalized E-plane (xy-plane) and H-plane (yz-plane) radiation patterns, respectively. These figures clearly show that the endfire patterns appear at both principle planes. The front-to-back ratio

Fig. 2. (a) The current distribution on the proposed antenna. (b) Normalized E-plane radiation patterns. (c) Normalized H-plane radiation patterns.

is greater than 12 dB and cross-polarization component is less at the E-plane and less than at the H-plane. than The half-power beamwidth at the E-plane and H-plane is about 60 , which is much narrower than that of the conventional dipole element (typically symmetric 150 at the E-plane, which is a relatively broad figure-of-eight field, and 360 at the H-plane).

CHEN et al.: V-BAND ON-CHIP DIPOLE-BASED ANTENNA

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Fig. 4. Normalized radiation patterns of the proposed antenna with different L . (a) E-plane; (b) H-plane.

Fig. 3. (a) The current distribution on the proposed antenna without the shortcircuit configuration. (b) Normalized E-plane radiation patterns. (c) Normalized H-plane radiation patterns.

For comparison, Fig. 3(a) presents the current distribution on the proposed antenna without the short-circuit configuration at

the resonant frequency. The current mainly flows on the upper titled dipole element since its input impedance is lower than the lower tilted dipole element or the straight dipole element. In this case, the upper titled dipole element is primarily responsible for the radiation patterns. Fig. 3(b) and (c) demonstrate the E-plane and H-plane radiation patterns of the antenna without the short-circuit configuration. The front-to-back ratio is only 5 dB and the cross-polarization component is about at both the E-plane and the H-plane. As compared with Fig. 2, the antenna directivity is lower than that with the short-circuit configuration, especially at the E-plane. Meanwhile, the backside and cross-polarization radiations are worse than that in Fig. 2.

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Fig. 5. Return loss (S ) of the proposed antenna with/without the radial open stubs and the planar half-wavelength dipole antenna.

The results indicate that the dipole-based antenna achieves a higher directivity and a better front-to-back ratio when it employs short-circuit configuration. The impact of the length on the radiation characteristics is investigated as below. Fig. 4(a) and (b) show the E-plane and H-plane radiation patterns with different at the resonant frequency. Fig. 4(a) shows the normalized radiation patterns at E-plane. The highest directivity and lowest cross-polarization is about quarter-wavelength radiation occur when the long at the resonant frequency. In contrast, at the H-plane, the antenna exhibits more directional patterns but has higher backside radiation as increases. Therefore, the antenna with quarter-wavelength long tilted and slotted dipole elements obtains well-behaved radiation characteristics. The radial open stubs are employed for impedance matching between the antenna and the CPS feed. Fig. 5 demonstrates that the antenna with the radial open stubs has much wider impedance bandwidth than that without the radial open stubs. Indeed, the impedance bandwidth of the antenna is better than that in conventional planar dipole antenna designs. The fractional bandwidth is 24% (55 to 70 GHz). The radial open stubs also impacts on the radiation patterns. As observed the current distribution in Fig. 2(a), the current flow between straight dipole element and radial open stubs is in the opposite direction which can suppress the backside radiation. Fig. 6(a) and (b) are the comparison of the normalized E-plane radiation pattern with/without the radial open stubs. The antenna without radial open stubs features less directional patterns, especially at the normalized E-plane (Fig. 6(b)). In the case of the E-plane, the backside radiation pattern becomes larger without using radial open stubs. III. MEASUREMENT RESULTS The antenna is fabricated on a semi-insulating GaAs substrate 500 thick, with a dielectric constant , and a low loss tangent of 0.005 in WIN 0.15 pHEMT technology. Fig. 7 shows a photograph of the fabricated antenna.

Fig. 6. Normalized E-plane radiation pattern of the proposed antenna (a) with the radial stubs and (b) without the radial stubs.

The measured return loss, transmission gain, absolute gain, and radiation patterns are presented as follows. A. Return Loss In order to de-embed the loading effect of the probe and isolate the influence of the input CPS feed on the input impedance of the proposed antenna, a standard SOLT (Short-Open-Load-Thru) calibration was performed by Agilent PNA 8361A vector network analyzer with a commercial Cascade impedance standard substrate (P/N 104–783 ISS) from pitch 100 MHz to 110 GHz frequency range [12]. Two 100 ground-signal (G-S) probes with low loss cables were used to measure the S-parameters. Fig. 8 shows the measured and simulated return losses of the antenna. The measured results agree well with the simulated ones. As shown in Fig. 8, the antenna resonates at about 61 GHz with a return loss of 30 dB. It can be concluded that the


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Fig. 7. Photograph of the fabricated dipole-based antenna.

Fig. 8. Simulated and measured return losses of the proposed on-chip antenna.

Fig. 9. The simulated and measured input impedance of the proposed antenna. (a) Real part and (b) imaginary part.

antenna obtains good quality and matching to the 50 termination. The measured 10-dB bandwidth is of 24% (55 to 70 GHz) which is greater than the simulated one (58 to 64 GHz). Indeed, the operating bandwidth is significantly wider than conventional resonant type antennas, such as dipole antenna and patch antenna. Again, the broad bandwidth is mainly attributed to the multi-path current produced by incorporating the tilted and shorted dipole elements. Fig. 9(a) and (b) show the measured and simulated real and imaginary parts of the antenna input impedance. The simulated results in Figs. 8 and 9 reasonably agree with the measured results. B. Antenna Transmission Gain and Absolute Gain Measurements The transmission gain of the antenna was measured on-chip with the technique presented in [17]. In transmission gain measurement, two identical on-chip antennas are placed face-to-face with a distance . One antenna is used as a transmitting antenna and the other as a receiving antenna. Fig. 10 plots the measured , transmission gain at different separations (R). At

Fig. 10. Measured transmission gain of the proposed antenna for different R.

the transmission gain is about at the antenna’s resonant frequency. Reference [12] proposes a desktop arrangement to determine the directional gain of an on-chip antenna. The antenna gain

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Fig. 11. Setup for absolute gain measurement. (a) block diagram, (b) photograph of measurement setup.

is measured using an on-wafer probe station, a ground-signal (GS) probe, impedance standard substrate (ISS) card and a vector network analyzer (VNA). Since a pair on-chip antennas sit on the metallic chuck of the probe station, the diffraction and multi-path effect always exists [13], which cause the discrepancy between the measurement and simulation results. Here, to minimize these uncertainties, a three-antenna measurement technique [14] is employed to obtain the antenna absolute gain of the on-chip antenna. The use of three-antenna method completely eliminates the need of unknown parameter, such as effective aperture, in two-antenna method; therefore, the antenna absolute gain can be accurately measured. Fig. 11(a) shows the block diagram of this measurement setup. Fig. 11(b) demonstrates the setup on the optical bench for precise alignment during the measurement. The AUT, 34.5-dBi V-band lens horn antenna (Flann 25810-TA), and 24-dBi V-band standard horn antenna (QuinStar QWH-VPRR00) are denoted respectively as the antenna A, B, and C and used as transmitting antenna or receiving antenna in the three-antenna measurement. The three power ratios, i.e., , , and , are measured in a face-to-face configuration with a distance of . It is noted that the distance should satisfy the far-field condition, which is equal to or greater than [12], where and are the largest aperture dimensions of the antennas and the free-space wavelength at the operating frequency. After measuring the

Fig. 12. Simulated and measured gain of the proposed antenna.

receiving power and the transmitting power , the antenna absolute gain of the proposed antenna (AUT) is calculated by applying the Friis transmission formula as follows:

(1)


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Fig. 13. Measurement setup for on-chip antenna radiation pattern. (a) block diagram, (b) cross view of the AUT and the probe station, (c) photograph of measurement setup.

Fig. 14. Simulated and measured radiation patterns of the proposed antenna at (a) E-plane and (b) H-plane pattern.

Note that a cable loss of 6.5 dB, an adapter loss of 2 dB, and an inter-connector loss of 2 dB are calibrated in the measurement. The measured peak gain of the on-chip antenna at the resonant frequency is 3.6 dBi. Fig. 12 shows the comparison between the simulated and measured antenna peak gain over the entire bandwidth. The measured peak gain is 0.8 dB less than the simulated one at the resonant frequency. The discrepancy increases as the frequency towards the high frequency. The discrepancy is attributed to the unwanted diffrac-

tion and scattering from the measurement facilities around the AUT. C. Radiation Pattern Characteristics Fig. 13(a) presents the setup block diagram for measuring on-chip antenna radiation patterns. Fig. 13(b) and (c) demonstrate the entire measurement setup and the setup in the vicinity of the antenna under test (AUT), respectively. In the setup, the AUT is mounted on a glass plate for measurement on the

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TABLE I PERFORMANCE COMPARISON OF THE RECENT V-BAND ON-CHIP ANTENNA DESIGNS

radiation characteristics. Both the AUT and the glass plate are supported by a metallic chuck. To minimize the impact of the metallic chuck, the AUT was extended 5 mm out from the edge of the metallic chuck. The RF signal was fed by a ground-signal (GS) on-wafer probe, which provided the differential mode signal. To measure on-chip antenna radiation patterns, the AUT was placed on a rotating table that can be adjusted with a horizontal plate, a vertical plate, and an angle plate. Fig. 13(b) shows a cross-section view of the AUT. Note that the lens horn antenna must be accurately aligned with the line of sight of the AUT to minimize unwanted scattering. Unlike the setup used in [15], the proposed measurement setup does not call for a custom-made hemisphere silicon lens or horn antennas. Fig. 14(a) and (b) respectively display the measured radiation patterns at the E- and H- planes compared to the simulated patterns. Owing to measurement facility limitations, this study does not provide the backside radiation patterns. Also, this limitation leads to a small data gap for the H-plane pattern. According to Fig. 14, the measured and simulated patterns are in reasonably good agreement. The measured beam is narrower than the

simulated beam at both the E- and H-planes. Besides, the measured main beam deviates at the H-plane. These discrepancies are caused by scattering from the probe. Table I compares the proposed antenna to previous studies [7]–[10], [15]–[19]. This table demonstrates that the proposed antenna features a higher gain, a wider bandwidth, and a smaller size among those in previous antenna designs. IV. CONCLUSION This study demonstrates a V-band on-chip dipole-based antenna which implemented using WIN 0.15 pHEMT process. The fabricated antenna has a compact size of 0.9 . The antenna performance is verified using various on-chip antenna techniques, two-antenna, three-antenna, and radiation pattern measurement methods for return loss, transmission gain, absolute gain, and radiation patterns. The measured fractional bandwidth of the proposed dipole-based antenna is 24% (55 to 70 GHz). Also, the demonstrated antenna exhibits endfire radiation patterns along with high gain and high front-to-back ratio. The designed differential-fed on-chip


dipole-based antenna achieves the transmission gain of (the separated distance ), the absolute gain of 3.6 dBi, the front-to-back ratio of 12 dB, and half-power beamwidth of 60 in E-plane and H-plane. These results demonstrate that the proposed antenna shows a strong potential for integration into monolithic millimeter-wave integrated circuits and single-chip wireless systems for short range communication applications.

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[17] Y. P. Zhang, L. H. Guo, and M. Sun, “High transmission gain inverted-F antenna on low-resistivity Si for wireless interconnect,” IEEE Electron Device Lett., vol. 27, pp. 374–376, May 2006. [18] Y. P. Zhang, M. Sun, and L. H. Guo, “On-chip antennas for 60-GHz radios in silicon technology,” IEEE Trans. Electron Devices, vol. 52, pp. 1664–1668, Jul. 2005. [19] S.-S. Hsu, K.-C. Wei, C.-Y. Hsu, and R.-C. Huey, “A 60-GHz millimeter-wave CPW-fed Yagi antenna fabricated by using 0.18- CMOS technology,” IEEE Electron Device Lett., vol. 29, pp. 625–627, Jun. 2008.

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I-Shan Chen (S’09) was born in Taipei in 1979, Taiwan, R.O.C. He is currently working toward the Ph.D. degree at the National Central University, Jhongli, Taiwan, R.O.C. His current research interests include the areas of millimeter-wave antennas, radio frequency integrated circuits, monolithic microwave and millimeter-wave integrated circuits in CMOS, pHEMT, and compound semiconductor technologies.

Hwann-Kaeo Chiou (M’05) was born in Taichung, Taiwan, R.O.C., in 1959. He received the B.S. degree in electrical physics from National Chiao Tung University, Hsinchu, R.O.C., in 1982, and the M.S.E.E degree and Ph.D. degrees in electrical engineering from National Taiwan University, Taipei, Taiwan, R.O.C., in 1985, and 1997, respectively. From 1985 to 2000, he was an Associate Researcher with the Chung-Shan Institute of Science and Technology (CSIST), where he was in charge of development of microwave integrated circuits (MICs) and MMICs and microwave subsystems for mobile communication. From 2000 to 2002, he was an Associate Vice President with BenQ Inc., where he was in charge of development of broadband wireless access technology. In August 2002, he joined the faculty of the Department of Electrical Engineering, National Central University, Jhongli, Taiwan, R.O.C., where he is currently a Professor. His current research interests include microwave and millimeter-wave passive components, RF integrated circuits (RFICs), MMICs, and millimeter-wave integrated circuits.

Nan-Wei Chen (M’03) received the B.S. degree in atmospheric sciences and the M.S. degree in space sciences from National Central University, Jhongli, Taiwan, in 1993 and 1995, respectively, and the Ph.D. degree in electrical engineering at the University of Illinois at Urbana-Champaign, in 2004. From 1998 to 2004, he was a Research Assistant at the Center for Computational Electromagnetics, University of Illinois, where he worked on time-domain integral equation methods for the solution of scattering and radiation problems. Since 2004, he has been an Assistant Professor of electrical engineering at the National Central University, Taiwan. His research interests include computational electromagnetics with special emphasis on time-domain integral-equations, periodic structures, and millimeter wave antennas. Prof. Chen received the Raj Mittra Outstanding Research Award from the University of Illinois in 2004.